TA2022

TA2022 TRIPATH | Alldatasheet

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Tripath Technology, Inc. - Technical Information 1 TA2022 – KLI/1.2/07-04 TA2022 STEREO 90W (4 Ω) CLASS-T™ DIGITAL AUDIO AMPLIFIER DRIVER USING DIGITAL POWER PROCESSING (DPP™) TECHNOLOGY Technical Information Revision 1.2 – JULY 2004 GENERAL DESCRIPTION The TA2022 is a 90W (4Ω) continuous average per channel Class-T Digital Audio Power Amplifier IC using Tripath’s proprietary Digital Power Processing (DPPTM) technology. Class-T amplifiers offer both the audio fidelity of Class-AB and the power efficiency of Class-D amplifiers.

APPLICATIONS

¾ DVD Players ¾ Mini/Micro Component Systems ¾ Home Theater ¾ Powered Speakers BENEFITS ¾ Fully integrated solution with internal FETs ¾ Dramatically improves efficiency versus Class-AB amplifiers ¾ Signal fidelity equal to high quality linear amplifiers ¾ High dynamic range compatible with digital media such as CD and DVD

FEATURES

¾ Class-T architecture ¾ High Power ¾ 100W @ 4Ω, 1.0% THD+N ¾ 90W @ 4Ω, 0.1% THD+N ¾ 60W @ 8Ω, 0.1% THD+N ¾ “Audiophile” Sound Quality ¾ 0.015% THD+N @ 70W 4Ω ¾ 0.015% THD+N @ 45W 8Ω ¾ 0.10% IHF-IM @ 25W 4Ω ¾ High Efficiency ¾ 92% @ 88W 8Ω ¾ 87% @ 125W 4Ω ¾ Dynamic Range = 102 dB ¾ Mute Input ¾ Integrated Gate Drive Supply ¾ Over-current protection ¾ Over and under-voltage protection ¾ Single ended outputs ¾ Outputs can be operated in bridged mode ¾ 32-pin SSIP package

Tripath Technology, Inc. - Technical Information 2 TA2022 – KLI/1.2/07-04 ABSOLUTE MAXIMUM RATINGS (Note 1) SYMBOL PARAMETER Value UNITS VPP, VNN Supply Voltage (VPP1, VPP2, VNN1, VNN2) +/- 40 V V5 Positive 5V Bias Supply Voltage at Input Pins (pins 18, 19, 23, 24, 26, 28-32) -0.3V to (V5+0.3V) V V VN10 Voltage for low-side FET drive VNN+13 V TSTORE Storage Temperature Range -55º to 150º C TA Operating Free-air Temperature Range (Note 2) -40º to 85º C TJ Junction Temperature 150º C ESDHB ESD Susceptibility – Human Body Model (Note 3) All pins (except pin 27) Pin 27 4000 1500 V V ESDMM ESD Susceptibility – Machine Model (Note 4) All pins 200 V Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. See the table below for Operating Conditions. Note 2: This is a target specification. Characterization is still needed to validate this temperature range. Note 3: Human body model, 100pF discharged through a 1.5KΩ resistor. Note 4: Machine model, 220pF – 240pF discharged through all pins. OPERATING CONDITIONS (Note 5) SYMBOL PARAMETER MIN. TYP. MAX. UNITS VPP, VNN Supply Voltage (VPP1, VPP2, VNN1, VNN2) +/- 12 +/-31 +/- 36 V V5 Positive 5 V Bias Supply 4.5 5 5.5 V VN10 Voltage for FET drive (Volts above VNN) 9 11 12 V Note 5: Recommended Operating Conditions indicate conditions for which the device is functional. See Electrical Characteristics for guaranteed specific performance limits. THERMAL CHARACTERISTICS SYMBOL PARAMETER VALUE UNITS θJC Junction-to-case Thermal Resistance 1.0° C/W θJA Junction-to-ambient Thermal Resistance (still air) 20° C/W

Tripath Technology, Inc. - Technical Information 3 TA2022 – KLI/1.2/07-04 ELECTRICAL CHARACTERISTICS (Notes 6, 7) TA = 25 °C. See Application/Test Circuit on page 8. Unless otherwise noted, the supply voltage is VPP=|VNN|=31V. SYMBOL PARAMETER CONDITIONS MIN. TYP. MAX. UNITS Iq Quiescent Current (No load, Mute = 0V) VPP = +31V VNN = -31V (Note 8) V5 = 5V (Note 9) VN10 = 11V (Note 10) mA mA mA mA IMUTE Mute Supply Current (No load, Mute = 5V) VPP = +31V VNN = -31V (Note 8) V5 = 5V (Note 9) 0.5 mA mA mA VIH High-level input voltage (MUTE) I IH = See Mute Control Section 3.5 V VIL Low-level input voltage (MUTE) I IL = See Mute Control Section 1.0 V VOH High-level output voltage (HMUTE) I OH = 3mA 4.0 V VOL Low-level output voltage (HMUTE) I OL = 3mA 0.5 V VOFFSET Output Offset Voltage No Load, MUTE = Logic low 0.1% RFBA, RFBB, RFBC resistors -750 750 mV IOC Over Current Sense Threshold TBD 7 8 A IVPPSENSE VPPSENSE Threshold Currents Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) 138 162 154 178 µA µA µA µA VVPPSENSE Threshold Voltages with RVPPSENSE = 249KΩ (Note 11, Note 12) Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) 36.5 17.8 42.8 40.9 22.2 20.4 47.3 24.4 V V V V IVNNSENSE VNNSENSE Threshold Currents Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) 152 174 169 191 µA µA µA µA VVNNSENSE Threshold Voltages with RVNNSENSE = 249KΩ (Note 11, Note 12) Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) -36.2 -14.8 -42.1 -40.8 -20.2 -17.9 -46.8 -22.6 V V V V

Tripath Technology, Inc. - Technical Information 4 TA2022 – KLI/1.2/07-04 PERFORMANCE CHARACTERISTICS – SINGLE ENDED (Notes 6, 7) TA = 25 °C. Unless otherwise noted, the supply voltage is VPP=|VNN|=31V, the input frequency is 1kHz and the measurement bandwidth is 20kHz. See Application/Test Circuit on Page 8. SYMBOL PARAMETER CONDITIONS MIN. TYP. MAX. UNITS POUT Output Power (Continuous Average/Channel) (Note 13) VPP = |VNN| = +/-31V, R L = 4Ω THD+N = 0.1% THD+N = 1.0% THD+N = 10% VPP = |VNN| = +/-35V, R L = 8Ω THD+N = 0.1% THD+N = 10% 100 125 W W W W W THD + N Total Harmonic Distortion Plus Noise POUT = 70W/Channel, RL = 4Ω VPP = |VNN| = +/-31V POUT = 45W/Channel, RL = 8Ω VPP = |VNN| = +/-35V 0.015 0.015 IHF-IM IHF Intermodulation Distortion 19kHz, 20kHz, 1:1 (IHF), RL = 4Ω POUT = 25W/Channel 0.1 % SNR Signal-to-Noise Ratio A-Weighted 0dB = 90W/Channel, RL = 4Ω 102 dB CS Channel Separation 0dB = 25W, RL = 4Ω 83 dB AV Amplifier Gain POUT = 10W/Channel, RL = 4Ω, See Application / Test Circuit

18.1 V/V

AVERROR Channel to Channel Gain Error POUT = 10W/Channel, RL = 4Ω See Application / Test Circuit 0.5 dB η Power Efficiency POUT = 88W/Channel, RL = 8Ω POUT = 125W/Channel, RL = 4Ω ISLOAD Source Current POUT = 125W/Channel, RL = 4Ω VPP = +31V VNN = -31V V5 = 5V 4.59 4.61 A A mA eNOUT Output Noise Voltage A-Weighted, input AC grounded 150 µV PERFORMANCE CHARACTERISTICS – BRIDGED TIED LOAD (Notes 6, 7) TA = 25 °C. Unless otherwise noted, the supply voltage is VPP=|VNN|=30V, the input frequency is 1kHz and the measurement bandwidth is 20kHz. SYMBOL PARAMETER CONDITIONS MIN. TYP. MAX. UNITS POUT Output Power (Continuous Average) (Note 13) VPP = |VNN| = +/-30V, R L = 8Ω THD+N = 0.1% THD+N = 10% 150 235 W W THD + N Total Harmonic Distortion Plus Noise POUT = 100W, RL = 8Ω 0.05 % IHF-IM IHF Intermodulation Distortion 19kHz, 20kHz, 1:1 (IHF), RL = 8Ω POUT = 25W 0.10 % η Power Efficiency POUT = 225W, RL = 8Ω 87 % SNR Signal-to-Noise Ratio A-Weighted, RL = 8Ω 0dB = 150W 104 dB eNOUT Output Noise Voltage A-Weighted, input AC grounded 220 µV Note 6: Minimum and maximum limits are guaranteed but may not be 100% tested. Note 7: For operation in ambient temperatures greater than 25 °C, the device must be derated based on the maximum junction temperatur e and the thermal resistance determined by the mounting technique.

Tripath Technology, Inc. - Technical Information 5 TA2022 – KLI/1.2/07-04 Note 8: This specification includes the current draw from the internal buck regulator. If an external floating supply is used, instead of the internal buck regulator, the quiescent current draw of the VNN supply will be approximately 20mA. Note 9: This specification includes the curr ent draw from both the TA2022 and the external feedback biasing. Note 10: This is the current draw of the VN10 pin if an external “floating” 11V supply is used instead of the internal buck regulator Note 11: These supply voltages are ca lculated using the IVPPSENSE AND IVNNSENSE values shown in the Electrical Characteri stics table. The typical voltage values shown are calculated using a R VPPSENSE and RVNNSENSE value of 249kohm without any tolerance variation. T he minimum and maximum voltage limits shown include either a +1% or –1% (+1% for Ov er-voltage turn on and Under-voltage turn off, -1% for Over-voltage turn off and Under-voltage turn on) variation of RVPPSENSE or RVNNSENSE off the nominal 249kohm value. These voltage specifications are examples to show both typical and worst case voltage ranges for a given RVPPSENSE and RVNNSENSE resistor va lue of 249kohm. Please refer to the Application Information section for a more detailed description of how to calculate the over and under voltage trip voltages for a given resistor value. Note 12: The fact that the ov er-voltage turn on and over-voltage turn off specifications exceed the absolute maximum of +/-40V for the TA2022 does not imply that the part will work at these elevated supply voltages. It also does not imply that TA2022 is tested or guaranteed at these supply voltages. The s upply voltages are simply a calculation based on the process spread of the I VPPSENSE and I VNNSENSE currents (see Note 11). The supply voltage must be maintained below the absolute maximum of +/-40V or permanent damage to the TA2022 may occur. Note 13: The supply voltage limitation for 4 ohm single ended (+/-31V), or 8 ohm bridged (+/- 30V), is based on the current limit protection ci rcuitry. The current limit circuitry may be activated during large output excurs ions if the recommended supply voltage ranges are exceeded. This will result in the amplifier being muted.

Tripath Technology, Inc. - Technical Information 6 TA2022 – KLI/1.2/07-04 TA2022 PINOUT PIN DESCRIPTION Pin Function Description 1, 13 VBOOT2, VBOOT1 Bootstrap voltages for gate drive of high side MOSFET’s 2 VN10 “Floating” supply input. Normally connected to the output of onboard VN10 buck converter. This voltage must be stable and referenced to VNN. 3 VN10GND Power ground for onboard VN10 generator. Electrically tied to the TA2022 case. 4, 12 VPP2, VPP1 Positive power supply input pins. 5 VN10SW Switching output voltage for onboard VN10 generator (buck converter). 6 NC Not connected internally. May be connected to pin 7 without any loss of functionality or performance. 7,10 OUT2, OUT1 Power amplifier outputs. 8, 9 VNN2, VNN1 Negative power supply inputs. 11 NC Not connected internally. May be connected to pin 10 without any loss of functionality or performance.

14 VN10FDBK Feedback for onboard VN10 generat or (nominally 11V above VNN)

15, 20 AGND Analog Ground. 16, 21 V5 5V power supply input. 17 REF Used to set internal bias current s. The pin voltage is typically 1.1V. 18 VNNSENSE Negative supply voltage sense input. This pin is used for both over and under voltage sensing for the VNN supply. 19 VPPSENSE Positive supply voltage sense input. This pin is used for both over and under voltage sensing for the VPP supply. 22, 25 OAOUT1, OAOUT2 Outputs of Input Stage op amps. 23, 26 INV1, INV2 Inverting inputs of Input Stage op amps. 24 MUTE Logic input. A logic high puts the amplifie r in mute mode. Ground pin if not used. Please refer to the section, Mute Control, in the Application Information. 27 BIASCAP Bandgap reference times two (typically 2.5VDC). Used to set the common mode voltage for the input op amps. This pin is not capable of driving external circuitry. 28, 29 FBKGND2, FBKOUT2 Output volt age differential feedback for channel 2. 30,31 FBKGND1, FBKOUT1 Output voltage differential feedback for channel 1. 32 HMUTE Logic Output. A logic high indicates both amplifiers are muted, due to the mute pin state, or a “fault” such as an overcurrent, undervoltage, or overvoltage condition. FBKGND1 AGND VNNSENSE VPPSENSE AGND OAOUT1 INV1 MUTE OAOUT2 INV2 VBOOT2 BIASCAP FBKGND2 FBKOUT2 FBKOUT1 VN10FDBK VBOOT1 VPP1 NC OUT1 VNN1 VNN2 OUT2 NC VPP2 VN10GND VN10 HMUTE 32-pin SSIP Package (Front View) REF VN10SW

Tripath Technology, Inc. - Technical Information 7 TA2022 – KLI/1.2/07-04 APPLICATION /TEST CIRCUIT TA2022 RL 4Ω or 8Ω HMUTE OAOUT1 INV1 OUT1 CI 3.3uF VPP1 VNN1 Processing Modulation CO 0.22uF LO 10uH, 10A Analog Ground Power Ground (Pin 9) RF 20KΩ RI 20KΩ CZ 0.22uF RZ 6.2Ω, 2W CB 0.1uF CBAUX 47uF VBOOT113 RB 250Ω VNN1 VN10 DB 11DQ09 VN10 FBKOUT131 FBKGND130 RFBA 1KΩ *RFBB 1.1KΩ *RFBB 1.1KΩ RFBA 1KΩ DO MUR120 *RFBC 9.1KΩ *RFBC 9.1KΩ V5 (Pin 21) AGND (Pin 20) AGND CFB 390pF CS 0.1uF CS 0.1uF AGND AGND V516 OAOUT2 INV2 OUT2 CI 3.3uF 26 7+ VPP2 VNN2 Processing Modulation CO 0.22uF LO 10uH, 10A (Pin 8) RF 20KΩ RI 20KΩ CZ 0.22uF RZ 6.2Ω, 2W CB 0.1uF CBAUX 47uF VBOOT21 RB 250Ω VNN2 VN10 DB 11DQ09 VN10 FBKOUT229 FBKGND228 RFBA 1KΩ *RFBB 1.1KΩ *RFBB 1.1KΩ RFBA 1KΩ DO MUR120 *RFBC 9.1KΩ *RFBC 9.1KΩ V5 (Pin 21) AGND CFB 560pF MUTE 24 BIASCAP CA 0.1uF 27 17REF RREF 8.25KΩ, 1% 12VPP1 9VNN1 +CHBR 0.1uF,100V Cs 220uF, 50V CS 0.1uF, 50V Cs 220uF, 50V CS 0.1uF, 50V 8VNN2 4VPP2 CHBR 0.1uF,100V VPP VNN VN10GND *RVPPSENSE(VPP1) VPP VNN

19 VPPSENSE

18 VNNSENSE

AGND (Pin 20) VN10SW5 CSWFB 0.1uF,50V VN10FBK14 VNN CSW 0.1uF,35V VN102 VNN DSW 11DQ09 LSW 100uH, 1A RSWFB 1KΩ CSW 100uF, 35V VN10 RL 4Ω or 8Ω 2.5V 200KΩ (Pin 20) (Pin 20) (Pin 20) ROFA 50KΩ ROFB 10KΩ to FBKGND1 (Pin30) Offset Trim Circuit V5 (Pin 21) AGND (Pin 20) VNN 249KΩ, 1% ROFA 50KΩ ROFB 10KΩ to FBKGND2 (Pin28) Offset Trim Circuit V5 (Pin 21) AGND (Pin 20) *RVNNSENSE(VNN1) 249KΩ, 1% 6 NC 11 NC *RVPP2 *RVNN2 Optional Components - See Application Information * The values of these components must be adjusted based on supply voltage range. See Application Information. VPP2 (Pin 4) DO MUR120 VPP1 (Pin 12) DO MUR120 Application / Test Diagram CS 0.1uF, 50V CS 0.1uF, 50V C S 0.1uF, 50V C S 0.1uF, 50V

Tripath Technology, Inc. - Technical Information 8 TA2022 – KLI/1.2/07-04 EXTERNAL COMPONENTS DESCRIPTION (Refer to the Application/Test Circuit) Components Description RI Inverting input resistance to provide AC gain in conjunction with RF. This input is biased at the BIASCAP voltage (approximately 2.5VDC). RF Feedback resistor to set AC gain in conjunction with RI. Please refer to the Amplifier Gain paragraph, in the Application Information section. CI AC input coupling capacitor which, in conjunction with RI, forms a high pass filter at )CR2(1f IIC π= . RFBA Feedback divider resistor connected to V5. This resistor is normally set at 1kΩ. RFBB Feedback divider resistor connected to AGND. This value of this resistor depends on the supply voltage setting and helps set the TA2022 gain in conjunction with RI, RF, RFBA, and RFBC. Please see the Modulator Feedback Design paragraphs in the Application Information Section. RFBC Feedback resistor connected from either the OUT1(OUT2) to FBKOUT1(FBKOUT2) or speaker ground to FBKGND1(FBKGND2). The value of this resistor depends on the supply voltage setting and helps set the TA2022 gain in conjunction with RI, RF, RFBA,, and RFBB. It should be noted that the resistor from OUT1(OUT2) to FBKOUT1(FBKOUT2) must have a power rating of greater than )(2RVPPP FBC2 DISS = . Please see the Modulator Feedback Design paragraphs in the Application Information Section. CFB Feedback delay capacitor that both lowers the idle switching frequency and filters very high frequency noise from the feedback signal, which improves amplifier performance. The value of CFB should be offset between channel 1 and channel 2 so that the idle switching difference is greater than 40kHz. Please refer to the Application / Test Circuit. ROFA Potentiometer used to manually trim the DC offset on the output of the TA2022. ROFB Resistor that limits the manual DC offset trim range and allows for more precise adjustment. RREF Bias resistor. Locate close to pin 17 and ground at pin 20. CA BIASCAP decoupling capacitor. Should be located close to pin 27 and grounded at pin 20. DB Bootstrap diode. This diode charges up the bootstrap capacitors when the output is low (at VNN) to drive the high side gate circuitry. Schottky or fast recovery diode rated at least 200mA, 90V, 50nS is recommended for the bootstrap circuitry. In addition, the bootstrap diode must be able to sustain the entire VPP-VNN voltage. Thus, for most applications, a 90V (or greater) diode should be used. CB High frequency bootstrap capacitor, which filters the high side gate drive supply. This capacitor must be located as close to pin 13 (VBOOT1) or pin1n (VBOOT2) for reliable operation. The other side of C B should be connected directly to the OUT1 (pin 10) or OUT2 (pin 7). Please refer to the Application / Test Circuit. CBAUX Bulk bootstrap capacitor that supplements CB during “clipping” events, which result in a reduction in the average switching frequency. RB Bootstrap resistor that limits CBAUX charging current during TA2022 power up (bootstrap supply charging). CSW VN10 generator filter capacitors. The high frequency capacitor (0.1uF) must be located close to pin 2 (VN10) to maximize device performance. The value of the bulk capacitor should be sized appropriately such that the VN10 voltage does not overshoot with respect to VNN during TA2022 turn on. Tripath recommends using a value of 100F for the bulk capacitor. LSW VN10 generator filter inductor. This inductor sized appropriately so that LSW does not

Tripath Technology, Inc. - Technical Information 9 TA2022 – KLI/1.2/07-04 saturate. If the recommended inductor value of 100H is not used, the VN10 may overshoot with respect to VNN during TA2022 turn on. DSW Flywheel diode for the internal VN10 buck converter. This diode also prevents VN10SW from going more than one diode drop negative with respect to VNN. This Diode can be a Fast Recovery, Switching or Shottky, but must be rated at least 200mA, 30V, 50nS. CSWFB VN10 generator feedback capacitor. This capacitor, in conjunction with RSWFB, filters the VN10 feedback signal such that the loop is unconditionally stable. RSWFB VN10 generator feedback resistor. This resistor sets the nominal VN10 voltage. With R SWFB equal to 1kΩ, the internally VN10 voltage will typically be 11V above VNN. CS Supply decoupling for the power supply pins. For optimum performance, these components should be located close to the TA2022 and returned to their respective ground as shown in the Application/Test Circuit. RVNNSNESE Overvoltage and undervoltage sense resistor for the negative supply (VNN). Please refer to the Electrical Characteristics Section for the trip points as well as the hysteresis band. Also, please refer to the Over / Under-voltage Protection section in the Application Information for a detailed discussion of the internal circuit operation and external component selection. RVPPSENSE Overvoltage and undervoltage sense resistor for the positive supply (VPP). Please refer to the Electrical Characteristics Section for the trip points as well as the hysteresis band. Also, please refer to the Over / Under-voltage Protection section in the Application Information for a detailed discussion of the internal circuit operation and external component selection. CHBR Supply decoupling for the high current Half-bridge supply pins. These components must be located as close to the device as possible to minimize supply overshoot and maximize device reliability. These capacitors should have good high frequency performance including low ESR and low ESL. In addition, the capacitor voltage rating must be twice the maximum VPP voltage. CZ Zobel capacitor, which in conjunction with RZ, terminates the output filter at high frequencies. Use a high quality film capacitor capable of sustaining the ripple current caused by the switching outputs. RZ Zobel resistor, which in conjunction with CZ, terminates the output filter at high frequencies. The combination of RZ and CZ minimizes peaking of the output filter under both no load conditions or with real world loads, including loudspeakers which usually exhibit a rising impedance with increasing frequency. The recommended power rating is 2 watts. DO Fast Recovery diodes that minimize overshoots and undershoots of the outputs with respect to power ground during switching transitions as well as output shorts to ground. For maximum effectiveness, these diodes must be located close to the output pins and returned to their respective VPP and VNN. Also, they should be rated with a maximum Forward Voltage of 1V at 10A. Please see Application/Test Circuit for VPP and VNN return pins. LO Output inductor, which in conjunction with C O, demodulates (filters) the switching waveform into an audio signal. Forms a second order filter with a cutoff frequency of )CL2(1f OOC π= and a quality factor of OOOL CLCRQ = . These inductors must be rated at least 10A with high linearity. Please see Output Filter Design section for details. CO Output capacitor, which, in conjunction with LO, demodulates (filters) the switching waveform into an audio signal. Forms a second order low-pass filter with a cutoff frequency of )CL2(1f OOC π= and a quality factor of OOOL CLCRQ = . Use a high quality film capacitor capable of sustaining the ripple current caused by the switching outputs. Electrolytic capacitors should not be used.

Tripath Technology, Inc. - Technical Information 10 TA2022 – KLI/1.2/07-04 TYPICAL PERFORMANCE CHARACTERISTICS – SINGLE ENDED 0.005 0.01 0.02 0.05 0.1 0.2 0.5 2 1005 10 20 50 Output Power (W) THD+N (%) THD+N vs Output Power f = 1kHz RL= 8Ω VPP=|VNN|=35V AES 17 Filter 0.005 0.01 0.02 0.05 0.1 0.2 0.5 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) THD+N (%) THD+N vs Frequency Po = 25W/ch RL = 8Ω VPP=|VNN|=35V BW = 22kHz BW = 30kHz 0.005 0.01 0.02 0.05 0.1 0.2 0.5 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) THD+N (%) THD+N vs Frequency Po = 50W/ch RL = 4Ω VPP=|VNN|=31V BW = 22kHz BW = 30kHz 0.005 0.01 0.02 0.05 0.1 0.2 0.5 1 2002 5 10 20 50 100 Output Power (W) THD+N (%) THD+N vs Output Power f = 1kHz RL= 4Ω VPP=|VNN|=31V AES 17 Filter -140 -130 -120 -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) Amplitude (dBr) Intermodulation Distortion 19kHz, 20kHz 1:1 Po = 25W/ch, 4Ω 0dBr = 10.0Vrms VPP=|VNN|=31V BW = 22Hz - 30kHz -140 -130 -120 -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) Amplitude (dBr) Intermodulation Distortion 19kHz, 20kHz 1:1 Po = 12.5W/ch, 8Ω 0dBr = 10.0Vrms VPP=|VNN|=31V BW = 22Hz - 30kHz

Tripath Technology, Inc. - Technical Information 11 TA2022 – KLI/1.2/07-04 TYPICAL PERFORMANCE CHARACTERISTICS – SINGLE ENDED TY 0.005 0.01 0.02 0.05 0.1 0.2 0.5 2 1005 10 20 50 Output Power (W) THD+N (%) THD+N vs Output Power f = 1kHz RL= 8Ω AES 17 Filter +/-25V +/-30V +/-35V -100 -40 -95 -90 -85 -80 -75 -70 -65 -60 -55 -50 -45 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) Channel Separation (dBr) Channel Separation VPP=|VNN|=31V Po = 25W/ch, 4Ω Po = 12.5W/ch, 8Ω BW = 22Hz - 22kHz RL = 8Ω RL = 4Ω -120 -70 -115 -110 -105 -100 -95 -90 -85 -80 -75 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) Amplitude (dBr) Noise Floor VPP=|VNN|=31V RF=RI=20kΩ 16kFFT AES 17 Filter 0.005 0.01 0.02 0.05 0.1 0.2 0.5 1 2002 5 10 20 50 100 Output Power (W) THD+N (%) THD+N vs Output Power f = 1kHz RL= 4Ω AES 17 Filter +/-23V +/-27V +/-31V 100 0 10 20 30 40 50 60 70 80 90 100 110 120 Output Power (W) Efficiency (%) VPP=|VNN|=31V RL = 4Ω AES 17 Filter THD+N < 10% Efficiency vs Output Power 100 0 1 02 03 04 05 06 07 08 09 0 Output Power (W) Efficiency (%) VPP=|VNN|=35V RL = 8Ω AES 17 Filter THD+N < 10% Efficiency vs Output Power

Tripath Technology, Inc. - Technical Information 12 TA2022 – KLI/1.2/07-04 TYPICAL PERFORMANCE CHARACTERISTICS – BRIDGED -140 -130 -120 -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) Amplitude (dBr) Intermodulation Distortion 19kHz, 20kHz 1:1 Po = 50W/ch, 8Ω BRIDGED 0dBr = 20.0Vrms VPP=|VNN|=30V BW = 22Hz - 30kHz 2 3005 10 20 50 100 200 0.005 0.01 0.02 0.05 0.1 0.2 0.5 Output Power (W) THD+N (%) THD+N vs Output Power f = 1kHz RL= 8Ω BRIDGED AES 17 Filter +/-23V +/-27V +/-30V 0.005 0.01 0.02 0.05 0.1 0.2 0.5 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) THD+N (%) THD+N vs Frequency Po = 100W/ch RL = 8Ω BRIDGED VPP=|VNN|=30V BW = 22kHz BW = 30kHz -120 -70 -115 -110 -105 -100 -95 -90 -85 -80 -75 20 20k50 100 200 500 1k 2k 5k 10k Frequency (Hz) Amplitude (dBr) Noise Floor VPP=|VNN|=30V RF=RI=20kΩ 8Ω BRIDGED 16kFFT AES 17 Filter 100 0 20 40 60 80 100 120 140 160 180 200 220 240 Output Power (W) Efficiency (%) VPP=|VNN|=30V RL = 8Ω BRIDGED AES 17 Filter THD+N < 10% Efficiency vs Output Power 0.005 0.01 0.02 0.05 0.1 0.2 0.5 2 3005 10 20 50 100 2001 Output Power (W) THD+N (%) f = 1kHz VPP=|VNN|=30V RL= 8Ω BRIDGED AES 17 Filter THD+N vs Output Power

Tripath Technology, Inc. - Technical Information 13 TA2022 – KLI/1.2/07-04

APPLICATION INFORMATION

TA2022 Basic Amplifier Operation The TA2022 has three major operational blocks: the signal processor, the MOSFET driver, and the power MOSFETs. The signal processor is a 5V CMOS block that amplifies the audio input signal and converts the audio signal to a switching pattern. Th is switching pattern is spread spectrum with a typical idle switching frequency of about 650kHz. The switching patterns for the two channels are not synchronized and the idle switching frequencies should differ by at least 40kHz to avoid increasing the audio band noise floor. The idle frequency difference can be accomplished by offsetting the value of CFB for each channel. Typical values of CFB are 390pF for channel 1 and 560pF for channel 2. The MOSFET driver level-shifts the signal proce ssor’s 5V switching patterns to the power supply voltages and drives the power MOSFETs. The MO SFET driver includes a switching power supply integrated to generate the VN10 supply. The VN10 supply powers the low side gate drivers as well provides the charging current need for the “boot strapped” supplies (VBOOT1 and VBOOT2) that power the high side MOSFET drivers. VN10 must be stable (regulated) at 10V to 12V above VNN. The VN10 circuitry shown in the Application / Test Circuit typically produces 11V above VNN. The power MOSFETs are N-channel devices configured in half-bridges and are used to supply power to the output load. The outputs of the power MOSFETs (OUT1 and OUT2) must be low pass filtered to remove the high frequency switching pattern. A residual voltage from the switching pattern will remain on the speaker outputs when the recommended out put LC filter is used, but this signal is outside of the audio band and will not affect audio performance. Circuit Board Layout The TA2022 is a power (high current) amplifier that operates at relatively high switching frequencies. The output of the amplifier sw itches between VPP and VNN at high speeds while driving large currents. This high-frequency digital signal is pa ssed through an LC low-pass filter to recover the amplified audio signal. Since the amplifier must drive the inductive LC output filter and speaker loads, the amplifier outputs can be pulled above the suppl y voltage and below ground by the energy in the output inductance. To avoid subjecting the TA2022 to potentially damaging voltage stress, it is critical to have a good printed circuit board layout. It is recommended that Tripath’s layout and application circuit be used for all applications and only be deviated from after careful analysis of the effects of any changes. Please refer to the TA2022 evaluati on board document, EB-TA2022, available on the Tripath website, at www.tripath.com. The following components are important to place near their associated TA2022 pins and are ranked in order of layout importance, either for proper device operation or performance considerations. - The capacitors C HBR provide high frequency bypassing of the amplifier power supplies and will serve to reduce spikes across the supply rails. C HBR should be kept within 1/8” (3mm) of the VNN(8,9) and VPP(4,12) pins. Pleas e note that both VNN1 and VPP1 as well as VNN2 and VPP2 must be decoupled separately. In addition, the voltage rating for C HBR should be 100V as this capacitor is exposed to the full supply range, VPP-VNN. - DO, fast recovery PN junction diodes minimize undershoots of the outputs with respect to power ground during switching transitions and abnormal load conditions such as output shorts to ground. For maximum effectiveness, these diodes must be located close to the output pins and returned to their respective V NN1(2). Please see Application/Test Circuit for ground return pin. - C FB removes very high frequency components from the amplifier feedback signals and lowers the output switching frequency by delay ing the feedback signals. In addition, the value of C FB is different for channel 1 and channel 2 to keep the average switching frequency difference greater than 40kHz. This minimizes in-band audio noise.

Tripath Technology, Inc. - Technical Information 14 TA2022 – KLI/1.2/07-04 - To minimize noise pickup and minimize THD+N, R FBC should be located as close to the TA2022 as possible. Make sure that the rout ing of the high voltage feedback lines is kept far away from the input op amps or significant noise coupling may occur. It is best to shield the high voltage feedback lines by using a gr ound plane around these traces as well as the input section. - CB provides high frequency bypassing for the boot strap supplies. Very high currents are present on these supplies. - CSW provides high frequency bypassing for the VN 10 generator circuit. Very high currents are present on these supplies. - CSWFB filters the feedback signal (VN10FDBK) for the hysteretic VN10 buck converter. The feedback signal is noise sensitive and the trace from CSWFB to VNN should be kept short. - DSW is the flywheel diode for the VN10 buck c onverter and prevents VN10SW(pin 5) from going more than one diode drop below VNN. In general, to enable placement as close to the TA 2022, and minimize PCB parasitics, the capacitors listed above should be surface mount types, located on the “solder” side of the board. Some components are not sensitive to location but are very sensitive to layout and trace routing. - To maximize the damping factor and reduce distortion and noise, the modulator feedback connections should be routed directly to the pins of the output inductors, L O. This was done on the EB-TA2022 board. - The output filter capacitor, C O, and zobel capacitor, C Z, should be star connected with the load return. The output ground feedback signal should be taken from this star point. This is suggested by the routing on the Application/Test schematic, but, for space/layout reasons, this was not fully implemented on the EB-2022. - The modulator feedback resistors, R FBA and R FBB should all be grounded and attached to 5V together. These connections will serve to minimize common mode noise via the differential feedback. Please refer to the EB-TA2022 evaluation board for more information. TA2022 Grounding Proper grounding techniques are required to ma ximize TA2022 functionality and performance. Parametric parameters such as THD+N, Noise Floor and crosstalk can be adversely affected if proper grounding techniques are not implemented on the PC B layout. The following discussion highlights some recommendations about grounding both with re spect to the TA2022 as well as general “audio system” design rules. The TA2022 is divided into two sections: the input section, which spans pin 15 through pin 32, and the output (high power) section, which spans pin 1 through pin 14. On the TA2022 evaluation board, the ground is also divided into distinct sections, one for the input and one for the output. To minimize ground loops and keep the audio noise floor as low as possible, the input and output ground must be only connected at a single point. Depending on the system design, the single point connection may be in the form of a ferrite bead or a PCB trace. The analog grounds, pin 15 and pin 20 must be connec ted locally at the TA2022 for proper device functionality. On the TA2022 evaluation board, Tripath has used an analog ground plane to minimize the impedances between pin 15 and pin 20 as well as the other analog ground connections, such as V5 supply bypassing, and feedback divider networks. The ground for the V5 power supply should connect directly to pin 20. Additionally, any external i nput circuitry such as preamps, or active filters, should be referenced to pin 20.

Tripath Technology, Inc. - Technical Information 15 TA2022 – KLI/1.2/07-04 For the power section, Tripath has traditionally used a “star” grounding scheme. Thus, the load ground returns and the power supply decoupling trac es are routed separately back to the power supply. In addition, any type of shield or c hassis connection would be connected directly to the ground star located at the power supply. Thes e precautions will both minimize audible noise and enhance the crosstalk performance of the TA2022. The TA2022 incorporates a differential feedback sy stem to minimize the effects of ground bounce and cancel out common mode ground noise. As su ch, the feedback from the output ground for each channel needs to be properly sensed. This c an be accomplished by connecting the output ground “sensing” trace directly to the star formed by the output ground return, output capacitor, C O, and the zobel capacitor, CZ. Refer to the Application / Test Circuit for a schematic description. Pin 3, VN10GND, is used for the VN10 buck converte r. Pin 3 can be connected to the main power supply decoupling ground trace (or plane) without any loss in functionality or reduction of performance. This pin is electric ally shorted to the copper heat sink (case) of the TA2022. Even if the internal VN10 regulator is not being used, VN10GND should still be connected to PGND. TA2022 Amplifier Gain The gain of the TA2022 is the product of the input stage gain and the modulator gain. Please refer to the sections, Input Stage Design, and Modulator Feedback Design, for a complete explanation of how to determine the external component values. MODULATORV EVINPUTSTAG VTA2022 A* AA =  ++−≈ 1R*R )R(R*R R RA FBBFBA FBBFBAFBC I F VTA2022 For example, using a TA2022 with the following external components, R I = 20kΩ RF = 20kΩ RFBA = 1kΩ RFBB = 1.13kΩ R FBC = 9.09kΩ V V18.13 11.13k Ω*1.0k Ω )1.13k(1.0k Ω*9.09k Ω 20k Ω 20k ΩAVTA2022 =  +Ω+−≈ Input Stage Design The TA2022 input stage is configured as an inverting amplifier, allowing the system designer flexibility in setting the input stage gain and frequency response. Figure 1 shows a typical application where the input stage is a constant gain inverting amplifier. The input stage gain should be set so that the maximum input signal level will drive the input stage output to 4Vpp.

Tripath Technology, Inc. - Technical Information 17 TA2022 – KLI/1.2/07-04 OUT 1 GROUND RFBB FBKGND1 RFBA OUT1 RFBC RFBCFBKOUT1 RFBB RFBA AGND 1/2 TA2022 Processing Modulation Figure 2: Modulator Feedback The modulator feedback resistors are: Ω= 1Ktypically specified, UserRFBA 4)-(VPP VPP*RR FBA FBB = VPP*RR FBA FBC = 1R*R )R(R*RA FBBFBA FBBFBAFBC MODULATOR-V ++≈ The above equations assume that VPP=|VNN|. For example, in a system with VPP MAX=36V and VNNMAX=-36V, R FBA = 1kΩ, 1% R FBB = 1.125kΩ, use 1.13kΩ, 1% R FBC = 9.0kΩ, use 9.09kΩ, 1% The resultant modulator gain is: 18.13V/V 11.13k Ω*1.0k Ω )1.13k(1.0k Ω*9.09k ΩA MODULATOR-V =+Ω+≈

Tripath Technology, Inc. - Technical Information 18 TA2022 – KLI/1.2/07-04 Mute The mute pin must be driven to a logic low or logi c high state for proper operat ion. The state of the mute pin is “latched in” to minimize the effects of noise on this pin, which could cause the TA2022 to switch state unintentionally. Controlling the mute pin with a push-pull output from a microcontroller, or a physical switch between V5 and AGND, works well as both solutions have low impedance drive capability. In some cases, it may be desirable to drive the mute pin with an alternative approach. When the device is in mute, the pin must be “pull ed low” via approximately 1kohm to overcome the internal latch and change the TA2022 state (i.e. out of mute). When the device is not in mute, the mute pin must be “pulled high” via approximately 2kohm to overcome the internal latch and change the TA2022 state (i.e. into mute). Figure 3 shows a si mple control circuit that buffers a Mute Control signal that is not capable of driving the Mute pin of the TA2022 directly. When the Mute Control signal is high, the Mute pin will be driven low and the TA 2022 will be on. If the Mute Control signal is low, the 2k resistor will pull the Mute pin high and the TA2022 will be muted. Figure 3: Low impedance drive for Mute Pin To ensure proper device operation, including minimizati on of turn on/off transients that can result in undesirable audio artifacts, Tripath recommends t hat the TA2022 device be muted prior to power up or power down of the 5V supply. The “sensing” of the V5 supply can be easily accomplished by using a “microcontroller supervisor” or equivalent to drive the TA2022 mute pin high when the V5 voltage is below 4.5V. This will ensure proper operation of the TA2022 input circuitry. A micro-controller supervisor such as the MCP101-450 from Microc hip Corporation has been used by Tripath to implement clean power up/down operation. If turn-on and/or turn-off noise is still present with a TA2022 amplifier, the cause may be other circuitry external to the TA2022. While the TA2022 has circ uitry to suppress turn-on and turn-off transients, the combination of power supply and other audio circ uitry with the TA2022 in a particular application may exhibit audible transients. One solution that will completely eliminate turn-on and turn-off pops and clicks is to use a relay to connect/disconnect that amplifier from the speakers with the appropriate timing during power on/off. TA2022 Output Capability The TA2022 can output two channels at 100 watts eac h into a 4ohm load at 1% THD+N. The maximum amplifier output power is determined by a number of factors including the TA2022 junction temperature, the load impedance and the power supply voltage. Tripath does not recommend driving loads below 4 ohm single ended as the amplifier efficiency will be seriously reduced and the amplifier may prematurely current limit. MUTE AGND TO MUTE (Pin 24) CONTROL 10k

Tripath Technology, Inc. - Technical Information 19 TA2022 – KLI/1.2/07-04 Bridging the TA2022 The TA2022 can be bridged by returning the signal from OAOUT1 to the input resistor at INV2. OUT1 will then be a gained version of OAOUT1, and OUT2 will be a gained and inverted version of OAOUT1 (see Figure 3). When the two amplif ier outputs are bridged, the apparent load impedance seen by each output is halved, so the mini mum recommended impedance for bridged operation is 8 ohms. Due to the internal current limit setting, the maximum supply voltage recommended for bridged operation is +/-30V. Bridged operation into loads below 8ohms is possible, but, as mentioned above, the amplifier efficiency will be reduced and the amplifier may prematurely current limit. The TA2022 is capable of 150W into 8 ohms bridged at 0.1% THD+N. AGND OAOUT2 OAOUT1 AGND BIASCAP RI 20k CI INV1 RF TA2022 INPUT INV2 20k Figure 4: Input Stage Setup for Bridging The switching outputs, OUT1 and OUT2, are not synchronized, so a common inductor may not be used with a bridged TA2022. For this same reas on, individual zobel networks must be applied to each output to load each output and lower the Q of each common mode differential LC filter. Output Voltage Offset The output offset voltage of the TA2022 is largely determined by the matching of the respective R FBA, RFBB, and R FBC networks for FBKOUT1(FBKOUT2) and FBKGND1 (FBKOUT2). Thus, the intrinsic offset of the TA2022 can be altered by the external feedback network resistor matching. To minimize the nominal untrimmed offset voltage, 1% tolerance resistors are recommended. In most applications, the output offset voltage will need to be trimmed via an external circuit (either passive or active). The output offset voltage of the TA2022 can be nulled by modifying the modulator feedback as shown in Figure 4. Potentiometer R OFA is used to trim the effective resistance seen by the output ground, and therefore the output offset. ROFB limits the trim range.

Tripath Technology, Inc. - Technical Information 20 TA2022 – KLI/1.2/07-04 Figure 5: Manual Output Offset Trim Circuit A DC servo can also be used to automatically nu ll any offset voltage. The TA2022 evaluation board incorporates a DC servo. Please refer to the TA2022 evaluation board document, EB-TA2022, available on the Tripath website, at www.tripath.com. Current Protection Design Although the over-current (Ioc) trip point is inter nally fixed, there are ex ternal components that can affect output current levels during a short circui t event. Referring to the Application/Test diagram these include the output inductor (Lo), output diodes (Do) and supply bypassing (Cs). The two output inductors, Lo, directly impact t he peak output current levels reached during short circuit events. For this reason they must be rat ed at least 10A regardless of the systems maximum operating current, ensuring the inductor does not sa turate which could lead to switching currents passing unimpeded to the load. When this occurs, the Ioc may be forced to trip on average current levels instead of peak current levels, directly impacting the current levels seen by the load and potentially causing damage to the TA2022. Please refer to the Output Filter Design section for more details on the inductor requirements and figure 6 fo r comparison of load currents with an unsaturated and saturated inductor. The four output diodes, Do, minimize overshoots and undershoots with respect to the supply rails VPP and VNN. In order for these diodes to work properly they must have a low Forward Voltage rating at 10A. We recommend a Fast Recovery diode or Ultra-Fast PN Rectifier diode with a Vf rating of 1V at 10A, or better. Also, These should be pl aced close to their respective output pins to maximum effectiveness. The two bulk supply voltage capacitors, Cs, will absorb the current generated when the diodes (Do) conduct. We recommend that a high frequency capac itor designed for low impedance and high ripple currents be used. Tripath recommends an impedance of .1 Ω or better at 100kHz and a ripple current rating of 1Arms at 100kHz. For maximum effectiveness, Tripath recommends that these capacitors be placed close to their respective supply pins. RFBC 10K Processing RFBB OUT1 OUT 1 GROUND RFBB RFBC FBKOUT1 AGND RFBA ROFB ROFA 50K 1/2 TA2022 RFBA FBKGND1 Modulation

Tripath Technology, Inc. - Technical Information 21 TA2022 – KLI/1.2/07-04 Figure 6: Short circuit load current with unsaturated toroidal inductor and saturated bobbin Inductor (shielded) Output Filter Design Tripath amplifiers generally have a higher swit ching frequency than PWM implementations allowing the use of higher cutoff frequency filters, reduc ing the load dependent peaking/drooping in the 20kHz audio band. This is especially important for applic ations where the end customer may attach any speaker to the amplifier (as opposed to a system where speakers are shipped with the amplifier), since speakers are not purely resistive loads and the impedance they present changes over frequency and from speaker model to speaker model. An RC network, or “zobel” (R Z, CZ) must be placed at the filter output to control the im pedance “seen” by the TA2022. The TA2022 works well with a 2 nd order, 107kHz LC filter with L O = 10uH and C O = 0.22uF and R Z = 6.2ohm/2W and C Z = 0.22uF. Some applications may require a more aggre ssive filter to reduce out of band noise. Below are some proven filter combinations: - 49.5kHz 2 nd order filter - 33.6kHz 2 nd order filter L O = 22uH L O = 33uH C O = .47uF C O = .68uF R Z = 8Ω R Z = 6.2Ω C Z = .47uF C Z = .68uF - 65kHz 4th order filter LO1 = 15uH C O1 = 1uF L O2 = 10uH C O2 = .22uF R Z = 10Ω C Z = .47uF Output inductor selection is a crit ical design step. The core materi al and geometry of the output filter inductor affects the TA2022 distortion levels, efficiency, over-current protection, power dissipation and EMI output. The inductor should have low loss at 700kHz with 80Vpp. It should be reiterated that regardless of the systems maximu m operating current, a 10A rating is required to ensure that peak current conditions will not cause the inductor to sa turate. During a short circuit event the inductor current increases very quickly in a saturated core (see figure 6), compromising the current protection scheme. A 10A rating is sufficient to ensure that current increases through the inductor are linear, and provides a safety margin for the TA2022. There are two types of inductors available in the 10A range that offers some EMI containment: they are the toroidal type and the bobbin (shielded) type inductor. In bobbin construction, a ferrite shield is placed around the core of a bobbin inductor to help contain radiated emissions. This shield can reduce the amount of energy the inductor can store in the core by reducing the air gap, which can lower the peak curr ent capability of the inductor. Typically, a 7-10A shielded bobbin inductor will not have the peak current capability necessary to ensure that the core

Tripath Technology, Inc. - Technical Information 22 TA2022 – KLI/1.2/07-04 will not saturate during short circuit events; this is why they are not recommended for use with the TA2022. Also it should be noted that shielded bobbin c onstruction is not as effective as toroidal construction for EMI containment. Tripath recommends that the customer use a toro idal inductor with a Carbonyl-E core for all applications of the TA2022. This core has a high peak current capability due to its low- µ Carbonyl-E metal powder. A distributed air gap increases its’ energy storage capability, which allows for a small footprint and high current capability. Carbonyl-E toroidal iron powder cores have low loss and good linearity. The toroidal shape is ideal for EMI containment. Also, EMI can be further contained by sizing the toroid to accept a full layer of windings. This aids in shielding the electric field. Tripath recommends: - Micrometals (www.micrometals.com) Type-2 (Carbonyl-E) toroidal iron powder cores. The specific core Tripath initially verifi ed and used on the EB-TA2022 was a T94-2 (23.9mm outer diameter) wound to 11uH with 19AWG wi re. Since then Tripath has determined that much smaller Carbonyl-E toroids will not satu rate during high current events. Tripath has also used T68-2 (17.5mm outer diameter) and the T60-2B/60(15.2mm outer diameter) cores wound to 11uH with 22AWG with good success. If a smaller core is required, core outer diameters as small as 15.2mm (T60-2) work well, but core temperature effects should be tested. The T60-2 core did not saturate dur ing short circuit testing, but maximum core temperatures must be considered and multip le layer winding must be used to achieve 11uH. Multiple winding can increase winding capacitance, which may cause ringing and increased radiated emissions. Bank winding tec hniques can minimize this effect. It should be noted that at core temperatures above 130C the single build wire used by most inductor manufacturers should be replaced with a heavy build wire. Micrometals does not provide winding services, but many companies purchase directly from them and provide completely finished inductors. Pulse Engineering has assigned a part number for the T68-2 wound with 44 turns of 22 AWG single build wire. The part number is PA0291. - Amidon Inc./American Cores type-06 (Carbonyl-E) toroidal iron powder cores. Tripath has used T690-06 (17.5mm outer diameter) cores wound to 11uH with good success. Amidon carries type-06 cores in the 23.9mm to 15.2 mm outer diameter range. They have assigned a part number for the T690-06 wound with 44 turns of 22 AWG single build wire. This part is approved by Tripath and is 690064422. Power Supplies The TA2022 requires the split supply rails VPP1(VPP2) and VNN1(VNN2), and V5. It also uses some additional voltages, VN10, VBOOT1 and VBOOT2 that are generated internally. The selection of components for the switching regulator is shown in the Application / Test Diagram. Minimum and Maximum Supply Voltage Operating Range The TA2022 can operate over a wide range of power supply voltages from +/-12V to +/-36V. In order to optimize operation for either the low or high range, the user must select the proper values for RVNNSENSE, RVPPSENSE, RFBA, RFBB, and RFBC. Please refer to the Modulator Feedback Design and Over/Under-voltage Protection sections for more additional information. VN10 Supply The TA2022 has an internal hysteretic buck converte r, which, in conjunction with a few passive components, generates the necessary floating power supply for the MOSFET driver stage (nominally 11V with the external components shown in Applic ation / Test Circuit). The performance curves shown in the data sheet as well as efficiency measurements were done using the internal VN10 generator. Tripath recommends that the internal VN10 generator be used. In some cases, though, a designer may wish to use an external VN10 generator. The specification for VN10 quiescent current (65mA typical, 80mA maxi mum) in the Electrical Characteristics section states the amount of current needed when an external floating supply is used. If the internal VN10

Tripath Technology, Inc. - Technical Information 24 TA2022 – KLI/1.2/07-04 Protection Circuits The TA2022 is guarded against over-current, over / under-voltage and over-temperature conditions. If the device goes into an over-current or over / under-voltage condition, the HMUTE goes to a logic HIGH indicating a fault condition. When this occurs, the amplifier is muted, all outputs are TRI- STATED, and will float to approximately 2.5VDC. Over-Current Protection An over-current fault occurs if more than approxim ately 8 amps of current flows from any of the amplifier output pins. This can occur if the speak er wires are shorted toget her or if one side of the speaker is shorted to ground. An over-current faul t sets an internal latch that can only be cleared if the MUTE pin is toggled or if the part is powered dow n. See the over-current curves in the Typical Characteristics section for more information. Over/Under Voltage Protection The TA2022 has built-in over and under voltage prot ection for both the VPP and VNN supply rails. The nominal operating voltage will typically be chosen as the supply “center point.” This allows the supply voltage to fluctuate, both above and below, the nominal supply voltage. VPPSENSE (pin 19) performs the over and undervolt age sensing for the positive supply, VPP. VNNSENSE (pin 18) performs the same function for the negative rail, V NN. In the simplest implementation, the supply is done via a single, exte rnal resistor per sense pin. This scheme is referred to as the “single resistor” sense circuit. Figure 9 shows the single resistor sense circuit. Figure 9: Single Resistor Sense Circuit When the current through R VPPSENSE (or R VNNSENSE) goes below or above the values shown in the muted. VPPSENSE is internally biased at 2.5V and VNNSENSE is biased at 1.25V. For the “single resistor” sense case (as shown in the Application / Test Diagram), these bias points must be taken into consideration when calculating the RVPPSENSE or RVNNSENSE resistor. Once the supply comes back into the supply voltage operating range (as defined by the supply sense resistors), the TA2022 will automatic ally be unmuted and will begin to amplify. There is a hysteresis range on both the VPPSENSE and VNNSENSE pins. If the amplifier is powered up in the hysteresis band the TA2022 will be muted. Thus, the usable s upply range is the difference between the over- voltage turn-off and under-voltage turn-off for both the VPP and VNN supplies. It should be noted that there is a timer of approximately 200mS with res pect to the over and under voltage sensing circuit. VPP TA2022 VNN RVPPSENSE VNNSENSE VPPSENSE RVNNSENSE

Tripath Technology, Inc. - Technical Information 25 TA2022 – KLI/1.2/07-04 Thus, the supply voltage must be outside of the user defined supply range for greater than 200mS for the TA2022 to be muted. The equation for calculating R VPPSENSE is as follows: VPPSENSE VPPSENSE I 2.5V - VPPR = The equation for calculating RVNNSENSE is as follows: VNNSENSE VNNSENSE I VNN-1.25V R = where IVPPSENSE or IVNNSENSE can be any of the currents shown in the Electrical Characteristics table for VPPSENSE and VNNSENSE, respectively. Example: Nominal supply voltage – +/-32.5V +/-10% From this information, a value of R VPPSENSE and R VNNSENSE can be calculated using the above formulas. 36V use35.75V1.132.5VVPP MAX =×= 36V- use35.75V1.132.5VVNN MAX −=×−= %1,k243use242.75A138 2.5V -36V R VPPSENSE Ω== kΩµ where IVPPSENSE is the minimum over-voltage turn off current for VPPSENSE. %1,k249use245.1A152 36V- -1.25V R VNNSENSE Ω== kΩµ where IVNNSENSE is the minimum over-voltage turn off current for VNNSENSE. Using the resistor values from above, the actual minimum over voltage turn off points will be: 36.03V2.5V138kΩ243VPP N_OFFMIN_OV_TUR =+×= µA 36.60V-A152kΩ4921.25VVNN N_OFFMIN_OV_TUR =×−=µ The other three trip points can be calculated usi ng the same formula but inserting the appropriate IVPPSENSE (or I VNNSENSE) current value. As stated earlier, the usable supply range is the difference between the minimum overvoltage turn off and maximum under voltage turn-off for both the VPP and VNN supplies. N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VPP-VPPVPP = N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VNN-VNNVNN = Using the resistor values from above, and the maximum under voltage trip currents shown in the Electrical Characteristics table, the maximum under voltage turn off points will be: 23.64V2.5V87kΩ243VPP N_OFFMAX_UV_TUR =+×= Aµ 22.41V-A95kΩ4921.25VVNN N_OFFMAX_UV_TUR =×−=µ

Tripath Technology, Inc. - Technical Information 26 TA2022 – KLI/1.2/07-04 and the resultant supply ranges will be: 12.39V23.64V03.36VPP RANGE =−= 14.19V-22.41V60.36VNN RANGE =−−−= It should also be noted that the tolerance of the R VPPSENSE (or R VNNSENSE) resistors will effect the trip voltages and thus, the usable supply range. To minimize the additional variance Tripath recommends 1% tolerance resistors. As a matter of completeness, the formulas below in clude the effect of resistor tolerance assuming a known value of R VPPSENSE or RVNNSENSE. 2.5V)100/TOL1()IR(VPP TRIPVPPSENSE N_OFFMIN_OV_TUR ++÷×= 2.5V)100/TOL1()IR(VPP TRIPVPPSENSE N_OFFMAX_UV_TUR ++××= )100/TOL1()IR(25.1VNN TRIPVNNSENSE N_OFFMIN_OV_TUR +÷×−= )100/TOL1()IR(25.1VNN TRIPVNNSENSE N_OFFMAX_UV_TUR +××−= Using a value of 243k Ω for R VPPSENSE and a value of 249k Ω for R VNNSENSE, assuming 5% tolerance, along with the appropriate value of ITRIP, the trip voltages and supply ranges can be calculated. 34.44V 2.5V)100/51()138243k Ω(VPP N_OFFMIN_OV_TUR =++÷×= Aµ 24.70V 2.5V)100/51()µA87243k Ω(VPP N_OFFMAX_UV_TUR =++××= 9.74V24.70V44.34VPP RANGE =−= 34.80V)100/51()µA152kΩ492(25.1VNN N_OFFMIN_OV_TUR −=+÷×−= 23.59V)100/51()µA95kΩ492(25.1VNN N_OFFMAX_UV_TUR −=+××−= 11.21V-23.59V80.34VNN RANGE =−−−= Thus, by using 5% resistors, the supply range for the VPP has been reduced by 2.65V while the VNN range has been reduced by approximately 3.0V (as compared to resistors with no tolerance variation). In actuality, if a 5% resistor was to be used, then the initial value of R VPPSENSE and RVNNSENSE would have had to be adjusted such that the minimum over voltage turn off points would have never been less than +/-36V as defined by the supply voltage and tolerance specification. It should be noted that the values for V VPPSENSE and VVNNSENSE shown in the Electrical Characteristics table were calculated using a value of 249k Ω for both R VPPSENSE and R VNNSENSE. In addition, for the maximum and minimum values, as opposed to the typical ones, a 1% tolerance resistor value around 249kΩ was chosen to show the effect on supply r ange. Thus, the minimum and maximum values would be “worst case” assuming a supply voltage of 5V for the input section of the TA2022. The entire discussion thus far has been for the “one resistor” sense circuit. This configuration requires a single resistor from either VPPSENSE or VNNSENSE to the respec tive power supply. While the simplest configuration, in terms of ex ternal components, there are some drawbacks to this configuration. The first drawback is that the range for VPP RANGE and VNNRANGE are asymmetric due to the different internal bias voltages of VPPSEN SE and VNNSENSE. A second issue is that current through RVPPSENSE or RVNNSENSE will change if the V5 voltage is not exactly 5V, since the bias voltages of pin 18 and pin 19 are set by resistor dividers between V5 and AGND. With an additional resistor per supply sense pi n (2 resistors per VPPSENSE or VNNSENSE), the drawbacks of the “one resistor” sense circuit can be e liminated. In addition, the calculations of the sense resistors are actually more straightforward for the “two resistor” sense circuit as opposed to the “one resistor” scheme. Figure 10 shows the proper connection for the “two resistor” sense circuit for both the VPPSENSE and VNNSENSE pins.

Tripath Technology, Inc. - Technical Information 27 TA2022 – KLI/1.2/07-04 Figure 10: Two Resistor Sense Circuit The equation for calculating R VPP1 is as follows: VPPSENSE VPP1 I VPPR = Set VPP1 VPP2 RR = . The equation for calculating R VNNSENSE is as follows: VNNSENSE VNN1 I VNNR = Set VNN1 VNN2 R3R ×= . IVPPSENSE or IVNNSENSE can be any of the currents shown in the Electrical Characteristics table for VPPSENSE and VNNSENSE, respectively. Example: Nominal supply voltage – +/-32.5V +/-10% From this information, a value of R VPP1, RVPP2, RVNN1, and R VNN2 can be calculated using the above formulas. 36V use35.75V1.132.5VVPP MAX =×= 36V- use35.75V1.132.5VVNN MAX −=×−= %1,k261use260.87A138 36VR VPP1 Ω== kΩµ Set %1,261 RVPP2 Ω= k . where IVPPSENSE is the minimum over-voltage turn off current for VPPSENSE. %1,k237use236.84A152 36VR VNN2 Ω== kΩµ Set %1,kΩ715 RVNN2 = . where IVNNSENSE is the minimum over-voltage turn off current for VNNSENSE. RVNN2 TA2022 RVNN1 RVPP1 VPP

Tripath Technology, Inc. - Technical Information 28 TA2022 – KLI/1.2/07-04 The two additional resistors, R VPP2 and R VNN2 compensate for the internal bias points. Thus, R VPP1 and R VNN1 can be used for the direct calculation of the actual VPP and VNN trip voltages without considering the effect of RVPP2 and RVNN2. Using the resistor values from above, the actual minimum over voltage turn off points will be: 36.02V138kΩ261VPP N_OFFMIN_OV_TUR =×= Aµ 36.24V-A152kΩ372VNN N_OFFMIN_OV_TUR =×−=µ The other three trip points can be calculated usi ng the same formula but inserting the appropriate IVPPSENSE (or I VNNSENSE) current value. As stated earlier, the usable supply range is the difference between the minimum overvoltage turn off and maximum under voltage turn-off for both the VPP and VNN supplies. N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VPP-VPPVPP = N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VNN-VNNVNN = Using the resistor values from above, and the maximum under voltage trip currents shown in the Electrical Characteristics table, the maximum under voltage turn off points will be: 22.71V87kΩ261VPP N_OFFMAX_UV_TUR =×= Aµ 22..51V-A95kΩ372 VNN N_OFFMAX_UV_TUR =×−=µ and the resultant minimum supply ranges will be: 13.31V22.71V02.36VPP RANGE =−= 13.73V-22.51V24.36VNN RANGE =−−−= By adding a total of 2 additional resistors (1 for VPPSENSE and 1 for VNNSENSE), the minimum supply range is now about 3% different as opposed to 13% for the “single resistor” sense case. In addition, the VPP range has been increased by nearly one volt. This represents a 7% improvement in supply range for VPP. As in the single resistor case, the tolerance of the R VPP1 and RVPP2 (or RVNN1 and RVNN2) resistors will affect the trip voltages and thus, the usable supply range. As the nominal supply voltage is decreased, the effect of R VPP2 and R VNN2 becomes more pronounced. To minimize the additional variance Tripath recommends 1% toleranc e resistors. It is possible to calculate the effect of resistor tolerances for the “two resistor” sense circuit, but ultimately, 1% resistors should be used for the sense circuit in all, but the tightest regulated supply schemes. Over Temperature Protection An over-temperature fault occurs if the junction temperature of the part exceeds approximately 165°C. The thermal hysteresis of the part is approximately 30°C, therefore the fault will automatically clear when the junction temperature drops below 135°C. HMUTE (pin 32) The HMUTE pin is a 5V logic output that indicates various fault conditions within the device. These conditions include: over-current, overvoltage and undervoltage. The HMUTE output is capable of directly driving an LED through a series 2kΩ resistor. Heat Sink Requirements most applications it will be necessary to fasten t he TA2022 to a heat sink. The determining factor is that the 150 °C maximum junction temperature, T J(max) cannot be exceeded, as specified by the following equation:

Tripath Technology, Inc. - Technical Information 29 TA2022 – KLI/1.2/07-04 PDISS = ( ) JA A)MAX(J TT θ where: PDISS = maximum power dissipation TJMAX = maximum junction temperature of TA2022 TA = operating ambient temperature θJA = junction-to-ambient thermal resistance θJA = θJC + θCS + θSA Example: What size heat sink is required to operate t he TA2022 at 80W per channel continuously in a 70ºC ambient temperature? P DISS is determined by: Efficiency = η = IN OUT P P = DISSOUT OUT PP P PDISS (per channel) = W88.519085.0 90 =−=− OUT OUT PP η Thus, P DISS for two channels = 31.76W θJA = ( ) DISS A)MAX(J P TT − = 76.31 70150 − = 2.52°C/W The θJC of the TA2022 is 1.0°C/W, so a heat sink of 1.32°C/W is required for this example (assuming a θCS = 0.2°C/W). In actual applications, other factors such as the average PDISS with a music source (as opposed to a continuous sine wave) and regulatory agency testing requirements will help determine the size of the heat sink required. Performance Measurements of the TA2022 The TA2022 operates by generating a high frequency switching signal based on the audio input. This signal is sent through a low-pass filter (external to the Tripath amplifier) that recovers an amplified version of the audio input. The frequency of the switching pattern is spread spectrum in nature and typically varies between 100kHz and 1MHz, which is well above the 20Hz – 20kHz audio band. The pattern itself does not alter or distort the audio input signal, but it does introduce some inaudible components. The measurements of certain performance parameters, particularly noise related specifications such as THD+N, are significantly affected by the design of the low-pass filter used on the output as well as the bandwidth setting of the measurement instrument used. Unless the filter has a very sharp roll-off just beyond the audio band or the bandwidth of the measurement instrument is limited, some of the inaudible noise components introduced by the TA2022 amplifier switching pattern will degrade the measurement. One feature of the TA2022 is that it does not require large multi-pole filters to achieve excellent performance in listening tests, usually a more critical factor than performance measurements. Though using a multi-pole filter may remove high-frequency noise and improve THD+N type measurements (when they are made with wide-bandwidth measuring equipment), these same filters degrade frequency response. The TA2022 Evaluation Board uses the Application/Test Circuit of this data sheet, which has a simple two-pole output filter and excellent performance in listening tests. Measurements in this data sheet were taken using this same circuit with a limited bandwidth setting in the measurement instrument.

Tripath Technology, Inc. - Technical Information 30 TA2022 – KLI/1.2/07-04

PACKAGE INFORMATION

Tripath Technology, Inc. - Technical Information 31 TA2022 – KLI/1.2/07-04 Tripath Technology, Inc. reserves the right to make changes without further notice to any products herein to improve reliability, function or design. Tripath does not assume any liability arising out of the application of use of any product or circuit described herein; neither does it convey any license under its patent rights nor the rights of others. Tripath and Digital Power Processing are trademarks of Tripath Technology. Other trademarks referenced in this document are owned by their respective companies. TRIPATH’S PRODUCT ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN CONSENT OF THE PRESIDENT OF TRIPATH TECHONOLOGY, INC. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used in accordance with instructions for use provided in this labeling, can be reasonably expected to result in significant injury of the user. 2. A critical component is any co mponent of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. Contact Information TRIPATH TECHNOLOGY, INC

2560 Orchard Parkway, San Jose, CA 95131

408.750.3000 - P 408.750.3001 - F For more Sales Information, please visit us @ www.tripath.com/cont_s.htm For more Technical Information, please visit us @ www.tripath.com/data.htm

Summary

STEREO 90W (4?? CLASS-T??DIGITAL AUDIO AMPLIFIER DRIVER USING DIGITAL POWER PROCESSING (DPP?? TECHNOLOGY

Description