RT6212A RICHTEK | Alldatasheet
Document overview
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Technical content
Features
Integrated 163m/86m MOSFETs 4.5V to 18V Supply Voltage Range 500kHz Switching Frequency ACOT Control Feedback Reference Voltage 0.8V ± 1.5% Feedback Reference Voltage 0.765V ± 1.5% Internal Start-Up into Pre-Biased Outputs Compact Package : TSOT-23-6 Pin Input Under-Voltage Lockout Over-Current Protection and Hiccup
Applications
Set-Top Boxes Portable TVs Access Point Routers DSL Modems LCD TVs Marking Information 2W= : Product Code DNN : Date Code RT6212AHRGJ6F 2W=DNN 2V= : Product Code DNN : Date Code RT6212BHRGJ6F 2V=DNN 0B= : Product Code DNN : Date Code RT6212AHGJ6F 0B=DNN 0A= : Product Code DNN : Date Code RT6212BHGJ6F 0A=DNN Simplified Application Circuit EN RT6212A/B GND VINVIN CIN BOOT L CBOOT LX Enable FB COUT VOUT CFF
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. www.richtek.com DS6212A/B-05 July 2017 Pin Configuration (TOP VIEW) GND LX VIN BOOT EN FB 2 3 TSOT-23-6 (FC) Functional Pin Description Pin No. Pin Name Pin Function 1 GND System ground. Provides the ground return path for the control circuitry and low-side power MOSFET. 2 LX Switch node. LX is the switching node that supplies power to the output and connect the output LC filter from LX to the output load. 3 VIN Power input. Supplies the power switches of the device. 4 FB Feedback voltage input. This pin is used to set the desired output voltage via an external resistive divider. The feedback voltage is 0.765V/0.8V typically. 5 EN Enable control input. Floating this pin or connecting this pin to GND can disable the device and connecting this pin to logic high can enable the device. 6 BOOT Bootstrap supply for high -side gate driv er. Connect a 100nF or greater capacitor from LX to BOOT to power the high-side switch. Functional Block Diagram UGATE LGATE Driver LX BOOT PVCC Control On-Time EN Comparator LX GND Reg VIBIAS VREF PVCC LX VCC Ripple Gen. VIN FB Minoff EN VIN GND LX VIN UV &OV OC VCC
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. DS6212A/B-05 July 2017 www.richtek.com Operation The RT6212A/B is a synchronous step-down converter with advanced constant on -time control mode. Using the ACOTTM control mode can reduce the output capacitance and provide fast transient response. It can minimize the component size without additional external compensation network. Current Protection The inductor current is monitored via the inte rnal switches cycle-by-cycle. Once the output voltage drops under UV threshold, the RT6212A/B will enter hiccup mode. UVLO Protection To protect the chip from operating at insufficient supply voltage, the UVLO is needed. When the input voltage of VIN is lower than the UVLO falling threshold voltage, the device will be lockout. Thermal Shutdown When the junction temperature exceeds the OTP threshold value, the IC will shut down the switching operation. Once the junction temperature cools down and is lower than the OTP lower threshold, t he converter will autocratically resume switching.
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. www.richtek.com DS6212A/B-05 July 2017 Absolute Maximum Ratings (Note 1) Power Dissipation, PD @ TA = 25C Package Thermal Resistance (Note 2) ESD Susceptibility (Note 3) Recommended Operating Conditions (Note 4)
Electrical Characteristics
(VIN = 12V, TA = 25C, unless otherwise specified) Parameter Symbol Test Conditions Min Typ Max Unit Supply Voltage VIN Supply Input Operating Voltage VIN 4.5 -- 18 V Under-Voltage Lockout Threshold VUVLO RT6212A/BHGJ6F 3.6 3.9 4.2 V RT6212A/BHRGJ6F 3.45 3.75 4.05 Under-Voltage Lockout Threshold Hysteresis VUVLO -- 340 -- mV Shutdown Current ISHDN VEN = 0V -- -- 5 µA Quiescent Current IQ VEN = 2V, VFB = 0.85V -- 0.5 -- mA Soft-Start Soft-Start Time tSS -- 1000 -- µs Enable Voltage Enable Voltage Threshold VEN_R VEN rising, RT6212A/BHGJ6F 1.4 1.5 1.6 V VEN_F VEN falling, RT6212A/BHGJ6F 1.18 1.28 1.38 VEN_R VEN rising, RT6212A/BHRGJ6F 1.33 1.43 1.53
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. DS6212A/B-05 July 2017 www.richtek.com Parameter Symbol Test Conditions Min Typ Max Unit VEN_F VEN falling, RT6212A/BHRGJ6F 1.11 1.21 1.31 Feedback Voltage Feedback Reference Voltage VREF 4.5V ≤ VIN ≤ 18V, RT6212A/BHGJ6F 0.788 0.8 0.812 V 4.5V ≤ VIN ≤ 18V, RT6212A/BHRGJ6F 0.753 0.765 0.777 Internal MOSFET High-Side On-Resistance RDS(ON)_H VBOOT − VLX = 4.8V -- 163 -- mΩ Low-Side On-Resistance RDS(ON)_L -- 86 -- Current Limit High-Side Switch Current Limit ILIM_H -- 5.8 -- A Low-Side Switch Valley Current Limit ILIM_L 2.6 3.3 -- Switching Frequency Switching Frequency fSW 400 500 640 kHz On-Time Timer Control Minimum On-Time tON_MIN 35 60 85 ns Minimum Off-Time tOFF_MIN 185 240 315 Output Under-Voltage and Over-Voltage Protections OVP Trip Threshold OVP detect -- 125 -- % OVP Propagation Delay -- 10 -- µs UVP Trip Threshold UVP detect 45 50 55 Hysteresis -- 10 -- UVP Propagation Delay -- 5 -- µs Thermal Shutdown Thermal Shutdown Threshold TSD -- 150 -- Thermal Shutdown Hysteresis TSD -- 20 -- Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to a bsolute maximum rating conditions may affect device reliability. Note 2. JA is measured under natural convection (still air) at TA = 25 C with the component mounted on a high effective-thermal-conductivity four-layer test board on a JEDEC 51 -7 thermal measurement standard. The first layer is filled with copper. JA is measured at the lead of the package. Note 3. Devices are ESD sensitive. Handling precaution recommended. Note 4. The device is not guaranteed to function outside its operating conditions.
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. Table 1. Recommended Components Selection (RT6212A/BHGJ6F) Table 2. Recommended Components Selection (RT6212A/BHRGJ6F)
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. DS6212A/B-05 July 2017 www.richtek.com Typical Operating Characteristics Efficiency vs. Output Current 100 0 0.5 1 1.5 2 Output Current (A) Efficiency (%) VIN = 4.5V VIN = 12V VIN = 18V RT6212A, VOUT = 1.2V Efficiency vs. Output Current 100 0 0.5 1 1.5 2 Output Current (A) Efficiency (%) VIN = 4.5V VIN = 12V VIN = 18V RT6212B, VOUT = 1.2V Efficiency vs. Output Current 100 0 0.5 1 1.5 2 Output Current (A) Efficiency (%) VOUT = 5V VOUT = 3.3V VOUT = 1.2V RT6212A, VIN = 12V Efficiency vs. Output Current 100 0 0.5 1 1.5 2 Output Current (A) Efficiency (%) RT6212B, VIN = 12V VOUT = 5V VOUT = 3.3V VOUT = 1.2V Output Voltage vs. Output Current 1.16 1.17 1.18 1.19 1.20 1.21 1.22 1.23 1.24 0 0.5 1 1.5 2 Output Current (A) Output Voltage (V) RT6212A RT6212B VIN = 12V, VOUT = 1.2V Output Voltage vs. Input Voltage 1.15 1.16 1.17 1.18 1.19 1.20 1.21 1.22 1.23 4 6 8 10 12 14 16 18 Input Voltage (V) Output Voltage (V) IOUT = 2A IOUT = 1A
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. www.richtek.com DS6212A/B-05 July 2017 Switching Frequency vs. Output Current 300 350 400 450 500 550 600 650 700 0 0.5 1 1.5 2 Output Current (A) Switching Frequency (kHz) 1 RT6212B Switching Frequency vs. Temperature 300 350 400 450 500 550 600 650 700 -50 -25 0 25 50 75 100 125 Temperature (°C) Switching Frequency (kHz) 1 IOUT = 1A Feedback Voltage vs. Input Voltage 0.76 0.77 0.78 0.79 0.80 0.81 0.82 0.83 0.84 4 6 8 10 12 14 16 18 Input Voltage (V) Feedback Voltage (V) RT6212A/BHGJ6F Feedback Voltage vs. Temperature 0.76 0.77 0.78 0.79 0.80 0.81 0.82 0.83 0.84 -50 -25 0 25 50 75 100 125 Temperature (°C) Feedback Voltage (V) RT6212A/BHGJ6F Feedback Voltage vs. Input Voltage 0.725 0.735 0.745 0.755 0.765 0.775 0.785 0.795 0.805 4 6 8 10 12 14 16 18 Input Voltage (V) Feedback Voltage (V) RT6212A/BHRGJ6F Feedback Voltage vs. Temperature 0.725 0.735 0.745 0.755 0.765 0.775 0.785 0.795 0.805 -50 -25 0 25 50 75 100 125 Temperature (°C) Feedback Voltage (V) RT6212A/BHRGJ6F
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. DS6212A/B-05 July 2017 www.richtek.com Inductor Valley Current Limit vs. Input Voltage 1.5 2.0 2.5 3.0 3.5 4.0 4.5 4 6 8 10 12 14 16 18 Input Voltage (V) Inductor Valley Current Limit (A) 1 Low-Side Switch Inductor Valley Current Limit vs. Temperature 1.5 2.0 2.5 3.0 3.5 4.0 4.5 -50 -25 0 25 50 75 100 125 Temperature (°C) Inductor Valley Current Limit (A) 1 Low-Side Switch Shutdown Current vs. Input Voltage 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4 6 8 10 12 14 16 18 Input Voltage (V) Shutdown Current (μA) 1 VEN = 0V Shutdown Current vs. Temperature -50 -25 0 25 50 75 100 125 Temperature (°C) Shutdown Current (μA) 1 VEN = 0V Quiescent Current vs. Input Voltage 500 600 700 800 900 1000 1100 1200 4 6 8 10 12 14 16 18 Input Voltage (V) Quiescent Current (μA) VEN = 2V, VFB = 0.85V Quiescent Current vs. Temperature 500 600 700 800 900 1000 1100 1200 -50 -25 0 25 50 75 100 125 Temperature (°C) Quiescent Current (μA) VEN = 2V, VFB = 0.85V
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. www.richtek.com DS6212A/B-05 July 2017 Input UVLO vs. Temperature 3.4 3.5 3.6 3.7 3.8 3.9 4.0 4.1 -50 -25 0 25 50 75 100 125 Temperature (°C) Input UVLO (V) RT6212A/BHGJ6F Rising Falling Enable Threshold vs. Temperature 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 -50 -25 0 25 50 75 100 125 Temperature (°C) Enable Threshold (V) RT6212A/BHGJ6F Rising Falling Input UVLO vs. Temperature 3.3 3.4 3.5 3.6 3.7 3.8 3.9 4.0 -50 -25 0 25 50 75 100 125 Temperature (°C) Input UVLO (V) Falling Rising RT6212A/BHRGJ6 F Enable Threshold vs. Temperature 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 -50 -25 0 25 50 75 100 125 Temperature (°C) Enable Threshold (V) Rising Falling RT6212A/BHRGJ6 F VIN = 12V, VOUT = 1.2V, IOUT = 0 to 2A Load Transient Response Time (100s/Div) VOUT (100mV/Div) IOUT (1A/Div) RT6212A VIN = 12V, VOUT = 1.2V, IOUT = 0 to 2A Load Transient Response Time (100s/Div) VOUT (100mV/Div) IOUT (1A/Div) RT6212B
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. DS6212A/B-05 July 2017 www.richtek.com VIN = 12V, VOUT = 1.2V, IOUT = 1A Voltage Ripple Time (1s/Div) VLX (10V/Div) VOUT (50mV/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2A Voltage Ripple Time (1s/Div) VLX (10V/Div) VOUT (50mV/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2A Power On from Input Voltage Time (2.5ms/Div) VIN (20V/Div) VLX (10V/Div) VOUT (1V/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2A Power Off from Input Voltage Time (25ms/Div) VIN (20V/Div) VLX (10V/Div) VOUT (1V/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2A Power On from Enable Time (2.5ms/Div) VEN (5V/Div) VLX (10V/Div) VOUT (1V/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2A Power Off from Enable Time (2.5ms/Div) VEN (5V/Div) VLX (10V/Div) VOUT (1V/Div) IOUT (2A/Div)
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. www.richtek.com DS6212A/B-05 July 2017
Application Information
Selecting an inductor involves specifying its inductance and also its required peak current. The exact inductor value is generally flexible and is ultimately chosen to obtain the best mix of cost, physical size, and circuit efficiency. Lower inductor values benefit from reduced size and cost and they can improve the circuit's transient response, but they increase the inductor ripple current and output voltage ripple and reduce the efficiency due to the resulting higher peak currents. Conversely, higher inductor values increase efficiency, but the inductor will either be physically larger or have higher resistance since more turns of wire are required and transient response will be slower since more time is required to change current (up or down) in the inductor. A good compromise between size, efficiency, and transient response is to use a ripple current ( IL) about 30% of the desired full output load current. Calculate the approximate inductor value by selecting the input and output voltages, the switching frequency (fSW), the maximum output current (I OUT(MAX)) and estimating a IL as some percentage of that current. OUT IN OUT IN SW L V V VL = V f I Once an inductor value is chosen, the ripple current (IL) is calculated to determine the required peak inductor current. OUT IN OUT LL L(PEAK) OUT(MAX) IN SW V V V II = and I = IV f L 2 To guarantee the requ ired output current, the inductor needs a saturation current rating and a thermal rating that exceeds IL(PEAK). These are minimum requirements. To maintain control of inductor current in overload and short circuit conditions, some applications may desire current ratings up to the current limit value. However, the IC's output under -voltage shutdown feature make this unnecessary for most applications. IL(PEAK) should not exceed the minimum value of IC's upper current limit level or the IC may not be able to meet the desired output current. If needed, reduce the inductor ripple current ( IL) to increase the average inductor current (and the output current) while ensuring that I L(PEAK) does not exceed the upper current limit level. For best efficiency, choose an inductor with a low DC resistance that meets the cost and size requirements. For low inductor core losses some type of ferrite core is usually best and a shielded core type, although possibly larger or more expensive, will probably give fewer EMI and other noise problems. Considering the T ypical Operating Circuit for 1.2V output at 2A and an input voltage of 12V, using an inductor ripple of 1A (50%), the calculated indu ctance value is : 1.2 12 1.2L 2.16 μH12 500kHz 1A The ripple current was selected at 1A and, as long as we use the calculated 2.16H inductance, that should be the actual ripple current amount. The ripple c urrent and required peak current as below : L 1.2 12 1.2I = = 1A12 500kHz 2.16 μH L(PEAK) 1Aand I = 2A = 2.5A 2 For th e 2.16H value, the inductor's sa turation and thermal rating should exceed at least 2.5A. For more conservative, the rating for inductor saturation current must be equal to or greater than switch current limit of the device rather than the inductor peak current. Input Capacitor Selection The input filter capacitors are needed to smooth out the switched current drawn from the input power source and to reduce voltage ripple on the input. The actual capacitance value is less important than the RMS current rating (and voltage rating, of course). The RMS input ripple current (I RMS) is a function of the input voltage, output voltage, and load current : OUT INRMS OUT(MAX) IN OUT V VI = I 1 VV
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. DS6212A/B-05 July 2017 www.richtek.com Ceramic capacitors are most often used because of their low cost, small size, high RMS current ratings, and robust surge current capabilities. However, take care when these capacitors are used at the input of circuits supplied by a wall adapter or other supply connected through long, thin wires. Current surges through the inductive wires can induce ringing at the RT 6212A/B input which could potentially cause large, damaging voltage spikes at VIN. If this phenomenon is observed, some bulk input capacitance may be required. Ceramic capacitors (to meet the RMS current requirement) can be placed in parallel with other types such as tantalum, electrolytic, or polymer (to reduce ringing and overshoot). Choose capacitors rated at higher temperatures than required. Several ceramic capacitors may be paralleled to meet the RMS current, size, and height requirements of the application . The typical operating circuit uses 22F and 0.1 F low ESR ceramic capacitors on the input. Output Capacitor Selection The RT 6212A/B are optimized for ceramic output capacitors and best performance will be obtained using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on positive load steps) and soar (overshoot on negative load steps). Output Ripple Output ripple at the switching frequency is caused by the inductor current ripple and its effect on the output capacitor's ESR and stored charge. These two ripple components are called ESR ripple and capacitive ripple. Since ceramic capacitors have extremely low ESR and relatively little capacitance , both components are similar in amplitude and both should be considered if ripple is critical. RIPPLE RIPPLE(ESR) RIPPLE(C) RIPPLE(ESR) L ESR LRIPPLE(C) OUT SW V = V V V = I R IV = 8 C f For the Typical Operating Circuit for 1.2V output and an inductor ripple of 1A, with 1 x 22F output capacitance each with about 5m ESR inclu ding PCB trace resistance, the output voltage ripple components are : RIPPLE(ESR) RIPPLE(C) RIPPLE V = 1A 5m = 5mV 1AV = 11.4mV 8 22 μF 500kHz V = 5mV 11.4mV 16.4mV Output Transient Undershoot and Overshoot In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT transient response is very quick and output transients are usually small. However, the combination of small ceramic output capacitors (with little capacitance), low output voltages (with little stored charge in the output capacitors), and low duty cycle applications (which require high inductance to get reasonable ripple currents with high input voltages) increases the size of voltage variations in response to very quick load changes. Typically, load changes occur slowly with respect to the IC's 500kHz switching frequency. But some modern digital loads can exhibit nearly instantaneous load changes and the following section shows how to calculate the worst -case voltage swings in response to very fast load steps. The output voltage transient undershoot and overshoot each have two components : the voltage steps caused by the output capacitor's ESR, and the voltage sag and soar due to the finite output capacitance and the inductor current slew rate. Use the following formulas to check if the ESR is low enough (typically not a problem with ceramic capacitors) and the output capacitance is large enough to prevent excessive sag and soar on very fast load step edges, with the chosen inductor value.
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. www.richtek.com DS6212A/B-05 July 2017 The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor : VESR _STEP = IOUT x RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input -to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on -time and the minimum off -time since the ACOT TM control scheme will ramp the current using on -times spaced apart with minimum off -times, which is as fast as allowed. Calculate the approximate on -time (neglecting parasites) and maximum duty cycle for a given inpu t and output voltage as : OUT ONON MAX IN SW ON OFF(MIN) Vtt = and D = V f t t The actual on -time will be slightly longer as the IC compensates for voltage drops in the circuit, but we can neglect both of these since the on -time increase compensates for the voltage losses. Calculate the output voltage sag as : OUTSAG OUT IN(MIN) MAX OUT L ( I )V = 2 C V D V The amplitude of the capacitive soar is a function of the load step, the output capacitor value, the inductor value and the output voltage : OUTSOAR OUT OUT L ( I )V = 2 C V For the Typical Operat ing Circuit for 1.2V output, the circuit has an inductor 2.16H and 1 x 22 F output capacitance with 5m ESR each. The ESR step is 2A x 5m = 10mV which is small, as expected. The output voltage sag and soar in response to full 0A -2A-0A instantaneous transients are : ON MAX 1.2Vt = = 200ns 12V 500kHz 200nsand D = = 0.455200ns 240ns where 240ns is the minimum off time. SAG SOAR 2.16μH (2A)V 46.1mV2 22 μF 12V 0.455 1.2V 2.16μH (2A)V 163.6mV2 22 μF 1.2V The sag is about 3.84% of the output voltage and the soar is a full 13.6% of the output voltage. The ESR step is negligible here but it does partially add to the soar, so keep that in mind whenever using higher -ESR output capacitors. The soar is typically much worse than the sag in high input, low -output step -down converters b ecause the high input voltage demands a large inductor value which stores lots of energy that is all transferred into the output if the load stops drawing current. Also, for a given inductor, the soar for a low output voltage is a greater voltage change an d an even greater percentage of the output voltage. Any sag is always short -lived, since the circuit quickly sources current to regain regulation in only a few switching cycles. With the RT 6212B, any overshoot transient is typically also short-lived since the converter will sink current, reversing the inductor current sharply until the output reaches regulation again. The RT6212A discontinuous operation at light loads prevents sinking current so, for that IC, the output voltage will soar until load current or leakage brings the voltage down to normal. Most applications never experience instantaneous full load steps and the RT 6212A/B high switching frequency and fast transient response can easily control voltage regulation at all times. Also, since the sag and soar both are proportional to the square of the load change, if load steps were reduced to 1A (from the 2A examples preceding) the voltage changes would be reduced by a factor of almost ten. For these reasons sag and soar are seldom an issue except in v ery low-voltage CPU core or DDR memory supply applications, particularly for devices with high clock frequencies and quick changes into and out of sleep modes. In such applications, simply increasing the amount of ceramic output capacitor (sag and soar are directly proportional to capacitance) or adding extra bulk capacitance can easily eliminate any excessive voltage transients.
Copyright © 2017 Richtek Technology Corporation. All rights reserved. is a registered trademark of Richtek Technology Corporation. www.richtek.com DS6212A/B-05 July 2017 Outline Dimension Symbol Dimensions In Millimeters Dimensions In Inches Min. Max. Min. Max. A 0.700 1.000 0.028 0.039 A1 0.000 0.100 0.000 0.004 B 1.397 1.803 0.055 0.071 b 0.300 0.559 0.012 0.022 C 2.591 3.000 0.102 0.118 D 2.692 3.099 0.106 0.122 e 0.950 0.037 H 0.080 0.254 0.003 0.010 L 0.300 0.610 0.012 0.024 TSOT-23-6 (FC) Surface Mount Package Richtek Technology Corporation 14F, No. 8, Tai Yuen 1st Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications witho ut notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should ver ify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent righ ts of Richtek or its subsidiaries.