TLE8386-2EL INFINEON | Alldatasheet
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Technical content
Rev. 1.0, 2010-10-25 TLE8386-2EL Basic Smart Boost Controller
Data Sheet 2 Rev. 1.0, 2010-10-25 TLE8386-2EL Table of Contents Table of Contents
TLE8386-2EL PG-SSOP-14 TLE8386-2EL Data Sheet 3 Rev. 1.0, 2010-10-25 Basic Smart Boost Controller TLE8386-2EL 1O v e r v i e w
Features
- Wide Input Voltage Range from 4.75 V to 45 V
- Constant Current or Constant Voltage Regulation
- Very Low Shutdown Current: Iq< 2µA
- Flexible Switching Frequency Range, 100 kHz to 700 kHz
- Synchronization with external clock source
- Available in a small thermally enhanced PG-SSOP-14 package
- Internal 5 V Low Drop Out Voltage Regulator
- Output Overvoltage Protection
- External Soft Start adjustable by capacitor
- Over Temperature Shutdown
- Automotive AEC Qualified
- Green Product (RoHS) Compliant
Description
The TLE8386-2EL is a boost controller with built in protection features. The main function of this device is to step- up (boost) an input voltage to a larger output voltage. The switching frequency is adjustable from 100 kHz to 700 kHz and can be synchronized to an external cloc k source. The TLE8386-2EL features an enable function reducing the shut-down current consumption to < 2 µA. The current mode regulation scheme of this device provides a stable regulation loop ma intained by small external compensati on components. The integrated soft- start feature with external components for adjustment limits the current peak as well as voltage overshoot at start- up. This IC is suited for use in the harsh automotive environments and provides protection functions such as output overvoltage protection and over temperature shutdown.
Data Sheet 4 Rev. 1.0, 2010-10-25 TLE8386-2EL Block Diagram
2 Block Diagram
Comp. Internal Supply On/Off Logic Leading Edge Blanking EN_INT Thermal Protection Over Voltage Protection PWM Generator Power Switch Gate Driver Feedback Voltage Error Amplifier Power On Reset BlockDiagram .vsd Switch Current Error Amplifier SYNC 10 Synchroni sation TLE8386-2EL
Data Sheet 5 Rev. 1.0, 2010-10-25
3 Pin Configuration
3.1 Pin Assignment
Figure 2 Pin Configuration
3.2 Pin Definitions and Functions
1I V C C Internal LDO Output; Used for internal biasing and gate drive. Do not leave open, bypass with external capacitor. Do not connect other circuitry to this pin. 2S W O Switch Output; Connect to the gate of external boost converter switching MOSFET. 3S G N D Current Sense Ground; Ground return for current sense switch, connect to bottom side of sense resistor. 4S W C S Current Sense Input; Detects the peak current through switch, connect to high side of sense resistor.
5 SST Soft Start;
Connect an external capacitor to adjust the soft start ramp, do not leave open. 6F B Feedback; Output voltage feedback, connect to output voltage via resistor divider from output capacitor to ground. 7N C Not Connected; 8C O M P Compensation Input; Connect R and C network to improve the stability of the regulation loop. &203 6<1& )5(4 *1' ,9&& 667 6*1' 6:&6 6:2 SLQFRQILJBVVRSVYJ
Data Sheet 6 Rev. 1.0, 2010-10-25 TLE8386-2EL Pin Configuration 9N C Not Connected;
10 SYNC Sync;
Synchronization Input, if feature synchronization is not used, leave open.
11 FREQ Frequency Select Input;
Connect external resistor to GND to set frequency, do not leave open.
12 GND Ground;
Connect to system ground.
13 EN Enable;
Apply logic high signal to enable device.
14 IN Supply Input;
Supply for internal biasing, connect to input voltage. Exposed Pad Connect to GND. Pin Symbol Function
General Product Characteristics Data Sheet 7 Rev. 1.0, 2010-10-25
4 General Product Characteristics
4.1 Absolute Maximum Ratings
Absolute Maximum Ratings1) Tj = -40 °C to +150 °C; all voltages with respect to ground, positive current flowing into pin (unless otherwise specified) Pos. Parameter Symbol Limit Values Unit Conditions Min. Max. Voltages 4.1.1 IN Supply Input VIN -0.3 45 V 4.1.2 EN Enable Input VEN -40 45 V
4.1.3 FB;
Feedback Error Amplifier Input VFB -0.3 5.5 V 4.1.4 -0.3 6.2 V t < 10s
4.1.5 SWCS
Switch Current Sense Input VSWCS -0.3 5.5 V 4.1.6 -0.3 6.2 V t < 10s
4.1.7 SWO
VSWO -0.3 5.5 V 4.1.8 -0.3 6.2 V t < 10s
4.1.9 SGND
VSGND -0.3 0.3 V
4.1.10 COMP
VCOMP -0.3 5.5 V 4.1.11 -0.3 6.2 V t < 10s 4.1.12 FREQ; Frequency Input VFREQ -0.3 5.5 V 4.1.13 -0.3 6.2 V t < 10s 4.1.14 SYNC; Synchronization Input VSYNC -0.3 5.5 V 4.1.15 -0.3 6.2 V t < 10s 4.1.16 SST; Softstart Setting Input VSST -0.3 5.5 V 4.1.17 -0.3 6.2 V t < 10s
4.1.18 IVCC
Internal Linear Voltage Regulator Output VIVCC -0.3 5.5 V 4.1.19 -0.3 6.2 V t < 10s Temperatures
4.1.20 Junction Temperature
Tj -40 150 °C–
4.1.21 Storage Temperature Tstg -55 150 °C–
Data Sheet 8 Rev. 1.0, 2010-10-25 TLE8386-2EL General Product Characteristics Note: Stresses above the ones listed here may cause permanent damage to the device. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Note: Integrated protection functions are designed to prevent IC destruction under fault conditions described in the data sheet. Fault conditions are considered as “outside” normal operating range. Protection functions are not designed for continuous repetitive operation.
4.2 Functional Range
Note: Within the functional range the IC operates as described in the circuit description. The electrical characteristics are specified within the conditions given in the related electrical characteristics table.
4.3 Thermal Resistance
Note: This thermal data was generated in accordance with JEDEC JESD51 standards. For more information, go to www.jedec.org. ESD Susceptibility
4.1.22 ESD Resistivity to GND VESD,HBM -2 2 kV HBM 2)
4.1.23 ESD Resistivity to GND VESD,CDM -500 500 V CDM 3)
4.1.24 ESD Resistivity Pin 1, 7, 8, 14 (corner
pins) to GND VESD,CDM,C -750 750 V CDM 3) 1) Not subject to production test, specified by design. 2) ESD susceptibility, Human Body Model “HBM” according to EIA/JESD 22-A114B 3) ESD susceptibility, Charged Device Model “CDM” EIA/JESD22-C101 or ESDA STM5.3.1 Pos. Parameter Symbol Limit Values Unit Conditions Min. Max. 4.2.1 Supply Voltage Input VIN 4.75 45 V VIVCC > VIVCC,RTH,d Pos. Parameter Symbol Limit Values Unit Conditions Min. Typ. Max.
4.3.1 Junction to Case 1)
1) Not subject to production test, specified by design. RthJC –1 0 –K / W –
4.3.2 Junction to Ambient 1) 2)
2) Specified RthJA value is according to JEDEC 2s2p (JESD 51-7) + (JESD 51-5) and JEDEC 1s0p (JESD 51-3) + heatsink area at natural convection on FR4 board; RthJA –4 7 –K / W 2 s 2 p
4.3.3 RthJA – 54 – K/W 1s0p + 600 mm 2
4.3.4 RthJA – 64 – K/W 1s0p + 300 mm 2
Absolute Maximum Ratings1) Tj = -40 °C to +150 °C; all voltages with respect to ground, positive current flowing into pin (unless otherwise specified) Pos. Parameter Symbol Limit Values Unit Conditions Min. Max.
Data Sheet 9 Rev. 1.0, 2010-10-25
5 Boost Regulator
5.1 Description
The TLE8386-2 boost (step-up) regulator provides a higher output voltage than input voltage. The PWM controller measures the output voltage via a resistor divider co nnected between Pin FB and ground, and determines the appropriate pulse width duty cycle (on time). An over volt age protection swit ches off the converter case if the voltage at Pin FB exceeds the over voltage limit. If the connection to the output voltage resistor divider should be lost, an internal current so urce connected to Pin FB will draw the volt age above this limit and shut the external MOSFET off. The current mode controller has a built-in slope compensation to prevent sub-harmonic oscillations which is a characteristic of current mode controllers operating at high duty cycles (>50% duty). An additional built- in feature is an integrated soft start that limits the current through the inductor and the external power switch during initialization. The soft-start time TSS is adjustable using an external capacitor CSST: The switching frequency may be adjusted by using an external resistor (please refer to chapter Oscillator and Synchronization). If synchronization to an external frequency source is used, the internal frequency has to be adjusted close to this external source. Figure 3 Boost Regulator Block Diagram TSS CSST 20 0 V, COMP FB FREQ Oscillator Gate Driver SWCS4 SWO2S R Q D VOVFB,TH SGND Logic Temp. Sensor Soft Start Feedback Error Amplifier Over Voltage Comparator VIVCC Current Sense OTA B oost_Diag .vsd SYNC Synchroni zation10 VRef Slope Comp. gmEA 5SST Soft Start PWM Curr Comparator
Data Sheet 10 Rev. 1.0, 2010-10-25 TLE8386-2EL Boost Regulator
5.2 Electrical Characteristics
VIN = 6V to 40V; Tj = -40 °C to +150 °C, all voltages with respect to ground, positive current flowing into pin; (unless otherwise specified) Pos. Parameter Symbol Limit Values Unit Conditions Min. Typ. Max. Boost Regulator: IBO = 100 to 500 mA
5.2.2 Voltage Line Regulation ∆VREF
/∆VIN – – 0.15 %/V VIN = 6 to 19 V; VBO= 30 V; IBO = 100 mA Figure 8
5.2.3 Voltage Load Regulation ∆VFB
/∆IBO ––5% / A VIN = 13V; VBO = 30V; IBO = 100 to 500 mA Figure 8
5.2.4 Switch Peak Over Current
VSWCS 120 150 180 mV VIN = 6 V VFB < VFBOV VCOMP = 3.5V
5.2.5 Current to Softstart setting
ISST -8 -10 -16 µA
5.2.6 Feedback Input Current IFB -200 nA
5.2.7 Switch Current Sense Input
ISWCS -10 -50 -100 µA VSWCS = 150 mV 5.2.8 Input Undervoltage Shutdown VIN,off 3.75 – – V VIN decreasing 5.2.9 Input Voltage Startup VIN,on ––4 . 7 5 V VIN increasing Gate Driver for Boost Switch
5.2.10 Gate Driver Peak Sourcing
Current1) ISWO,SRC –- 3 8 0 –m A VSWO = 3.5V
5.2.11 Gate Driver Peak Sinking
Current1) ISWO,SNK –5 5 0 –m A VSWO = 1.5V 5.2.12 Gate Driver Output Rise Time tR,SWO – 3 06 0n s CL,SWO = 3.3nF; VSWO = 1V to 4V 5.2.13 Gate Driver Output Fall Time tF,SWO – 2 04 0n s CL,SWO = 3.3nF; VSWO = 1V to 4V 1) Not subject to production test, specified by design
Data Sheet 11 Rev. 1.0, 2010-10-25 Efficiency depending on Input Voltage VIN and output Current IBO ,%2>$@ (IILFLHQF\\>@ (IILFLHQF\\IRU9,1 9 ,%2>$@ (IILFLHQF\\>@ (IILFLHQF\\IRU9,1 9 ,%2>$@ (IILFLHQF\\>@ (IILFLHQF\\IRU9,1
Data Sheet 12 Rev. 1.0, 2010-10-25 TLE8386-2EL Boost Regulator Load regulation Input Voltage VIN = 6V Load regulation Input Voltage VIN = 13.5 Load regulation Input Voltage VIN = 19V /RDG5HJXODWLRQYV7HPS $,RXW$ 7HPS /LQH5HJ /RDG5HJXODWLRQYV7HPS $,RXW$ 7HPS /LQH5HJ /RDG5HJXODWLRQYV7HPS $,RXW$ 7HPS /LQH5HJ
Oscillator and Synchronization Data Sheet 11 Rev. 1.0, 2010-10-25
6 Oscillator and Synchronization
6.1 Description
R_OSC vs. switching frequency The internal oscillator is used to determine the switching frequency of the boost regulator. The switching frequency can be selected from 100 kHz to 700 kHz with an external resistor to GND. To set the switching frequency with an external resistor the following formula can be applied. In addition, the oscillator is capable of changing from the frequency set by the external resistor to a synchronized frequency from an external clock source. If an external clock source is provided on the pin SYNC, the internal oscillator should adjusted clos e to this frequency. Then it synchronizes to this exte rnal clock frequency and the boost regulator switches at the synchronized frequency. The synchronization frequency capture range is from 250 kHz to 700 kHz. Figure 4 Oscillator and Synchronization Block Diagram and Simplified Application Circuit Figure 5 Synchronization Timing Diagram Clock Frequency Detector Oscillator SYNC Multiplexer PWM Logic Gate Driver SWO RFREQ VCLK Oscillator_BlkDiag.vsd FREQ TLE8386-2EL 96<1& W W6<1&75 W6<1&75 96<1&+ 96<1&/ 76<1& I6<1& W6<1&3:+ 2VFLOODWRUB7LPLQJVYJ Ω − 3 112 105.310141 sFREQ s fR FREQ
Data Sheet 12 Rev. 1.0, 2010-10-25 TLE8386-2EL Oscillator and Synchronization
6.2 Electrical Characteristics
VIN = 6V to 40V; Tj = -40 °C to +150 °C, all voltages with respect to ground, positive current flowing into pin; (unless otherwise specified) Pos. Parameter Symbol Limit Values Unit Conditions Min. Typ. Max. Oscillator:
6.2.1 Oscillator Frequency fFREQ 250 300 350 kHz RFREQ = 20kΩ
6.2.2 Oscillator Frequency
fFREQ 100 – 700 kHz 17% internal tolerance + external resistor tolerance
6.2.3 FREQ Supply Current IFREQ ––- 7 0 0 µ A VFREQ = 0 V
6.2.4 SYNC input internal pull-
RSYNC 150 250 350 k Ω VSYNC= 5V
6.2.5 Maximum Duty Cycle DMAX,fixed 90 93 95 % Fixed frequency mode
6.2.6 Maximum Duty Cycle DMAX,sync 88 – – % Synchronization mode,
internal frequency (set by resistor) is 0.8 to 1.2
6.2.7 Synchronization Frequency
fSYNC 250 – 700 kHz ratio between synchronization and internal frequency (set by resistor) is 0.8 to 1.2
6.2.8 Synchronization Signal Duty
TD_SYNC 20 80 %
6.2.9 Synchronization Signal
VSYNC,H 3 . 0 ––V 1) 1) Synchronization of external SWO ON signal to falling edge
6.2.10 Synchronization Signal
VSYNC,L ––0 . 8 V 1)
Oscillator and Synchronization Data Sheet 13 Rev. 1.0, 2010-10-25 Typical Performance Characteristics of Oscillator Switching Frequency fSW versus Frequency Select Resistor to GND RFREQ Oscillator_fFreq_vs_Rfreq.vsd 100 200 300 400 500 0 1 02 03 04 05 06 07 08 0 R FREQ [kohm] fFREQ [kHz] Tj = 25 °C 600 700
Data Sheet 14 Rev. 1.0, 2010-10-25 TLE8386-2EL Enable Function
7 Enable Function
7.1 Description
The enable function powers on or off the device. A valid l ogic low signal on enable pin EN powers off the device and current consumption is less than 2 µA. A valid logic high enable signal on enable pin EN powers on the device. The Enable Startup Time tEN,START is the time between the Enable signal is recognized as valid and the device starts to switch. During this period of time the internal supplies, bandgap are initalized and reach their nominal values. The TLE8386-2 will start to switch after the nominal values are reached. Figure 6 Timing Diagram Enable 9(1 9(121 9,9&& 9,9&&21 96:2 9(12)) W W W 3RZHU2II ,T$ 1RUPDO 6:22Q (1B7LPLQJVYJ 3RZHU2Q W(167$57V
Data Sheet 15 Rev. 1.0, 2010-10-25
7.2 Electrical Characteristics
VIN = 6V to 40V; Tj = -40 °C to +150 °C, all voltages with respect to ground, positive current flowing into pin; (unless otherwise specified) Pos. Parameter Symbol Limit Values Unit Conditions Min. Typ. Max. Enable Input:
7.2.1 Enable
VEN,ON 3.0 – V –
7.2.2 Enable
VEN,OFF ––0 . 8 V –
7.2.3 Enable Hysteresis VEN,HYS 50 200 400 mV –
7.2.4 Enable
IEN,H ––3 0 µ A VEN = 16.0 V
7.2.5 Enable
IEN,L –0 . 1 1µ A VEN = 0.5 V
7.2.6 Enable Startup Time 1)
1) Not subject to production test, specified by design. tEN,START 100 – – µs – Current Consumption
7.2.7 Current Consumption,
Iq_off ––2µ A VEN = 0.8 V; Tj ≤ 105C; VIN = 16V
7.2.8 Current Consumption,
Active Mode2) 2) Dependency on switching frequency and gate charge of boost. Iq_on ––7m A VEN ≥ 4.75 V; IBO = 0 mA; VIN = 16V VSWO = 0% Duty
Data Sheet 16 Rev. 1.0, 2010-10-25 TLE8386-2EL Linear Regulator
8 Linear Regulator
8.1 Description
The internal linear voltage regulator supplies the internal gate drivers with a typical voltage of 5 V and current up to 50 mA. An external output capacito r with low ESR is required on pin IVCC for stability and buffering transient load currents. During normal operatio n the external boost MOSFET switch will draw transient currents from the linear regulator and its output capacitor. Proper sizing of the output capacitor must be considered to supply sufficient peak current to the gate of the external MOSFET switch. Please refer to application section for recommendations on sizing the output capacitor. An integrated power-on reset circuit monitors the linear regulator output voltage and resets the device in case the output voltage falls below the power-on reset threshold. The power-on reset helps protect the external switches from excessive power dissipation by ensuring the gate drive voltage is sufficient to enhance the gate of an external logic level n-channel MOSFET. IVCC stays at around 300 mV when Enable signal is off. No external circuit should be connected to IVCC Figure 7 Voltage Regulator Block Diagram and Simplified Application Circuit
8.2 Electrical Characteristics
VIN = 6V to 40V; Tj = -40 °C to +150 °C, all voltages with respect to ground, positive current flowing into pin; (unless otherwise specified) Pos. Parameter Symbol Limit Values Unit Conditions Min. Typ. Max. 8.2.1 Output Voltage VIVCC 4.6 5 5.4 V 6 V ≤ VIN ≤ 45 V 0.1 mA ≤ IIVCC ≤ 50 mA 8.2.2 Output Current Limitation ILIM 51 110 mA VIN = 13.5 V VIVCC = 4.5V
8.2.3 Drop out Voltage VDR 1000 mV IIVCC = 50mA 1)
1) Measured when the output voltage VCC has dropped 100 mV from its nominal value. 8.2.4 Output Capacitor CIVCC 0.47 3 µF 2) 2) Minimum value given is needed for regulator stability; a pplication might need higher capacitance than the minimum. 8.2.5 Output Capacitor ESR RIVCC,ESR 0.5 Ω f = 10kHz
8.2.6 Undervoltage Reset Headroom VIVCC,HDRM 1 0 0 ––m V VIVCC decreasing
VIVCC - VIVCC,RTH,d 8.2.7 Undervoltage Reset Threshold VIVCC,RTH,d 4 . 0 ––V VIVCC decreasing 8.2.8 Undervoltage Reset Threshold VIVCC,RTH,i ––4 . 5 V VIVCC increasing ,9&& /LQHDU5HJXODWRU /LQ5HJB%OFN'LDJVYJ *DWH 'ULYHU
Protection and Diagnostic Functions Data Sheet 17 Rev. 1.0, 2010-10-25
9 Protection and Diag nostic Functions
9.1 Description
The TLE8386-2EL has integrated circuits to prot ect against output over voltage, open feedback and overtemperature faults. During an ov ervoltage the gate driver outputs SW O will turn off. In the event of an overtemperature condition the integrated thermal shutdown function turns off the gate drivers and internal linear voltage regulator. If the connection from pin FB to the ou tput voltage resistor divider should be lost, an internal current source connected to Pin FB will draw the voltage above this limit and shut the external MOSFET off. The typical junction shutdown te mperature is 175°C. After cooling down t he IC will automatically restart operation. Thermal shutdown is an integrated prot ection function designed to prevent immediate IC destruction and is not intended for continuous use in normal operation.
9.2 Electrical Characteristics
Note: Integrated protection functions are designed to prevent IC destruction under fault conditions described in the data sheet. Fault conditions are considered as “outside” normal operating range. Protection functions are not designed for continuous repetitive operation. VIN = 6V to 40V; Tj = -40 °C to +150 °C, all voltages with respect to ground, positive current flowing into pin; (unless otherwise specified) Pos. Parameter Symbol Limit Values Unit Conditions Min. Typ. Max. Temperature Protection:
9.2.1 Over Temperature Shutdown Tj,SD 160 175 190 °C –
9.2.2 Over Temperature Shutdown
Tj,SD,HYST –1 5 –° C – Overvoltage Protection:
9.2.3 Output Over Voltage Feedback
VOVFB,TH 8 10 12 % 10% higher of regulated voltage
9.2.4 Output Over Voltage Feedback
VOVFB,HYS 5 % Output Voltage decreasing
9.2.5 Over Voltage Reaction Time tOVPRR 2 – 10 µs Output Voltage
Data Sheet 18 Rev. 1.0, 2010-10-25 TLE8386-2EL
Application Information
Note: The following information is given as a hint for the implementation of the device only and shall not be regarded as a description or warranty of a certain functionality, condition or quality of the device.
10.1 Boost Converter Application Circuit
Figure 8 Boost Converter Application Circuit Figure 9 Boost Application Circuit Bill of Material Note: This is a simplified example of an application circuit. The function must be verified in the real application. 9%2 6<1& &203 )5(4 667 *1' 6:2 6:&6 6*1' ,9&& 7/((/ 5&6 5)%/ 5)%+ &287 /%2267/,1387 &,1&,1 9,1 '%2267 &667&&203 5&203 5)5(4 &,9&& 532/ '32/ $SS'LDJ%RRVW &,1XQG/,1387UHFRPPHQGHGIRU VXSSUHVVLRQRI(0( &&203 LBOOST Reference Designator DBOOST Part NumberManufacturer COUT RCS RFBH CCOMP RCOMP CIVCC RFREQ IC1 Vishay Coilcraft MSS1278T-104ML_ EEVFK1K101QPanasonic Value 100 uH 100 uF, 80V Type Diode Capacitor Capacitor Capacitor IC Inductor Resistor Resistor Resistor Resistor Panasonic100 uF, 6.3V EEFHD0J101R Infineon TLE8386-2EL-- 11 kΩ, 1% Panasonic ERJ3EKF1102V 20 kΩ, 1% Panasonic ERJ3EKF2002V Panasonic SS3H10Schottky, 3 A, 100 VR 10 nF -- Panasonic10 kΩ ERJ3EKF1002V Quantity CIN1 EEEFK1H101GPPanasonic100 uF, 50V Capacitor 1 50 mΩ, 1% ERJB1CFR05U RFBL Resistor1 kΩ, 1% Panasonic ERJ3EKF1001V 1 CSST -- Capacitor4,7 nF -- 1
Data Sheet 19 Rev. 1.0, 2010-10-25
10.1.1 Principle:
The TLE8386-2EL can be configured as a boost converter, where the desired output voltage VBO is always higher than the input voltage VIN. A boost convertor is not short-circuit protected. If the output voltage VBO is shorted, the output current will only be limited by the input voltage VIN capability. A typical boost converter application is shown in Figure 8, the elements and abbreviations and their meanings are:
- L BOOST = boost inductor
- L INPUT = input filter inductor, recommended to reduce electromagnetic emissions
- C IN1 = input filter capacitor
- C IN2 = additional input filter capacitor, recommended to reduce electromagnetic emissions
- C OUT = output filter capacitor
- D BOOST = output diode
- V IN = input voltage
- V INMIN = minimum input voltage
- V BO = boost output voltage
- R CS = current sense resistor
- R FBH = boost output voltage resistor divider, highside resistor
- R FBL = boost output voltage resistor divider, lowside resistor
- R COMP, CCOMP = compensation network elements
- R FREQ = frequency setting resistor
- C SST = softstart setting capacitor
- C IVCC = capacitor for internal LDO
- D = duty cycle MAX = maximum duty cycle
- f FREQ = Switching Frequency
- I IN = input current
- I BO = output current
- I BOMAX = maximum output current The ratio between input voltage VIN and output voltage VBO in continuos conduction mode (CCM) is: In discontinous conduction mode (DCM) the conversion ratio at a fixed frequency is higher, the switching current increases and efficiency is reduced. The maximum duty cycle DMAX occurs for minimum input voltage VINMIN. VBO VIN VBO
Data Sheet 20 Rev. 1.0, 2010-10-25 TLE8386-2EL
10.1.2 Component Selection:
Power MOSFET selection: The important parameters for the choice of the power MOSFET are:
- Drain-source voltage rating V DS: The power MOSFET will see the full output voltage VBO plus the output diode (DBOOST) forward voltage. During its off-time additional ringing across drain-to-source will occur.
- On-resistance R DSON for efficiency reasons and power dissipation
- Maximum drain current I DMAX
- Gate-to-source charge and gate-to-drain charge
- Thermal resistance It is recommended to choose a power MOSFET with a drain-source voltage rating V DS of at least 10 V higher than the output voltage VBO. The power dissipation PLOSSFET in the power MOSFET can be calculated using the following formula:
- C RSS = reverse transfer capacitance, please refer to power MOSFET data sheet
- I BOOSTMAX = maximum average current through the boost inductor LBOOST. The first term in the equation above gives the conductio n losses in the power MOSFET, the second term the switching losses. To optimize the efficiency, RDSON and CRSS should be minimized. PLOSSFET IBOOSTMAX
2 RDSON× 2V BO
2 IBOOSTMAX CRSS
Data Sheet 21 Rev. 1.0, 2010-10-25 Current sense resistor RCS selection: For control and protection, the TLE8386-2EL measures the power MOSF ET current by a current sense resistor RCS, which is located between the power MOSFET source and ground. For proper function it is very important:
- To locate the current sense resistor as close as possible to the TLE8386-2EL
- To use short (low resistive and low inductive) tr aces between the power MOSFET source and ground.
- To use short (low resistive and low inductive) traces between the current sense resistor RCS highside and lowside and the pins SWCS and SGND (it is not recommended to use pin GND instead of pin SGND for power MOSFET current measurement).
- The value of R CS should be selected to make sure that the maximum peak sense voltage VSENSEPEAK during steady state normal operation will be lower than the adjusted current limit threshold (current limit function!). It is recommended to give a 20% margin.
- The value of R CS should be selected to make sure that the power MOSFET maximum drain current IDMAX will not be exceeded (please refer to power MOSFET data sheet). The figure below shows the voltage waveform over the current sense resistor RCS during a switching cycle: Figure 10 Sense voltage VSENSE waveform during a switching cycle
- V SENSEMAX = maximum average sense voltage at maximum output current IBO measured during on-time.
- V SENSEPEAK = maximum peak sense voltage at maximum output current IBO at end of on-time.
- ∆VSENSE = ripple voltage across RCS (switch ripple current) during on-time, represents the peak-to-peak ripple current in the boost inductor LBOOST. The maximum (peak-to-peak) switch current ripple percentage χ (will be needed for further calculations of inductor values) can be calculated considering the 20% margin by following equation:
- V SWCS = Switch peak over current threshold
- χ is recommended to fall in the range between 0.2 to 0.6 (please refer to calculations in the following chapters) t On-Time VSENSE VSENSEMAX VSENSEPEAK ∆VSENSE Switching Cycle χ ∆VSENSE
Data Sheet 22 Rev. 1.0, 2010-10-25 TLE8386-2EL The value of the sense resistor RCS can be calculated as follows:
- I BOOSTPEAK = peak current through the boost inductor LBOOST (will be calculated at boost inductor selection) Boost inductor LBOOST selection: The important parameters for selecting the boost inductor are:
- Inductor L BOOST
- Maximum RMS current rating I BOOSTRMS for thermal design
- Saturation current threshold I BOOSTSAT The maximum average inductor current is: The ripple current through the boost inductor is: The peak current through the boost inductor is: (The peak current trough the boost inductor must be smaller than the saturation current threshold!) The RMS current through the boost inductor is: The boost inductor value LBOOST can be calculated by the following equation: RCS 08 0, VSWCS× IBOOSTPEAK IBOOSTMAX IBOMAX ∆IBOOST χ IBOOSTMAX×χ I× BOMAX IBOOSTPEAK IBOOSTMAX 1 χ 2---+ × IBOOSTSAT<= IBOOSTRMS IBOOSTMAX 1 χ2 LBOOST VINMIN
Data Sheet 23 Rev. 1.0, 2010-10-25 In fixed frequency mode an external resistor determines the switching frequency. The minimum boost inductor for fixed frequency is given by the formula below:
- L BOOSTMIN = minimum Inductance required (minimum value of LBOOST) Following the previous equations the user should choose the boost inductor having sufficient saturation and RMS current ratings. The boost inductor value influences the current ripple ∆IBOOST:
- A larger boost inductor value decreases the current ripple ∆IBOOST, but reduces also the current loop gain.
- A lower boost inductor value increases the current ripple ∆IBOOST, but provides faster transient response. A lower boost inductor value also results in higher input current ripple and greater core losses. Output diode DBOOST selection: Guidelines to choose the diode:
- Fast switching diode
- Low forward drop
- Low reverse leakage current
- It is recommended to choose the repetitive reverse voltage rating V RRM (please refer to diode data sheet) at least 10V higher than the boost converter output voltage VBO. The average forward current in normal operation is equal to the boost converter output current I BO and the peak current through the diode IDPEAK (occurs in off-time of the power MOSFET) is: The power dissipation PLOSSDIO in the output diode DBOOST is:
- V D = forward drop voltage of diode DBOOST (please refer to diode data sheet). LBOOSTMIN VBO V[] RCS Ω[]× 106 3–×10 V[] fFREQ Hz[]× IDPEAK I= BOOSTPEAK IBOOSTMAX 1 χ 2---+ ×= PLOSSDIO IBOMAX VD×=
Data Sheet 24 Rev. 1.0, 2010-10-25 TLE8386-2EL Output filter capacitor COUT selection: Choosing the correct output capacitor for given output ripple voltage, the influence of
- ESR = equivalent series resistance,
- ESL = equivalent series inductance and
- bulk capacitance have to be considered. The effects of these three parameters is additional ringing on the output voltage VBO. The voltage ripple at the output voltage VBO depends on:
- ∆VESR: in percent, related to the ESR of the output capacitor(s)
- ∆VCOUT: in percent, related to the bulk capacitance of the output capacitor(s)
- To receive the total voltage ripple, the influence of ∆VESR and ∆VCOUT must be counted together. The output capacitor can be calculated using the followin g equation (which contains the influence of the bulk capacitance on the output voltage ripple): Influence of the capacitor ESR on the output voltage ripple: The output capacitor experiences high RMS ripple currents, the RMS ripple current rating can be determined using the following formula:
- I COUTRMS = RMS ripple current rating at switching frequency IFREQ. To meet the ESR requirements often multiple capacitor s are paralleled. Typically, once the ESR requirement is met, the output capacitance is adequate for filtering and has the required RMS current rating. Additional ceramic capacitors are commonly used to reduce the effects of parasitic inductance to reduce high frequent switching noise on the boost converter output. COUT IBOMAX ESR COUT ∆VESR IDPEAK ICOUTRMS IBOMAX DMAX
Data Sheet 25 Rev. 1.0, 2010-10-25 Input filter capacitor CIN1 selection: The input filter capacitor C IN1 has to compensate the alternate current c ontent or current ripple on the input line, recommended values are from 10µF to 100µF, to improve the suppression of high frequent distortions a parallel ceramic capacitor might be necessary. The RMS input capacitor ripple current IIN1RMS for a boost converter is: Compensation network elements RCOMP, CCOMP selection: To compensate the feedback loop of the TLE8386-2EL a series network of RCOMP, CCOMP is usually connected from pin COMP to ground. For most applications the capacitor CCOMP should be in the range of 470pF to 22nF, and the resistor RCOMP should be in the range of 5kΩ to 100kΩ. An additional capacitor CCOMP2 might be usefull to improve stability. CCOMP and CCOMP2 should be a low ESR ceramic capacitors. A practical approach to determine the compensation network is to start with the application circuit as shown in the data sheet and tune the comp ensation network to optimize the performance. Stability of the loop should then be checked under all operating conditions, including output current and variations and over the entire temperature range. Output boost voltage VBO adjustment by determining the output voltage resistor divider RFBH, RFBL:
- V FB = feedback reference voltage (VBO is always higher than VIN during operation of the boost converter) Additional input filter inductor LINPUT and capacitor CIN2 selection:
- f FILTER = resonance frequency of the additional input filter The input filter inductor LINPUT should have a saturation current value equal to L BOOST, capacitor CIN2 should be a low ESR ceramic capacitor. Both elements are forming a low pass filter to suppress conducted disturbances on the VIN line. To obtain an optimum suppression, the input filter resonance frequency fFILTER should be at least ten times lower than the switching frequency fFREQ: The use of an additional input filter is depending on the requirements of the application. For selection of RFREQ, CSST and CIVCC please refer to previous chapters. IIN1RMS 03 0, ∆IBOOST×= VBO VFB RFBH RFBL+ RFBL fFREQ 10 f FILTER 2Π LINPUT CIN2× ×>
Data Sheet 26 Rev. 1.0, 2010-10-25 TLE8386-2EL
10.2 Further Information on TLE8386-2EL
10.2.1 General Layout recommendations
Introduction: A boost converter is a potential source of electromagnetic disturbances which may affect the environment as well as the device itself and cause sporadic malfunction up to damages depending on the amount of noise. In principal we may consider the following basic effects:
- Radiated magnetic fields caused by circular currents, occurring mostly with the switching frequency and their harmonics
- Radiated electric fields, often c aused by (voltage) oscillations
- Conducted disturba nces (voltage spikes or oscillations) on the lines, mostly input and output lines. Radiated magnetic fields: Radiated magnetic fields are caused by circular currents occurring in so called “current windows”. These circular currents are alternating currents which are driven by th e switching transistor. The alternating current in these windows are driving magnetic fields. The amount of magnetic emissions is mostly depending on the amplitude of the alternating current and the size of the so-called “wi ndow” (this is the area, whic h is defined by the circular current paths. We can divide into two windows:
- the input current “window” (path consisting of CIN1, LBOOST and the power MOSFET): Only the alternate content of the input current IIN is considered.
- the output current “window”: (path consisting of the power MOSFET, DBOOST and COUT): Output current ripple ∆I The area of these “windows” has to be kept as small as possible, with the relating elements placed next to each others. It is highly recommended to use a ground plane as a single layer which covers the complete regulator area with all components shown in this figure. All connections to ground shall be as short as possible Radiated electric fields: Radiated electric fields are caused by voltage osc illations occurring due to st ray inductances and stray capacitances at the connection be tween power MOSFET, output diode D BOOST and output capacitor C OUT. They are also of course influenced by the commutation of the current from the power MOSFET to the output diode DBOOST. Their frequencies might be between 10 and 100 MHz. Therefore it is recommended to use a fast Schottky diode and to keep the connections in this area as low in ductive as possible. This can be achieved by using short and broad connections and to arrange the related parts as close as possible. Following the recommendation of using a ground layer these low inductive connections will form together with the ground layer small capacitances which are desirable to damp the slope of these oscillations. The oscillations use connections or wires as antennas, this effect can also be minimized by the short and broad connections.
Data Sheet 27 Rev. 1.0, 2010-10-25 Conducted disturbances: Conducted disturbances are voltage spikes or voltage osc illations, occurring permanently or by occasion mostly on the input or output connections. Comparable to th e radiated electric fields they are caused by voltage oscillations occurring due to stray inductances and stray capacitances at the connection between power MOSFET, output diode DBOOST and output capacitor COUT. Their frequencies might be between 10 and 100 MHz. Th ey are super positioned to the input and output voltage and might thus disturb other components of the application. The countermeasures against conducted disturbances are similar to the radiated electric fields:
- it is recommended to use short an thick connecti ons between the single parts of the converter
- all parts shall be mounted close together
- additional Filter capacitors (ceramic , with low ESR) in parallel to the output and input capacitor and as close as possible to the switching parts. Input and load current must be forced to pass these devices, do not connect them via thin lines. Recommended values from 10nF to 220nF
- for the input filter a so called “p ” – Filter for maximum suppression might be necessary, which requires additional capacitors on the input
10.2.2 Additional information
- Please contact us for information regarding the Pin FMEA.
- and for existing application notes with more detailed information about the possibilities of this device
- For further information you may contact http://www.infineon.com/
Data Sheet 29 Rev. 1.0, 2010-10-25 Figure 11 PG-SSOP-14 Green Product (RoHS compliant) To meet the world-wide customer requirements for environmentally friendly products and to be compliant with government regulations the device is available as a green product. Green products are RoHS-Compliant (i.e Pb-free finish on leads and suitable for Pb-free soldering according to IPC/JEDEC J-STD-020). PG-SSOP-14-1,-2,-3-PO V02 1 7 14 8 14x0.25±0.05 2) M0.15 DC A-B 0.65 C Stand Off 0 ... 0.1 (1.45) 1.7 MAX. 0.08 C A B 4.9±0.11) A-BC0.1 2x 1) Does not include plastic or metal protrusion of 0.15 max. per side 2) Does not include dambar protrusion Bottom View ±0.23 ±0.22.65 0.2 ±0.2 D 6 M D 8x 0.64±0.25 3.9±0.11) 0.35 x 45˚ 0.1 CD +0.06 0.19 8˚ MAX. Index Marking Exposed Diepad For further package information, please visit our website: http://www.infineon.com/packages. Dimensions in mm
Data Sheet 30 Rev. 1.0, 2010-10-25 TLE8386-2EL
Revision History
1.0 Revision Date Changes 1.0 2010-10-25 Data Sheet
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