UM0670 STMICROELECTRONICS | Alldatasheet
Document overview
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Technical content
Datasheet sections
- 1 Introduction
- 2 Getting started
- 3 Design concept
- 4 STEVAL-ILL013V1 technical details
- 5 Schematic diagram
- 6 Bill of material
- 7 STEVAL-ILL013V1 performance
- 8 Dimming function
- 9 Measurement
- 9.1 Output waveform measurement
- 9.2 Power MOSFET turn ON and OFF time
- 9.3 LED current ripple reduction
- 9.4 Standard EN61000-3-2 measurement
- 9.5 EMI measurement (EN55015)
- 10 Design features
- 10.1 Proper startup circuit design
- 10.2 Zero dimming design implementation
- 11 References and related materials
- 12 Revision history
80 W offline LED driver with PFC
1 Introduction
robust and offer wider design flexibility compared to other light sources. which is described in detail in this user manual. can be ordered using order code STEVAL-ILL013V1, and is shown in Figure 1. Figure 1. STEVAL-ILL013V1 demonstration board
2 Getting started
to the calculated output voltages shown in Table 1. and 265 VAC, and the LE Ds begin illuminating. charged to 400 V and can cause uncontrolled peak LED current. Table 1. LED values for different output currents Table 2. Output LED current adjustment on the demonstration board
3 Design concept
the STEVAL-ILL013V1, the LED brightness is pre-adjusted to 100%. side switch, as in a standard buck topology) and therefore it is easier to control the switch. note AN2928 (see Section 11: References and related materials : 2). Figure 2. STEVAL-ILL013V1 block schematic
265 V AC
400 V 80 W LOAD
Figure 4. Modified buck converter with dimming design concept
80 W LEDs
4 STEVAL-ILL013V1 technical details
- 80 W LED driver
- 350 mA, 700 mA and 1 A LED current settings
- PF = 0.99 with VIN = 110 V or PF = 0.98 with V IN = 230 V
- THD (total harmonic distortion) = 4.6 and V IN = 110 V or THD = 10.3 and V IN = 230 V
- High PFC boost converter operating in transition mode
- Modified buck converter working in CCM and using FOT network
- Switching frequency f = 125 kHz / 350 mA (modified buck converter)
- Switching frequency f = 69 kHz / 700 mA (modified buck converter)
- Switching frequency f = 55 kHz / 1000 mA (modified buck converter)
- The same inductor and transformer core used (E25)
- Supply voltage provided for external PWM generator
- Board size: 130 mm x 60 mm x 27 mm
- Optional external PWM generator (non isolated)
- Full brightness if PWM generator is not connected
- Two output connectors for LEDs
- High efficiency (~90%)
- Wide input voltage range: 88 V to 265 VAC
- Brightness regulation between 0% and 100%
- EMI filter implemented
- EN55015 and EN61000-3-2 tested
Figure 5. STEVAL-ILL013V1 with PWM module
5 Schematic diagram
Figure 6. High PFC boost converter with the L6562A
Figure 7. Modified buck converter with the L6562A
6 Bill of material
Table 3. STEVAL-ILL013V1 demonstration board bill of material (1)
41 N T C 1 0 Ω NTC thermistor EPCOS B57235S100M
- The power MOSFET STF9NM50N can be replaced by STF10NM60N.
Table 3. STEVAL-ILL013V1 demonstration board bill of material (continued) (1)
7 STEVAL-ILL013V1 performance
References and related materials : 6.). Figure 8. Efficiency over the whole input voltage range
1 A at 4 LEDs
Figure 9. Power factor for wide input voltage range Figure 10. Detailed power factor for wide input voltage range
Figure 11. Total harmonic distortion for wide input voltage range
8 Dimming function
LEDs as a light source are very often used in applications where the brightness regulation is required. Their biggest advantage is that their minimum brightness can be easily regulated by changing their current, and they are stable even at very low brightness. Generally, there are two basic concepts regarding how the brightness is regulated. The first is called “analog dimming”, which means that the brightness is regulated by changing the continuous forward LED current. This concept is not used on the STEVAL-ILL013V1. The second solution is to use a low frequency (~200 Hz) PWM signal and change the brightness by pulse width modulation. This is the approach used in the design of the STEVAL-ILL013V1. Any external PWM generator can be used for brightness regulation, but it should be taken into account that the STEVAL-ILL013V1 is not isolated. In order to demonstrate the dimming function on the STEVAL-ILL013V1, an external PWM generator using STMicroelectronics’ ST7LITEU05 microcontroller was connected to the board, and the output LED current was measured. The microcontroller generates a PWM signal with a frequency of 250 Hz. The duty cycle is set by a potentiometer from 0% up to 100%.The result with duty cycles of 50%, 10% and 2% is shown in Figure 12. The input voltage was, in this case, 230 VAC and the output LED current was set to 350 mA. It is also possible to achieve LED brightness regulation below 2%. In this case the nominal LED current is slightly decreased.
Figure 12. Output LED current dimming capability
9 Measurement
9.1 Output waveform measurement
(OST AR LED LE UW E3B; see Section 11: References and related materials 6). different LED voltages is demonstrated in Figure 14. Figure 13. Output LED current waveform (I
Figure 18. Output LED current for different LED voltage (I LED = 1000 mA)
9.2 Power MOSFET turn ON and OFF time
Equation 13 in the appendix). Figure 19. Power MOSFET turn ON and OFF measurement
9.3 LED current ripple reduction
current ripple and good dimming resolution, as illustrated in Figure 12. Figure 20. LED current ripple for the 100 nF output capacitor
9.4 Standard EN61000-3-2 measurement
Figure 21. EN61000-3-2 analysis for LED current of 350 mA and V IN from 85 V to 160 VAC
85 V 100 V 110 V 120 V 140 V 160 V
Figure 22. EN61000-3-2 analysis for LED current of 350 mA and V IN from 180 V to 265 VAC Figure 23. EN61000-3-2 analysis for LED current of 700 mA and V IN from 85 V to 160 VAC Figure 24. EN61000-3-2 analysis for LED current of 700 mA and V IN from 180 V to 265 VAC
180 V 200 V 220 V 230 V 240 V 260V
180 V 200 V 220 V 230 V 240 V 260 V
Figure 25. EN61000-3-2 analysis for LED current of 1000 mA and V IN from 85 V to 160 VAC Figure 26. EN61000-3-2 analysis for LED current of 1000 mA and V IN from 180 V to 265 VAC
9.5 EMI measurement (EN55015)
Figure 27. Average limit measurement from 150 kHz to 30 MHz (I LED = 350 mA) Figure 28. Quasi-peak limit measurement from 9 kHz to 150 kHz (I LED = 350 mA)
10 Design features
10.1 Proper startup circuit design
voltage reaches 400 V, as shown in Figure 40. Figure 39. Proper startup circuit design
Figure 40. Proper startup using diode D104 and capacitor C110 Figure 41. Improper startup without using diode D104 and capacitor C110
10.2 Zero dimming design implementation
and R128 (upper limit set to 16.6 V) the transistor is opened and charges C110 and C107. voltage on capacitor C110 is not below 12.5 V during no brightness. Figure 42. Design improvement allowing zero dimming
Figure 43. Voltage on capacitor C110 and output bus voltage of 400 V
UM0670 References and related materials Doc ID 15327 Rev 2 36/43
11 References and related materials
- STMicroelectronics, EVL6562A-TM-80W, 80 W high performance transition mode PFC evaluation board, data brief; see www.st.com 2. STMicroelectronics, AN2928, Modified buck converter for LED applications, application note; see www.st.com 3. STMicroelectronics, STF9NM50N, N-channel second generation MDmesh™ power MOSFET, datasheet; see www.st.com 4. STMicroelectronics, STPSC806D, 600 V power Schottky silicon carbide diode, datasheet; see www.st.com 5. EPCOS, B66317, Ferrites and accessories E25/13/7 (EF25) core and accessories, datasheet; see www.epcos.com 6. OSRAM, LE UW E3B, OSTAR Lighting with optics, datasheet; see www.osram-os.com.
Appendix A Design calculation The aim of this section is to demonstrate how the components for the modified buck converter are calculated. Design calculation follows precisely the equations used in application note AN2928 ( Section 11: References and related materials : 2.). Therefore, please refer to this application note for more information. A.1 Design specifications for a modified buck convertor
- VIN = 400 V
- VLED = 80 V
- IAVR = 1 A
- IMAX = 1.4 A
- IMIN = 0.6 A
- f = 50 kHz
- TA = 30 °C
- TJMAX_MOSFET = 70 °C Modified buck converter working with duty cycle (output LED current is 1 A): Equation 1 Calculated OFF time for selected switching frequency of 50 kHz is: Equation 2 Now, the FOT network should be calculated. First, resistor R203 is selected: R203 = 3900 Ω Two capacitors in parallel (C204 and C206) are used for the 1 A output LED current (jumper JP3 is connected) and their size is determined using the following equation: Equation 3 Therefore the capacitors C204 and C206 have the following size: C204 = 390 pF C206 = 1.5 nF D VLED VIN tOFF 1D–() C204 C206|| tOFF
Resistor R202 limits the charging current and should be in the following range: Equation 4 Equation 5 Equation 6 A 1 kΩ resistor is chosen for R202 . Capacitor C203 should be lower than 1.25 nF , and therefore a value of 220 pF was chosen: Equation 7 C203 = 220 pF Inductor size is calculated using following equation: Equation 8 Two sense resistors are connected in parallel and their size is calculated: Equation 9 The output LED current of 1 A was precisely set by adjusting resistors R204 and R206, and therefore their optimal resistance values are 1.5 Ω and 2.2 Ω. R204 = 1.5 Ω R206 = 2.2 Ω In the next step the power MOSFET and its heat sink are calculated. The power MOSFET RMS current is derived using the following equation: VGD MAX VZCD CLAMP VF–– IZCD MAX VZCD CLAMP R203 VGD MIN V– ZCD CLAMP VF– VZCD CLAMP 15 5.7– 0.7– 0.01 5.7 ⎛⎞+
750 R 202 2326<<
C203 C204 C206||() VZCD CLAMP L VLED tOFF× R204 R206 VCS IMAX
Power MOSFET conduction loss is: Equation 11 Where the power MOSFET chosen is the STF9NM50N (see datasheet Section 11: References and related materials : 3.) and its RDS(on) for 70 °C is: Equation 12 Power MOSFET switching losses can be approximately calculated (turn OFF time was measured 120 ns - see Figure 19): Equation 13 The total power loss on the power MOSFET is 1.839 W, so the heat sink can be calculated from following equation: Equation 14 And maximum heat sink-to-ambient resistance is: Equation 15 The heat sink used in the power MOSFET on the STEVAL-ILL013V1 has a thermal resistance of 13.5 °C / W, and therefore this heat sink is optimized for this design. The last power component remaining to be calculated is the power diode. The diode conducts during the OFF time, and therefore its average current is: Equation 16 lRMS
2 D × lAVR
VIN IMAX tOFF SW f××× PTOT TJMAX MOSFET TA– RthHA TJMAX MOSFET TA– PTOT 70 30– IAVR D 1D–() IMAX IMIN+
Power loss on the STPSC806D diode is: Equation 17 where forward diode voltage was found for the average diode current 0.8 A in the datasheet (see datasheet Section 11: References and related materials : 4.). Calculated junction diode temperature without using heat sink is: Equation 18 Junction-to-case thermal resistance is available in the datasheet for the STPSC806D and case-to-ambient temperature is determined by the device package used. In this case, the TO-220 package is used and its thermal resistance is typically 60 °C / W. Calculated junction diode temperature without using a heat sink is much lower than the maximum junction temperature for the STPSC806D, and therefore this diode is suitable for the design. One of the most important things to consider is proper inductor design. The inductor size was calculated in Equation 8, but generally the inductor size by itself is not enough to ensure proper inductor design and therefore several additional equations are used for completing overall inductor construction. First, the inductor core size must be selected and for this selection it is very helpful to calculate the minimum area product using application parameters. Minimum required core area product (AP), where the flux swing is limited by core saturation is: Equation 19 where the constant is Cl = J MAX x CR x 10–4 = 420 x 0.5 x 10–4. The inductor core E25 from EPCOS was selected. The minimum core cross section is 51.5 mm2 and the winding cross section is 61 mm 2 (see datasheet Section 11: References and related materials: 5.) and the calculated product area is: Equation 20 The calculated product area is bigger than the minimum required product area, and therefore the inductor core E25 can be used. PLOSS D IAVR D VF× 0.8 0.7 0.56W=×== TJ PLOSS D RthJC RthCA+()× TA 0.56 2.4 60 +() 30 65 ° C=+×=+= APMIN Ll PEAK× lRMS× 3--- 1.6 10 3– 1.4 1××× 3--- 0.2518cm 4=== AP A N AMIN 61 51.5 0 31415cm 4,=×=×=
The number of turns for the inductor is: Equation 21 where the inductance factor Al for the E25 core and 2 mm gap is calculated: Equation 22 K1 = 70 (see datasheet Section 11: References and related materials : 5.) K2 = – 0,73 (see datasheet Section 11: References and related materials : 5.) s = E25 core air gap [mm]. The last step to complete the inductor design is to calculate the wire diameter. Maximum inductor power dissipation is: Equation 23 The wire resistance on the inductor is (copper wire with diameter of 0.28 mm is chosen): Equation 24 where average turn length I N is written in the core datasheet (see datasheet Section 11: References and related materials 5.). The power dissipation on the wire is: Equation 25 The power loss in the wire is much lower than the maximum power loss in the inductor, and so a wire with a diameter of 0.28 mm is suitable for this inductor. N L AL 54.22 10 9–× AL K1 s × 90 2 × 54.22nH=== PMAX LOSS TMAX TA– RT R ρ× l s--- ρ lN N× PWIRE Rl AVR
Table 4. Document revision history 15-May-2009 1 Initial release.