TPA721DR

TPA721_07 TI | Alldatasheet

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IN± VO± GND V DD VO+ D OR DGN PACKAGE (TOP VIEW) Audio Input Bias Control VDD 700 mW VO+ VDD BYPASS IN± VDD/2 CI RI CS CB RF SHUTDOWN VO± 8 GND From System Control

3 IN+

3.3-V and 5-V Operation The TPA721 is a bridge-tied load (BTL) audio power amplifier developed especially for low-voltage appli- Wide Power Supply Compatibility 2.5 V 5.5 V cations where internal speakers are required. Output Power for R L Ω Operating with a 3.3-V supply, the TPA721 can 700 mW at V DD BTL deliver 250-mW of continuous power into a BTL Ω 250 mW at V DD 3.3 BTL load at less than 0.6% THD+N throughout voice band frequencies. Although this device is characterized out Integrated Depop Circuitry to kHz, its operation is optimized for narrower Thermal and Short-Circuit Protection band

applications

communications. Surface-Mount Packaging The BTL configuration eliminates the need for exter- nal coupling capacitors on the output in most appli- SOIC cations, which is particularly important for small PowerPAD MSOP battery-powered equipment. This device a shutdown mode for power-sensitive a supply current of µ A during shutdown. The TPA721 is available in an 8-pin SOIC surface-mount package and the surface-mount PowerPAD MSOP, which reduces board space by 50% and height by 40%. Please be aware that an important notice concerning availability, standard warranty, and use in critical sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Copyright 1998 2004, Texas Instruments Incorporated Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

www.ti.com ABSOLUTE MAXIMUM RATINGS TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. AVAILABLE OPTIONS PACKAGED DEVICES MSOP T A SMALL OUTLINE (1) MSOP (2) SYMBOLIZATION (D) (DGN) C to C TPA721D TPA721DGN ABC (1) In the D package, the maximum output power is thermally limited to 350 mW; 700 mW peaks can be driven, as long as the RMS value is less than 350 mW. (2) The D and DGN packages are available taped and reeled. To order a taped and reeled part, add the suffix R to the part number (e.g., TPA301DR). Terminal Functions TERMINAL I/O NO. BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected to a BYPASS I 0.1- µ F to 2.2- µ F capacitor when used as an audio amplifier. GND GND is the ground connection. IN- I IN- is the inverting input. IN- is typically used as the audio input terminal. IN+ I IN+ is the noninverting input. IN+ is typically tied to the BYPASS terminal. SHUTDOWN I SHUTDOWN places the entire device in shutdown mode when held high. V DD V DD is the supply voltage terminal. V O O V O is the positive BTL output. V O O V O is the negative BTL output. over operating free-air temperature range (unless otherwise noted) (1) UNIT V DD Supply voltage V V I Input voltage 0.3 V to V DD +0.3 V Continuous total power dissipation Internally limited (see Dissipation Rating Table) T A Operating free-air temperature range C to C T J Operating junction temperature range C to 150 C T stg Storage temperature range C to 150 C Lead temperature 1,6 mm (1/16 inch) from case for seconds 260 C (1) Stresses beyond those listed under "absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

www.ti.com DISSIPATION RATING TABLE RECOMMENDED OPERATING CONDITIONS ELECTRICAL CHARACTERISTICS OPERATING CHARACTERISTICS ELECTRICAL CHARACTERISTICS TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 PACKAGE T A C DERATING FACTOR T A C T A C D 725 mW 5.8 mW/ C 464 mW 377 mW DGN 2.14 W (1) 17.1 mW/ C 1.37 W 1.11 W (1) See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD on page of that document. MIN MAX UNIT V DD Supply voltage 2.5 5.5 V V IH High-level voltage, (SHUTDOWN) 0.9 V DD V V IL Low-level voltage, (SHUTDOWN) 0.1 V DD V T A Operating free-air temperature C at specified free-air temperature, V DD 3.3 T A C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V OO Output offset voltage (measured differentially) SHUTDOWN R L Ω RF k Ω mV PSRR Power supply rejection ratio V DD 3.2 V to 3.4 V dB I DD Supply current SHUTDOWN RF k Ω 1.25 2.5 mA I DD(SD) Supply current, shutdown mode (see Figure SHUTDOWN V DD RF k Ω µ A IH SHUTDOWN, V DD 3.3 V i 3.3 V µ A IL SHUTDOWN, V DD 3.3 V i V µ A V DD 3.3 T A R L Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT P O Output power (1) THD 0.5%, See Figure 250 mW THD N Total harmonic distortion plus noise P O 250 mW, f 200 Hz to kHz, See Figure 0.55% B OM Maximum output power bandwidth Gain THD 2%, See Figure kHz B Unity-gain bandwidth Open loop, See Figure 1.4 MHz k SVR Supply ripple rejection ratio f kHz, C B µ See Figure dB V n Noise output voltage Gain C B 0.1 µ See Figure µ V(rms) (1) Output power is measured at the output terminals of the device at f kHz. at specified free-air temperature, V DD T A C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V OO Output offset voltage (measured differentially) SHUTDOWN R L Ω RF k Ω mV PSRR Power supply rejection ratio V DD 4.9 V to 5.1 V dB I DD Supply current SHUTDOWN RF k Ω 1.25 2.5 mA I DD(SD) Supply current, shutdown mode (see Figure SHUTDOWN V DD RF k Ω 100 µ A IH SHUTDOWN, V DD 5.5 V i V DD µ A IL SHUTDOWN, V DD 5.5 V i V µ A

www.ti.com OPERATING CHARACTERISTICS PARAMETER MEASUREMENT INFORMATION Audio Input Bias Control VDD VO+ VDD BYPASS IN± VDD/2 CI RI CS CB RF SHUTDOWN VO± 8 RL = 8 Ω GND V DD T A R L Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT P O Output power THD 0.5%, See Figure 700 (1) mW THD N Total harmonic distortion plus noise P O 250 mW, f 200 Hz to kHz, See Figure 0.5% B OM Maximum output power bandwidth Gain THD 2%, See Figure kHz B Unity-gain bandwidth Open loop, See Figure 1.4 MHz k SVR Supply ripple rejection ratio f kHz, C B µ See Figure dB V n Noise output voltage Gain C B 0.1 µ See Figure µ V(rms) (1) The DGN package, properly mounted, can conduct 700-mW RMS power continuously. The D package can only conduct 350-mW RMS power continuously with peaks to 700 mW. Figure BTL Mode Test Circuit

www.ti.com TYPICAL CHARACTERISTICS −50 −60 −80 −100 20 100 1k −30 −20 f − Frequency − Hz 10k 20k −10 −40 −70 −90 VDD = 5 V VDD = 3.3 V RL = 8 Ω CB = 1 µF BTL kSVR −Supply Ripple Rejection Ratio − dB VDD − Supply Voltage − V 1.8 0.8 0.6 3 4 5.5 IDD− Supply Current − mA 2.5 3.5 4.5 1.6 1.2 1.4 SHUTDOWN = 0 V RF = 10 kΩ TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 Table of Graphs FIGURE k SVR Supply ripple rejection ratio vs Frequency I DD Supply current vs Supply voltage vs Supply voltage P O Output power vs Load resistance vs Frequency 11, THD+N Total harmonic distortion plus noise vs Output power 10, 13, Open-loop gain and phase vs Frequency 15, Closed-loop gain and phase vs Frequency 17, V n Output noise voltage vs Frequency 19, P D Power dissipation vs Output power 21, SUPPLY RIPPLE REJECTION RATIO SUPPLY CURRENT vs vs FREQUENCY SUPPLY VOLTAGE Figure Figure

www.ti.com VDD − Supply Voltage − V 3 43.5 4.5 SHUTDOWN = VDD RF = 10 kΩ 5.52.5 IDD− Supply Current − Aµ VDD − Supply Voltage − V 600 400 200 2.5 3.53 4 5.5 1000 P 4.5 5 O − Output Power − mW 800 THD+N 1% f = 1 kHz BTL RL = 32 Ω RL = 8 Ω RL − Load Resistance − Ω 300 200 100 16 3224 40 64 800 P 48 56 O − Output Power − mW 400 THD+N = 1% f = 1 kHz BTL VDD = 5 V 500 600 VDD = 3.3 V 700 f − Frequency − Hz THD+N −Total Harmonic Distortion + Noise − % AV = −2 V/V VDD = 3.3 V PO = 250 mW RL = 8 Ω BTL 20 1k 10k 0.01 0.1 20k100 AV = −20 V/V AV = −10 V/V TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 TYPICAL CHARACTERISTICS (continued) SUPPLY CURRENT OUTPUT POWER vs vs SUPPLY VOLTAGE SUPPLY VOLTAGE Figure Figure OUTPUT POWER TOTAL HARMONIC DISTORTION NOISE vs vs LOAD RESISTANCE FREQUENCY Figure Figure

www.ti.com f − Frequency − Hz THD+N −Total Harmonic Distortion + Noise − % PO = 125 mW VDD = 3.3 V RL = 8 Ω AV = −2 V/V BTL 20 1k 10k 0.01 0.1 20k100 PO = 50 mW PO = 250 mW PO − Output Power − W THD+N −Total Harmonic Distortion + Noise − % 0 0.15 0.4 0.01 0.1 0.2 0.25 0.3 0.35 VDD = 3.3 V f = 1 kHz AV = −2 V/V BTL 0.05 0.1 RL = 8 Ω PO − Output Power − W THD+N −Total Harmonic Distortion + Noise − % f = 20 kHz VDD = 3.3 V RL = 8 Ω CB = 1 µF AV = −2 V/V BTL 0.01 0.1 1 0.01 0.1 f = 1 kHz f = 10 kHz f = 20 Hz f − Frequency − Hz THD+N −Total Harmonic Distortion + Noise − % AV =− 2 V/V VDD = 5 V PO = 700 mW RL = 8 Ω BTL 20 1k 10k 0.01 0.1 20k100 AV = −20 V/V AV = −10 V/V TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 TYPICAL CHARACTERISTICS (continued) TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY OUTPUT POWER Figure Figure TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs OUTPUT POWER FREQUENCY Figure 10. Figure 11.

www.ti.com f − Frequency − Hz THD+N −Total Harmonic Distortion + Noise − % PO = 700 mW VDD = 5 V RL = 8 Ω AV = −2 V/V BTL 20 1k 10k 0.01 0.1 20k100 PO = 50 mW PO = 350 mW PO − Output Power − W THD+N −Total Harmonic Distortion + Noise − % RL = 8 Ω VDD = 5 V f = 1 kHz AV = −2 V/V BTL 0.01 0.1 0.3 0.6 0.9 PO − Output Power − W THD+N −Total Harmonic Distortion + Noise − % f = 20 Hz VDD = 5 V RL = 8 Ω CB = 1 µF AV = −2 V/V BTL 0.01 0.1 1 0.01 0.1 f = 1 kHz f = 10 kHz f = 20 kHz TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 TYPICAL CHARACTERISTICS (continued) TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY OUTPUT POWER Figure 12. Figure 13. TOTAL HARMONIC DISTORTION NOISE vs OUTPUT POWER Figure 14.

www.ti.com −20 −30 f − Frequency − kHz −10 180° −180° Phase 60° −60° Open-Loop Gain − dB Phase 1 101 102 103 104 140° 100° 20° − 20° −100° −140° VDD = 3.3 V RL = Open BTL Gain −20 −30 f − Frequency − kHz −10 Gain Phase Open-Loop Gain − dB 101 102 103 104 VDD = 5 V RL = Open BTL 180° −180° 60° −60° Phase 140° 100° 20° − 20° −100° −140° TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 TYPICAL CHARACTERISTICS (continued) OPEN-LOOP GAIN AND PHASE vs FREQUENCY Figure 15. OPEN-LOOP GAIN AND PHASE vs FREQUENCY Figure 16.

www.ti.com −0.5 −1.5 f − Frequency − Hz −0.25 −0.75 −1.25 −1.75 0.5 Closed-Loop Gain − dB 0.25 0.75 130° 120° 140° Phase 150° 160° VDD = 3.3 V RL = 8 Ω PO = 250 mW BTL 170° 180° Gain Phase 101 102 103 104 105 106 −0.5 −1.5 f − Frequency − Hz −0.25 −0.75 −1.25 −1.75 0.5 Closed-Loop Gain − dB 0.25 0.75 130° 120° 140° Phase 150° 160° VDD = 5 V RL = 8 Ω PO = 700 mW BTL 170° 180° Gain Phase 101 102 103 104 105 106 TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 TYPICAL CHARACTERISTICS (continued) CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 17. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 18.

www.ti.com − Output Noise Voltage − VµVn f − Frequency − Hz 20 1k 10k 100 20k100 VO BTL VDD = 3.3 V BW = 22 Hz to 22 kHz RL = 8 Ω or 32 Ω AV = −1 V/V Vo+ − Output Noise Voltage − VµVn f − Frequency − Hz 20 1k 10k 100 20k100 VDD = 5 V BW = 22 Hz to 22 kHz RL = 8 Ω or 32 Ω AV = −1 V/V VO BTL Vo+ PD − Output Power − mW 6000 150 100 350 PD − Power Dissipation − mW 200 250

300 RL = 8 Ω

RL = 32 Ω BTL Mode VDD = 3.3 V PD − Output Power − mW 400 6000 1000 400 300 100 800 PD − Power Dissipation − mW 500 700 600 200 RL = 32 Ω 200 800 BTL Mode VDD = 5 V RL = 8 Ω TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 TYPICAL CHARACTERISTICS (continued) OUTPUT NOISE VOLTAGE OUTPUT NOISE VOLTAGE vs vs FREQUENCY FREQUENCY Figure 19. Figure 20. POWER DISSIPATION POWER DISSIPATION vs vs OUTPUT POWER OUTPUT POWER Figure 21. Figure 22.

www.ti.com APPLICATION INFORMATION BRIDGE-TIED LOAD Power V(RMS) RL V(RMS) VO(PP) 2 2 (1) RL 2x VO(PP) VO(PP) ±VO(PP) VDD VDD f(corner) 1 2RLCC (2) TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 Figure shows a linear audio power amplifier (APA) in a BTL configuration. The TPA721 BTL amplifier consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive configuration, but initially consider power to the load. The differential drive to the speaker means that as one side is slewing up, the other side is slewing down, and vice versa. This, in effect, doubles the voltage swing on the load as compared to a ground-referenced load. Plugging V O(PP) into the power equation, where voltage is squared, yields the output power from the same supply rail and load impedance (see Equation Figure 23. Bridge-Tied Load Configuration In a typical portable handheld equipment sound channel operating at 3.3 bridging raises the power into an Ω speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power, that is a 6-dB improvement, which is loudness that can be heard. In addition to increased power, there are frequency response concerns. Consider the single-supply SE configuration shown in Figure A coupling capacitor is required to block the dc offset voltage from reaching the load. These capacitors can be quite large (approximately µ F to 1000 µ so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting low-frequency performance of the system. This frequency-limiting effect is due to the high-pass filter network created with the speaker impedance and the coupling capacitance and is calculated with Equation For example, a 68- µ F capacitor with an Ω speaker would attenuate low frequencies below 293 Hz. The BTL configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency performance is then limited only by the input network and speaker response. Cost and PCB space are also minimized by eliminating the bulky coupling capacitor.

www.ti.com RL CC VO(PP) VO(PP) VDD ±3 dB fc BTL AMPLIFIER EFFICIENCY VL(RMS) VO IDD IDD(RMS) TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 APPLICATION INFORMATION (continued) Figure 24. Single-Ended Configuration and Frequency Response Increasing power to the load does carry a penalty of increased internal power dissipation. The increased dissipation is understandable considering that the BTL configuration produces the output power of a SE configuration. Internal dissipation versus output power is discussed further in the thermal considerations section. The primary cause of linear amplifier inefficiencies is voltage drop across the output stage transistors. The internal voltage drop has two components. One is the headroom or dc voltage drop that varies inversely to output power. The second component is due to the sine-wave nature of the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from V DD The internal voltage drop multiplied by the RMS value of the supply current, I DD(RMS) determines the internal power dissipation of the amplifier. An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure Figure 25. Voltage and Current Waveforms for BTL Amplifiers Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different. Keep in mind that for most of the waveform, both the push and pull transistors are not on at the same time, which supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform. The following equations are the basis for calculating amplifier efficiency.

www.ti.com IDD(RMS) 2VP RL PSUP VDD IDD(RMS) VDD 2VP RL Efficiency PL PSUP where PL VL(RMS) RL Vp 2RL VL(RMS) VP (3) Efficiency of a BTL configuration VP 4VDD

2 PLRL

(4) TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 APPLICATION INFORMATION (continued) Table employs Equation to calculate efficiencies for three different output power levels. The efficiency of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resulting in a nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper power supply design. Table Efficiency vs Output Power in 3.3-V, Ω BTL Systems INTERNAL OUTPUT POWER EFFICIENCY PEAK VOLTAGE DISSIPATION (W) (%) (V) (W) 0.125 33.6 1.41 0.26 0.25 47.6 2.00 0.29 0.375 58.3 2.45 (1) 0.28 (1) High-peak voltage values cause the THD to increase. A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the efficiency equation to utmost advantage when possible. In Equation V DD is in the denominator. This indicates that as V DD goes down, efficiency goes up.

www.ti.com APPLICATION SCHEMATICS Audio Input Bias Control VDD 700 mW VO+ VDD BYPASS IN± VDD/2 CI CS 1 µF CB 2.2 µF SHUTDOWN VO± 8 GND From System Control 10 kΩ RF 50 kΩ COMPONENT SELECTION Gain-Setting Resistors, R F and R I BTL gain 2 RF RI (5) Effective impedance RFRI RF RI (6) TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 Figure is a schematic diagram of a typical handheld audio application circuit, configured for a gain of V/V. Figure 26. TPA721 Application Circuit The following sections discuss the selection of the components used in Figure The gain for each audio input of the TPA721 is set by resistors R F and R I according to Equation for BTL mode. BTL mode operation brings about the factor in the gain equation due to the inverting amplifier mirroring the voltage swing across the load. Given that the TPA721 is a MOS amplifier, the input impedance is high; consequently, input leakage currents are not generally a concern, although noise in the circuit increases as the value of R F increases. In addition, a certain range of R F values is required for proper startup operation of the amplifier. Taken together, it is recommended that the effective impedance seen by the inverting node of the amplifier be set between k Ω and k Ω The effective impedance is calculated in Equation As an example, consider an input resistance of k Ω and a feedback resistor of k Ω The BTL gain of the amplifier would be V/V, and the effective impedance at the inverting terminal would be 8.3 k Ω which is well within the recommended range. For high-performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of R F above k Ω the amplifier tends to become unstable due to a pole formed from R F and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately pF should be placed in parallel with R F when R F is greater than k Ω This, in effect, creates a low-pass filter network with the cutoff frequency defined in Equation

www.ti.com fco(lowpass) 1 2RFCF −3 dB fc (7) Input Capacitor, C I fco(highpass) 1 2RICI −3 dB fc (8) CI 1 2RIfco (9) Power Supply Decoupling, C S TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 For example, if R F is 100 k Ω and C F is pF, then f co is 318 kHz, which is well outside of the audio range. In the typical application an input capacitor, C I is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, C I and R I form a high-pass filter with the corner frequency determined in Equation The value of C I is important to consider as it directly affects the bass (low-frequency) performance of the circuit. Consider the example where R I is k Ω and the specification calls for a flat bass response down to Hz. Equation is reconfigured as Equation In this example, C I is 0.40 µ so, one would likely choose a value in the range of 0.47 µ F to µ A further consideration for this capacitor is the leakage path from the input source through the input network I C I and the feedback resistor F to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high-gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most V DD /2, which is likely higher than the source dc level. It is important to confirm the capacitor polarity in the application. The TPA721 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µ placed as close as possible to the device V DD lead, works best. For filtering lower frequency noise signals, a larger aluminum electrolytic capacitor of µ F or greater placed near the audio power amplifier is recommended.

www.ti.com Midrail Bypass Capacitor, C B CB 250 kΩ RF RI CI (10) USING LOW-ESR CAPACITORS 5-V VERSUS 3.3-V OPERATION HEADROOM AND THERMAL CONSIDERATIONS PdB 10LogPW 10Log 700 mW ±1.5 dB TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 The midrail bypass capacitor, C B is the most critical capacitor and serves several important functions. During start-up or recovery from shutdown mode, C B determines the rate at which the amplifier starts up. The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and THD The capacitor is fed from a 250-k Ω source inside the amplifier. To keep the start-up pop as low as possible, the relationship shown in Equation should be maintained. This insures the input capacitor is fully charged before the bypass capacitor is fully charged and the amplifier starts up. As an example, consider a circuit where C B is 2.2 µ C I is 0.47 µ R F is k Ω and R I is k Ω Inserting these values into the Equation results in: 18.2 35.5 which satisfies the rule. Recommended value for bypass capacitor C B is 0.1- µ F to 2.2- µ ceramic or tantalum low-ESR, for the best THD and noise performance. Low-ESR capacitors are recommended throughout this section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor behaves like an ideal capacitor. The TPA721 operates over a supply range of 2.5 V to 5.5 This data sheet provides full specifications for 5-V and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability. The most important consideration is that of output power. Each amplifier in TPA721 can produce a maximum voltage swing of V DD This means, for 3.3-V operation, clipping starts to occur when V O(PP) 2.3 V as opposed to V O(PP) V for 5-V operation. The reduced voltage swing subsequently reduces maximum output power into an Ω load before distortion becomes significant. Operation from 3.3-V supplies, as can be shown from the efficiency formula in Equation consumes approximately two-thirds the supply power of operation from 5-V supplies for a given output-power level. Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions. A typical music CD requires dB to dB of dynamic headroom to pass the loudest portions without distortion as compared with the average power output. The TPA721 data sheet shows that when the TPA721 is operating from a 5-V supply into an Ω speaker, 700 mW peaks are available. Converting watts to dB: Subtracting the headroom restriction to obtain the average listening level without distortion yields: 1.5 dB dB 16.5 (15-dB headroom) 1.5 dB dB= 13.5 (12-dB headroom) 1.5 dB dB 10.5 (9-dB headroom) 1.5 dB dB 7.5 (6-dB headroom) 1.5 dB dB= 4.5 (3-dB headroom

www.ti.com TPA721 SLOS231E NOVEMBER 1998 REVISED JUNE 2004 Converting dB back into watts: P W PdB/10 mW (15-dB headroom) mW (12-dB headroom) mW (9-dB headroom) 175 mW (6-dB headroom) 350 mW (3-dB headroom) This is valuable information to consider when attempting to estimate the heat dissipation requirements for the amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with dB of headroom, against 12-dB and 15-dB system. Using the power dissipation curves for a 5-V, Ω system, the internal dissipation in the TPA721 and maximum ambient temperatures is shown in Table Table TPA721 Power Rating, 5-V, Ω BTL D PACKAGE DGN PACKAGE (SOIC) (MSOP) PEAK OUTPUT AVERAGE POWER POWER OUTPUT DISSIPATION MAXIMUM AMBIENT MAXIMUM AMBIENT (mW) POWER (mW) TEMPERATURE TEMPERATURE CFM) CFM) 700 700 mW 675 C 110 C 700 350 mW dB) 595 C 115 C 700 176 mW dB) 475 C 122 C 700 mW dB) 350 C 125 C 700 mW (12 dB) 225 111 C 125 C Table shows that the TPA721 can be used to its full 700-mW rating without any heat sinking in still air up to 110 C and C for the DGN package (MSOP) and D package (SOIC), respectively.

Orderable Device Status(1) Package Type Package Drawing Pins Package Qty Eco Plan(2) Lead/Ball FinishMSL Peak Temp (3) TPA721D ACTIVE SOIC D 8 75 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA721DGN ACTIVE MSOP- Power PAD DGN 8 80 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA721DGNG4 ACTIVE MSOP- Power PAD DGN 8 80 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA721DGNR ACTIVE MSOP- Power PAD DGN 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA721DGNRG4 ACTIVE MSOP- Power PAD DGN 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA721DR ACTIVE SOIC D 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA721DRG4 ACTIVE SOIC D 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA721EVM OBSOLETE TBD Call TI Call TI (1)The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2)Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontentfor the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS):TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt):This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br):TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. PACKAGE OPTION ADDENDUM www.ti.com 6-Dec-2006 Addendum-Page 1

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