TPA2006D1DRBR

TPA2006D1 TI | Alldatasheet

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88% efficiency, 75-dB PSRR, Optimized PWM Output Stage Eliminates improved RF-rectification immunity, and very small LC Output Filter total PCB footprint make the TPA2006D1 ideal for Internally Generated 250-kHz Switching cellular handsets. A fast start-up time of ms with Frequency Eliminates Capacitor and minimal pop makes the TPA2006D1 ideal for PDA Resistor applications. Improved PSRR dB) and Wide In cellular handsets, the earpiece, speaker phone, Supply Voltage (2.5 V to 5.5 Eliminates and melody ringer can each be driven by the Need for a Voltage Regulator TPA2006D1. The TPA2006D1 allows independent gain while summing signals from separate sources, Fully Differential Design Reduces RF and has a low µ V noise floor, A-weighted. Rectification and Eliminates Bypass Capacitor Improved CMRR Eliminates Two Input Coupling Capacitors Space Saving mm x mm QFN Package (DRB) Please be aware that an important notice concerning availability, standard warranty, and use in critical

applications

sheet. PRODUCTION DATA information is current as of publication date. Copyright 2006, Texas Instruments Incorporated Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

www.ti.com ORDERING INFORMATION ABSOLUTE MAXIMUM RATINGS RECOMMENDED OPERATING CONDITIONS PACKAGE DISSIPATION RATINGS TPA2006D1 SLOS498 SEPTEMBER 2006 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. T A PACKAGE (1) PART NUMBER SYMBOL C to C 8-pin QFN (DRB) TPA2006D1DRB BTQ (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI Web site at www.ti.com over operating free-air temperature range unless otherwise noted (1) TPA2006D1 In active mode 0.3 V to V V DD Supply voltage In SHUTDOWN mode 0.3 V to V V I Input voltage 0.3 V to V DD 0.3 V Continuous total power dissipation See Dissipation Rating Table T A Operating free-air temperature C to C T J Operating junction temperature C to 125 C T stg Storage temperature C to 150 C (1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. MIN NOM MAX UNIT V DD Supply voltage 2.5 5.5 V V IH High-level input voltage SHUTDOWN 1.3 V DD V V IL Low-level input voltage SHUTDOWN 0.35 V R I Input resistor Gain V/V (26 dB) k Ω V IC Common mode input voltage range V DD 2.5 5.5 CMRR dB 0.5 V DD 0.8 V T A Operating free-air temperature C T A C T A C T A C PACKAGE DERATING FACTOR (1) POWER RATING POWER RATING POWER RATING DRB 21.8 mW/ C 2.7 W 1.7 W 1.4 W (1) Derating factor measure with High K board. Submit Documentation Feedback

www.ti.com ELECTRICAL CHARACTERISTICS V V 285 k/C0087 R I 300 k/C0087 R I 315 k/C0087 R I OPERATING CHARACTERISTICS TPA2006D1 SLOS498 SEPTEMBER 2006 T A C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT Output offset voltage OS V I A V V/V, V DD 2.5 V to 5.5 V mV (measured differentially) PSRR Power supply rejection ratio V DD 2.5 V to 5.5 V dB V DD 2.5 V to 5.5 V IC V DD to 0.5 CMRR Common mode rejection ratio dB V IC V DD to V DD 0.8 V IH High-level input current V DD 5.5 V I 5.8 V 100 µ A IL Low-level input current V DD 5.5 V I 0.3 V µ A V DD 5.5 no load 3.4 4.9 I (Q) Quiescent current V DD 3.6 no load 2.8 mA V DD 2.5 no load 2.2 3.2 I (SD) Shutdown current V SHUTDOWN 0.35 V DD 2.5 V to 5.5 V 0.5 µ A V DD 2.5 V 770 Static drain-source on-state r DS(on) V DD 3.6 V 590 m Ω resistance V DD 5.5 V 500 Output impedance in SHUTDOWN V SHUTDOWN 0.35 V k Ω f (sw) Switching frequency V DD 2.5 V to 5.5 V 200 250 300 kHz Gain V DD 2.5 V to 5.5 V Resistance from shutdown to GND 300 k Ω T A Gain V/V, R L Ω (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V DD V 1.45 THD N 10%, f kHz, R L Ω V DD 3.6 V 0.73 W V DD 2.5 V 0.33 P O Output power V DD V 1.19 THD N 1%, f kHz, R L Ω V DD 3.6 V 0.59 W V DD 2.5 V 0.26 V DD P O R L Ω f kHz 0.19% Total harmonic distortion plus THD+N V DD 3.6 P O 0.5 R L Ω f kHz 0.19% noise V DD 2.5 P O 200 mW, R L Ω f kHz 0.20% f 217 Hz, V DD 3.6 Inputs ac-grounded k SVR Supply ripple rejection ratio V (RIPPLE) 200 dB with C i µ F mV PP SNR Signal-to-noise ratio V DD P O R L Ω A-weighted dB No weighting V DD 3.6 f Hz to kHz, V n Output voltage noise µ V RMS Inputs ac-grounded with C i µ F A weighting CMRR Common mode rejection ratio V DD 3.6 V IC V PP f 217 Hz dB Z I Input impedance 142 150 158 k Ω Start-up time from shutdown V DD 3.6 V ms Submit Documentation Feedback

www.ti.com FUNCTIONAL BLOCK DIAGRAM SC 150□k/c87 300□k/c87 150□k/c87 150□k/c87 150□k/c87 TPA2006D1 SLOS498 SEPTEMBER 2006 Terminal Functions TERMINAL I/O

DESCRIPTION

I Negative differential input IN+ I Positive differential input V DD I Power supply V O Positive BTL output GND O High-current ground V O Negative BTL output SHUTDOWN I Shutdown terminal (active low logic) NC No Connect, not connected internal to the device. May be left unconnected Thermal Pad O Should be soldered to a grounded thermal pad on PCB for best thermal performance Submit Documentation Feedback

www.ti.com TYPICAL CHARACTERISTICS TABLE OF GRAPHS TEST SET-UP FOR GRAPHS TP A2006D1 IN+ IN- OUT+ OUT- V DD GND CI CI RI RI Measurement Output

1 F/c109+

P D Power dissipation vs Output power Supply current vs Output power I (Q) Quiescent current vs Supply voltage I (SD) Shutdown current vs Shutdown voltage vs Supply voltage P O Output power vs Load resistance vs Output power THD+N Total harmonic distortion plus noise vs Frequency 10, 11, vs Common-mode input voltage K SVR Supply ripple rejection ratio vs Frequency 14, vs Time GSM power supply rejection vs Frequency K SVR Supply ripple rejection ratio vs Common-mode input voltage vs Frequency CMRR Common-mode rejection ratio vs Common-mode input voltage C I is shorted for any common-mode input voltage measurement. A 33- µ H inductor is placed in series with the load resistor to emulate a small speaker for efficiency measurements. The 30-kHz low-pass filter is required even if the analyzer has an internal low-pass filter. An RC low-pass filter (100 Ω nF) is used on each output for the data sheet graphs. Submit Documentation Feedback

www.ti.com VDD L =□2.5□V, Class-AB, V =□5□V, R =□8 DD L /c87 P -□Output□Power□-□WO Efficiency□-□% VDD L =□5□V, 100 150 200 250 300 PO -□Output□Power□-□W V =□2.5□V, R =□8 ,□33 H DD L /c87 /c109 V =□3.6□V, R =□8 ,□33 H DD L /c87 /c109 V =□5□V, R =□8 ,□33 H DD L /c87 /c109 Supply□Current□-□mA 0.1 0.2 0.3 0.4 0.5 0.6 0.7 P D Power□Dissipation□-□W PO -□Output□Power□-□W Class-AB, V =□5□V,□R =□8DD L /c87 Class-AB, V =□3.6□V, R =□8 DD L /c87 V =□3.6□V, R =□8 ,□33 H DD L /c87 /c109 V =□5□V, R =□8 ,□33 H DD L /c87 /c109 0.5 1.5 Shutdown Voltage − V − Shutdown Current − I(SD) Aµ VDD = 5 V VDD = 3.6 V VDD = 2.5 V 0.2 0.4 0.6 0.8 1.2 1.4 1.6 8 12 16 20 24 28 32 VDD =□5□V V =□3.6□VDD V =□2.5□VDD R -□Load□Resistance□-L /c87 P -□Output□Power□-□W O f□=□1□kHz THD+N□=□10% Gain□=□2□V/V 2.2 2.4 2.6 2.8 3.2 3.4 3.6 3.8 2.5 3 3.5 4 4.5 5 5.5 I(Q) − Quiescent□Current − mA V − VDD − Supply□Voltage No□Load R =□8 ,□33 HL /c87 /c109 2.5 3 3.5 4 4.5 5 V -□Supply□VDD oltage□-□V P -□Output□Power□-□W O R =□8 f□=□1□kHz L /c87 Gain□=□2□V/V THD+N□=□1% THD+N□=□10% 0.2 0.4 0.6 0.8 1.2 1.4 1.6 0.1 0.001 0.01 1□k 10□k Power□Output − W THD+N − Total□Harmonic□Distortion□+□Noise − %0.1□k R =□8 f□=□1□kHz L /c87 2.5□V 3.6□V 5□V10 0.2 0.4 0.6 0.8 1.2 1.4 8 12 16 20 28 R -□Load□Resistance□-L /c87 P -□Output□Power□-□W O 3224 f□=□1□kHz THD+N□=□1% Gain□=□2□V/V V =□2.5□VDD V =□3.6□VDD VDD =□5□V TPA2006D1 SLOS498 SEPTEMBER 2006 EFFICIENCY POWER DISSIPATION SUPPLY CURRENT vs vs vs OUTPUT POWER OUTPUT POWER OUTPUT POWER Figure Figure Figure QUIESCENT CURRENT SUPPLY CURRENT OUTPUT POWER vs vs vs SUPPLY VOLTAGE SHUTDOWN VOLTAGE LOAD RESISTANCE Figure Figure Figure TOTAL HARMONIC DISTORTION OUTPUT POWER OUTPUT POWER NOISE vs vs vs LOAD RESISTANCE SUPPLY VOLTAGE OUTPUT POWER Figure Figure Figure Submit Documentation Feedback

www.ti.com 0.01 0.1 20 100 10□k 20□k f − Frequency − Hz THD+N − Total□Harmonic□Distortion□+□Noise − % P =□1□WO P =□0.25□WO P =□0.5□WO V =□5□VDD R =□8L /c87 1□k 0.001 0.1 20 100 10□k 20□k f − Frequency − Hz THD+N − Total□Harmonic□Distortion□+□Noise − % 0.01 1□k V =□3.6□VDD R =□8L /c87 P =□0.25□WO P =□0.5□WO P =□0.125□WO 0.001 0.1 20 100 10□k 20□k f − Frequency − Hz THD+N − Total□Harmonic□Distortion□+□Noise − % 0.01 1□k V =□2.5□VDD R =□8L /c87 P =□0.2□WO P =□0.075□WO P =□0.015□WO −90 −80 −70 −60 −50 −40 −30 20 100 1□k 10□k VDD =□5□V VDD =□2.5□V f − Frequency − Hz Sopply□Ripple□Rejection□Ratio − dB Inputs□floating R =□8L /c87 20□k VDD =□3.6□V −90 −80 −70 −60 −50 −40 −30 20 100 1□k 20□k f − Frequency − Hz Supply□Ripple□Rejection□Ratio − dB VDD =□2.□5□V VDD =□3.6□V VDD =□5□V Inputs□ac-grounded C I =□2 F/c109 RL =□8 /c87 Gain□=□2□V/V 10□k 0.1 0 0.5 1 1.5 2 2.5 f = 1 kHz P O = 200 mW VIC − Common Mode Input Voltage − V THD+N − Total Harmonic Distortion + Noise − % 3 3.5 4 4.5 5 VDD = 2.5 V VDD = 5 V VDD = 3.6 V C1 − High 3.6 V C1 − Amp 512 mV C1 − Duty 12% t − Time − 2 ms/div V DD 200 mV/div VOUT 20 mV/div −150 −100 −50 0 400 800 1200 1600 2000 −150 −100 −50 f − Frequency − Hz − Output Voltage − dBVVO − Supply Voltage − dBVV DD VDD Shown in Figure 22 C I = 2 µF, Inputs ac-grounded Gain = 2V/V TPA2006D1 SLOS498 SEPTEMBER 2006 TOTAL HARMONIC DISTORTION TOTAL HARMONIC DISTORTION TOTAL HARMONIC DISTORTION NOISE NOISE NOISE vs vs vs FREQUENCY FREQUENCY FREQUENCY Figure 10. Figure 11. Figure 12. TOTAL HARMONIC DISTORTION NOISE SUPPLY RIPPLE REJECTION RATIO SUPPLY RIPPLE REJECTION RATIO vs vs vs COMMON MODE INPUT VOLTAGE FREQUENCY FREQUENCY Figure 13. Figure 14. Figure 15. GSM POWER SUPPLY REJECTION GSM POWER SUPPLY REJECTION vs vs TIME FREQUENCY Figure 16. Figure 17. Submit Documentation Feedback

www.ti.com −100 −90 −80 −70 −60 −50 −40 −30 −20 −10 0 1 2 3 4 5 VIC − Common Mode Input Voltage − V CMRR − Common Mode Rejection Ratio − dB VDD = 5 V, Gain = 2 VDD = 2.5 V VDD = 3.6 V −75 −70 −65 −60 −55 −50 20 100 1 k 20 k VDD = 3.6 V f − Frequency − Hz CMRR − Common Mode Rejection Ratio − dB VIC = 200 mVPP R L = 8 Ω Gain = 2 V/V 10 k −80 −70 −60 −50 −40 −30 −20 −10 DC Common Mode V oltage − V Sopply Ripple Rejection Ratio − dB VDD = 2. 5 V VDD = 3.6 V VDD = 5 V TPA2006D1 SLOS498 SEPTEMBER 2006 SUPPLY RIPPLE REJECTION RATIO COMMON-MODE REJECTION RATIO COMMON-MODE REJECTION RATIO vs vs vs DC COMMON MODE VOLTAGE FREQUENCY COMMON-MODE INPUT VOLTAGE Figure 18. Figure 19. Figure 20. Submit Documentation Feedback

www.ti.com APPLICATION INFORMATION FULLY DIFFERENTIAL AMPLIFIER Advantages of Fully Differential Amplifiers COMPONENT SELECTION TPA2006D1 SLOS498 SEPTEMBER 2006 The TPA2006D1 is a fully differential amplifier with differential inputs and outputs. The fully differential amplifier consists of a differential amplifier and a common-mode amplifier. The differential amplifier ensures that the amplifier outputs a differential voltage on the output that is equal to the differential input times the gain. The common-mode feedback ensures that the common-mode voltage at the output is biased around V DD regardless of the common-mode voltage at the input. The fully differential TPA2006D1 can still be used with a single-ended input; however, the TPA2006D1 should be used with differential inputs when in a noisy environment, like a wireless handset, to ensure maximum noise rejection. Input-coupling capacitors not required: The fully differential amplifier allows the inputs to be biased at voltage other than mid-supply. For example, if a codec has a mid-supply lower than the mid-supply of the TPA2006D1, the common-mode feedback circuit will adjust, and the TPA2006D1 outputs will still be biased at mid-supply of the TPA2006D1. The inputs of the TPA2006D1 can be biased from 0.5 V to V DD 0.8 If the inputs are biased outside of that range, input-coupling capacitors are required. Mid-supply bypass capacitor, C (BYPASS) not required: The fully differential amplifier does not require a bypass capacitor. This is because any shift in the midsupply affects both positive and negative channels equally and cancels at the differential output. Better RF-immunity: GSM handsets save power by turning on and shutting off the RF transmitter at a rate of 217 Hz. The transmitted signal is picked-up on input and output traces. The fully differential amplifier cancels the signal much better than the typical audio amplifier. Figure shows the TPA2006D1 typical schematic with differential inputs and Figure shows the TPA2006D1 with differential inputs and input capacitors, and Figure shows the TPA2006D1 with single-ended inputs. Differential inputs should be used whenever possible because the single-ended inputs are much more susceptible to noise. Table Typical Component Values REF DES VALUE EIA SIZE MANUFACTURER PART NUMBER R I 150 k Ω 0.5%) 0402 Panasonic ERJ2RHD154V C S µ F (+22%, -80%) 0402 Murata GRP155F50J105Z C I (1) 3.3 nF 10%) 0201 Murata GRP033B10J332K (1) C I is only needed for single-ended input or if V ICM is not between 0.5 V and V DD 0.8 C I 3.3 nF (with R I 150 k Ω gives a high-pass corner frequency of 321 Hz. Submit Documentation Feedback

www.ti.com Input Resistors I Gain /C00432 x 150 k/C0087 R I /C0466V V /C0467 (1) Decoupling Capacitor S Input Capacitors I fc /C0043 1 /C04662/C0112R IC I/C0467 (2) C I/C0043 1 /C04662/C0112R Ifc/C0467 (3) TPA2006D1 SLOS498 SEPTEMBER 2006 The input resistors I set the gain of the amplifier according to Equation Resistor matching is very important in fully differential amplifiers. The balance of the output on the reference voltage depends on matched ratios of the resistors. CMRR, PSRR, and cancellation of the second harmonic distortion diminish if resistor mismatch occurs. Therefore, it is recommended to use tolerance resistors or better to keep the performance optimized. Matching is more important than overall tolerance. Resistor arrays with matching can be used with a tolerance greater than 1%. Place the input resistors very close to the TPA2006D1 to limit noise injection on the high-impedance nodes. For optimal performance the gain should be set to V/V or lower. Lower gain allows the TPA2006D1 to operate at its best, and keeps a high voltage at the input making the inputs less susceptible to noise. The TPA2006D1 is a high-performance class-D audio amplifier that requires adequate power supply decoupling to ensure the efficiency is high and total harmonic distortion (THD) is low. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically µ placed as close as possible to the device V DD lead works best. Placing this decoupling capacitor close to the TPA2006D1 is very important for the efficiency of the class-D amplifier, because any resistance or inductance in the trace between the device and the capacitor can cause a loss in efficiency. For filtering lower-frequency noise signals, a µ F or greater capacitor placed near the audio power amplifier would also help, but it is not required in most device. The TPA2006D1 does not require input coupling capacitors if the design uses a differential source that is biased from 0.5 V to V DD 0.8 V (shown in Figure If the input signal is not biased within the recommended common-mode input range, if needing to use the input as a high pass filter (shown in Figure or if using a single-ended source (shown in Figure input coupling capacitors are required. The input capacitors and input resistors form a high-pass filter with the corner frequency, f c determined in Equation The value of the input capacitor is important to consider as it directly affects the bass (low frequency) performance of the circuit. Speakers in wireless phones cannot usually respond well to low frequencies, so the corner frequency can be set to block low frequencies in this application. Equation is reconfigured to solve for the input coupling capacitance. If the corner frequency is within the audio band, the capacitors should have a tolerance of 10% or better, because any mismatch in capacitance causes an impedance mismatch at the corner frequency and below. For a flat low-frequency response, use large input coupling capacitors µ F). However, in a GSM phone the ground signal is fluctuating at 217 Hz, but the signal from the codec does not have the same 217-Hz fluctuation. The difference between the two signals is amplified, sent to the speaker, and heard as a 217-Hz hum. Submit Documentation Feedback

www.ti.com IN- IN+ PWM H- Bridge VO+ VO- Internal Oscillator CS To□Battery VDD GNDBias Circuitry RI RI Differential Input TPA2006D1 Filter-Free□Class□D SHUTDOWN IN- IN+ PWM H- Bridge VO+ VO- Internal Oscillator CS To□Battery VDD GND Bias Circuitry RI RI Differential Input TPA2006D1 Filter-Free□Class□D SHUTDOWN CI CI TPA2006D1 Filter-Free□Class□D SHUTDOWN TPA2006D1 SLOS498 SEPTEMBER 2006 Figure 21. Typical TPA2006D1 Application Schematic With Differential Input for a Wireless Phone Figure 22. TPA2006D1 Application Schematic With Differential Input and Input Capacitors Figure 23. TPA2006D1 Application Schematic With Single-Ended Input Submit Documentation Feedback

www.ti.com SUMMING INPUT SIGNALS WITH THE TPA2006D1 Summing Two Differential Input Signals Gain 1/C0043 V O V I1 /C00432 x 150 k/C0087 R I1 /C0466V V /C0467 (4) Gain 2/C0043 V O V I2 /C00432 x 150 k/C0087 R I2 /C0466V V /C0467 (5) Filter-Free□Class□D SHUTDOWN TPA2006D1 SLOS498 SEPTEMBER 2006 Most wireless phones or PDAs need to sum signals at the audio power amplifier or just have two signal sources that need separate gain. The TPA2006D1 makes it easy to sum signals or use separate signal sources with different gains. Many phones now use the same speaker for the earpiece and ringer, where the wireless phone would require a much lower gain for the phone earpiece than for the ringer. PDAs and phones that have stereo headphones require summing of the right and left channels to output the stereo signal to the mono speaker. Two extra resistors are needed for summing differential signals total of components). The gain for each input source can be set independently (see Equation and Equation and Figure If summing left and right inputs with a gain of V/V, use R R 300 k Ω Figure 24. Application Schematic With TPA2006D1 Summing Two Differential Inputs Submit Documentation Feedback

www.ti.com Summing a Differential Input Signal and a Single-Ended Input Signal Gain 1/C0043 V O V I1 /C00432 x 150 k/C0087 R I1 /C0466V V /C0467 (6) Gain 2/C0043 V O V I2 /C00432 x 150 k/C0087 R I2 /C0466V V /C0467 (7) C I2/C0043 1 /C04662/C0112R I2fc2/C0467 (8) C I2/C0117 1 /C04662/C0112150k/C008720Hz /C0467 (9) C >□□53□nFI2 (10) IN- IN+ PWM H- Bridge VO+ VO- Internal Oscillator C S To Battery VDD GNDBias Circuitry R I2 R I2 Differential Input 1 Filter-Free Class D SHUTDOWN R I1 R I1 Single-Ended Input 2 C I2 C I2 TPA2006D1 SLOS498 SEPTEMBER 2006 Figure shows how to sum a differential input signal and a single-ended input signal. Ground noise can couple in through IN+ with this method. It is better to use differential inputs. The corner frequency of the single-ended input is set by C shown in Equation To assure that each input is balanced, the single-ended input must be driven by a low-impedance source even if the input is not in use If summing a ring tone and a phone signal, the phone signal should use a differential input signal while the ring tone might be limited to a single-ended signal. The high pass corner frequency of the single-ended input is set by C If the desired corner frequency is less than Hz: Figure 25. Application Schematic With TPA2006D1 Summing Differential Input and Single-Ended Input Signals Submit Documentation Feedback

www.ti.com Summing Two Single-Ended Input Signals Gain 1/C0043 V O V I1 /C00432 x 150 k/C0087 R I1 /C0466V V /C0467 (11) Gain 2/C0043 V O V I2 /C00432 x 150 k/C0087 R I2 /C0466V V /C0467 (12) C I1/C0043 1 /C04662/C0112R I1fc1/C0467 (13) C I2/C0043 1 /C04662/C0112R I2fc2/C0467 (14) C P /C0043C I1 /C0041C I2 (15) R P /C0043 R I1 /C0032R I2 /C0466R I1 /C0041R I2/C0467 (16) IN- IN+ PWM H- Bridge VO+ VO- Internal Oscillator C S To Battery VDD GND Bias Circuitry R I2 R P Filter-Free Class D SHUTDOWN R I1 Single-Ended Input 2 C I2 C P Single-Ended Input 1 C I1 Component Location TPA2006D1 SLOS498 SEPTEMBER 2006 Four resistors and three capacitors are needed for summing single-ended input signals. The gain and corner frequencies and f for each input source can be set independently (see Equation through Equation and Figure Resistor, R P and capacitor, C P are needed on the IN+ terminal to match the impedance on the IN- terminal. The single-ended inputs must be driven by low impedance sources even if one of the inputs is not outputting an ac signal. Figure 26. Application Schematic With TPA2006D1 Summing Two Single-Ended Inputs Place all the external components very close to the TPA2006D1. The input resistors need to be very close to the TPA2006D1 input pins so noise does not couple on the high impedance nodes between the input resistors and the input amplifier of the TPA2006D1. Placing the decoupling capacitor, C S close to the TPA2006D1 is important for the efficiency of the class-D amplifier. Any resistance or inductance in the trace between the device and the capacitor can cause a loss in efficiency. Submit Documentation Feedback

www.ti.com EFFICIENCY AND THERMAL INFORMATION Derating Factor 0.0218 =□□45.9 C/W o /c113JA = (17) TA J JA DmaxMax□□= T Max P =□□125 45.9(0.2)□□=□□115.8 C/c45 /c113 /c45 o (18) ELIMINATING THE OUTPUT FILTER WITH THE TPA2006D1 Effect on Audio Traditional Class-D Modulation Scheme 0 V -5 V +5 V Current OUT+ Differential Voltage Across Load OUT- TPA2006D1 SLOS498 SEPTEMBER 2006 The maximum ambient temperature depends on the heat-sinking ability of the PCB system. The derating factor for the DRB package is shown in the dissipation rating table. Converting this to θ JA Given θ JA of 45.9 C/W, the maximum allowable junction temperature of 125 and the maximum internal dissipation of 0.2 W (Po=1.45 Ω load, 5-V supply, from Figure the maximum ambient temperature can be calculated with the following equation. Equation shows that the calculated maximum ambient temperature is 115.8 C at maximum power dissipation with a 5-V supply and Ω a load, see Figure The TPA2006D1 is designed with thermal protection that turns the device off when the junction temperature surpasses 150 C to prevent damage to the IC. This section focuses on why the user can eliminate the output filter with the TPA2006D1. The class-D amplifier outputs a pulse-width modulated (PWM) square wave, which is the sum of the switching waveform and the amplified input audio signal. The human ear acts as a band-pass filter such that only the frequencies between approximately Hz and kHz are passed. The switching frequency components are much greater than kHz, so the only signal heard is the amplified input audio signal. The traditional class-D modulation scheme, which is used in the TPA005Dxx family, has a differential output where each output is 180 degrees out of phase and changes from ground to the supply voltage, V DD Therefore, the differential pre-filtered output varies between positive and negative V DD where filtered 50% duty cycle yields volts across the load. The traditional class-D modulation scheme with voltage and current waveforms is shown in Figure Note that even at an average of volts across the load (50% duty cycle), the current to the load is high causing a high loss and thus causing a high supply current. Figure 27. Traditional Class-D Modulation Scheme's Output Voltage and Current Waveforms Into an Inductive Load With no Input Submit Documentation Feedback

www.ti.com TPA2006D1 Modulation Scheme 0 V -5 V +5 V Current OUT+ OUT- Differential Voltage Across Load 0 V -5 V +5 V Current OUT+ OUT- Differential Voltage Across Load Output = 0 V Output > 0 V Efficiency: Why You Must Use a Filter With the Traditional Class-D Modulation Scheme TPA2006D1 SLOS498 SEPTEMBER 2006 The TPA2006D1 uses a modulation scheme that still has each output switching from to the supply voltage. However, OUT+ and OUT- are now in phase with each other with no input. The duty cycle of OUT+ is greater than 50% and OUT- is less than 50% for positive voltages. The duty cycle of OUT+ is less than 50% and OUT- is greater than 50% for negative voltages. The voltage across the load sits at volts throughout most of the switching period greatly reducing the switching current, which reduces any I R losses in the load. Figure 28. The TPA2006D1 Output Voltage and Current Waveforms Into an Inductive Load The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform results in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple current is large for the traditional modulation scheme because the ripple current is proportional to voltage multiplied by the time at that voltage. The differential voltage swing is V DD and the time at each voltage is half the period for the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from each half cycle for the next half cycle, while any resistance causes power dissipation. The speaker is both resistive and reactive, whereas an LC filter is almost purely reactive. The TPA2006D1 modulation scheme has very little loss in the load without a filter because the pulses are very short and the change in voltage is V DD instead of V DD As the output power increases, the pulses widen making the ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for most needed. An LC filter with a cutoff frequency less than the class-D switching frequency allows the switching current to flow through the filter instead of the load. The filter has less resistance than the speaker that results in less power dissipated, which increases efficiency. Submit Documentation Feedback

www.ti.com Effects of Applying a Square Wave Into a Speaker P SPKR /C0043P SUP ±P SUP THEORETICAL (at max output power) (19) P SPKR /C0043 P SUP P OUT P SUP THEORETICAL P OUT (at max output power) (20) P SPKR /C0043P OUT /C04661 /C0104MEASURED /C0042 1 /C0104THEORETICAL /C0467(at max output power) (21) /C0104THEORETICAL /C0043 R L R L /C00412rDS(on) (at max output power) (22) TPA2006D1 SLOS498 SEPTEMBER 2006 If the amplitude of a square wave is high enough and the frequency of the square wave is within the bandwidth of the speaker, a square wave could cause the voice coil to jump out of the air gap and/or scar the voice coil. A 250-kHz switching frequency, however, is not significant because the speaker cone movement is proportional to 1/f for frequencies beyond the audio band. Therefore, the amount of cone movement at the switching frequency is very small. However, damage could occur to the speaker if the voice coil is not designed to handle the additional power. To size the speaker for added power, the ripple current dissipated in the load needs to be calculated by subtracting the theoretical supplied power, P SUP THEORETICAL from the actual supply power, P SUP at maximum output power, P OUT The switching power dissipated in the speaker is the inverse of the measured efficiency, η MEASURED minus the theoretical efficiency, η THEORETICAL The maximum efficiency of the TPA2006D1 with a 3.6 V supply and an Ω load is 86% from Equation Using equation Equation with the efficiency at maximum power (84%), we see that there is an additional mW dissipated in the speaker. The added power dissipated in the speaker is not an issue as long as it is taken into account when choosing the speaker. Submit Documentation Feedback

www.ti.com When to Use an Output Filter 1□nF Ferrite Chip□Bead VO- Ferrite Chip□Bead 1□nF VO+

0.1 F/c109

33 H/c109

33 H/c109VO-

0.1 F/c1090.47 F/c109 TPA2006D1 SLOS498 SEPTEMBER 2006 Design the TPA2006D1 without an output filter if the traces from amplifier to speaker are short. The TPA2006D1 passed FCC and CE radiated emissions with no shielding with speaker trace wires 100 mm long or less. Wireless handsets and PDAs are great a filter. A ferrite bead filter can often be used if the design is failing radiated emissions without an LC filter, and the frequency sensitive circuit is greater than MHz. This is good for circuits that just have to pass FCC and CE because FCC and CE only test radiated emissions greater than MHz. If choosing a ferrite bead, choose one with high impedance at high frequencies, but very low impedance at low frequencies. Use an LC output filter if there are low frequency MHz) EMI sensitive circuits and/or there are long leads from amplifier to speaker. Figure and Figure show typical ferrite bead and LC output filters. Figure 29. Typical Ferrite Chip Bead Filter (Chip bead example: NEC/Tokin: N2012ZPS121) Figure 30. Typical LC Output Filter, Cutoff Frequency of kHz Submit Documentation Feedback

Orderable Device Status(1) Package Type Package Drawing Pins Package Qty Eco Plan(2) Lead/Ball FinishMSL Peak Temp (3) TPA2006D1DRBR ACTIVE SON DRB 8 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA2006D1DRBRG4 ACTIVE SON DRB 8 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA2006D1DRBT ACTIVE SON DRB 8 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA2006D1DRBTG4 ACTIVE SON DRB 8 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR (1)The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2)Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontentfor the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS):TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt):This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br):TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. PACKAGE OPTION ADDENDUM www.ti.com 2-Oct-2006 Addendum-Page 1

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