TPA2005D1DRBQ1

TPA2005D1-Q1 TI | Alldatasheet

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84% efficiency, -71-dB PSRR at Internally Generated 250-kHz Switching 217 Hz, improved RF-rectification immunity, and Frequency Eliminates Capacitor and 15-mm total PCB area make the TPA2005D1 ideal Resistor for cellular handsets. A fast start-up time of ms with Improved PSRR (-71 dB at 217 Hz) and minimal pop makes the TPA2005D1 ideal for PDA Wide Supply Voltage (2.5 V to 5.5 applications. Eliminates Need for a Voltage Regulator In cellular handsets, the earpiece, speaker phone, Fully Differential Design Reduces RF and melody ringer can each be driven by the Rectification and Eliminates Bypass TPA2005D1. The device allows independent gain Capacitor control by summing the signals from each function, while minimizing noise to only µ V RMS Improved CMRR Eliminates Two Input Coupling Capacitors (1) Contact factory for details. Q100 qualification data available on request. Please be aware that an important notice concerning availability, standard warranty, and use in critical

applications

sheet. MicroStar Junior, PowerPAD are trademarks of Texas Instruments. PRODUCTION DATA information is current as of publication date. Copyright 2005, Texas Instruments Incorporated Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

www.ti.com ABSOLUTE MAXIMUM RATINGS RECOMMENDED OPERATING CONDITIONS DISSIPATION RATINGS TPA2005D1-Q1 SLOS474 AUGUST 2005 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. ORDERING INFORMATION T A PACKAGE PART NUMBER SYMBOL MicroStar Junior (GQY) TPA2005D1GQYR (1) PREVIEW MicroStar Junior (ZQY) (2) TPA2005D1ZQYR (1) PREVIEW -40 C to C 8-pin QFN (DRB) TPA2005D1DRBR (1) BIQ 8-pin MSOP (DGN) TPA2005D1DGN(R) PREVIEW (1) The GQY, ZQY, and DRB packages are only available taped and reeled. An R at the end of the part number indicates the devices are taped and reeled. (2) The GQY is the standard MicroStar Junior package. The ZQY is lead-free option, and is qualified for 260 lead-free assembly. over operating free-air temperature range unless otherwise noted (1) UNIT In active mode -0.3 V to V V DD Supply voltage (2) In SHUTDOWN mode -0.3 V to V V I Input voltage -0.3 V to V DD 0.3 V Continuous total power dissipation See Dissipation Rating Table T A Operating free-air temperature -40 C to C T J Operating junction temperature -40 C to 150 C T stg Storage temperature -65 C to C Lead temperature 1,6 mm (1/16 inch) from case for seconds 260 C (1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. (2) For the MSOP (DGN) package option, the maximum V DD should be limited to V if short-circuit protection is desired. MIN NOM MAX UNIT V DD Supply voltage 2.5 5.5 V V IH High-level input voltage SHUTDOWN V DD V V IL Low-level input voltage SHUTDOWN 0.7 V R I Input resistor Gain V/V (26 dB) k Ω V IC Common-mode input voltage range V DD 2.5 5.5 CMRR -49 dB 0.5 V DD -0.8 V T A Operating free-air temperature -40 C DERATING T A C T A C T A C PACKAGE FACTOR POWER RATING POWER RATING POWER RATING GQY, ZQY mW/ C W 1.28 W 1.04 W DRB 21.8 mW/ C 2.7 W 1.7 W 1.4 W DGN 17.1 mW/ C 2.13 W 1.36 W 1.11 W

www.ti.com ELECTRICAL CHARACTERISTICS V V 2 142 k RI 2 158 k RI 2 150 k RI OPERATING CHARACTERISTICS TPA2005D1-Q1 SLOS474 AUGUST 2005 T A -40 C to C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT Output offset voltage OS V I A V V/V, V DD 2.5 V to 5.5 V mV (measured differentially) PSRR Power-supply rejection ratio V DD 2.5 V to 5.5 V -75 -55 dB V DD 2.5 V to 5.5 T A C -68 -49 CMRR Common-mode rejection ratio V IC V DD to 0.5 dB T A -40 C to C -35 V IC V DD to V DD 0.8 V IH High-level input current V DD 5.5 V I 5.8 V µ A IL Low-level input current V DD 5.5 V I 0.3 V µ A V DD 5.5 no load 3.4 4.5 I (Q) Quiescent current V DD 3.6 no load 2.8 mA V DD 2.5 no load 2.2 3.2 I (SD) Shutdown current V (SHUTDOWN) 0.8 V DD 2.5 V to 5.5 V 0.5 µ A V DD 2.5 V 770 Static drain-source r DS(on) V DD 3.6 V 590 m Ω on-state resistance V DD 5.5 V 500 Output impedance in V (SHUTDOWN) 0.8 V k Ω SHUTDOWN f (sw) Switching frequency V DD 2.5 V to 5.5 V 200 250 300 kHz Gain T A Gain V/V, R L Ω (unless otherwise noted) PARAMETER TEST CONDITIOINS MIN TYP MAX UNIT V DD V 1.18 THD 1%, f kHz, V DD 3.6 V 0.58 W R L Ω V DD 2.5 V 0.26 P O Output power V DD V 1.45 THD 10%, f kHz, V DD 3.6 V 0.75 W R L Ω V DD 2.5 V 0.35 P O f kHz, R L Ω V DD V 0.18% Total harmonic distortion plus THD+N P O 0.5 f kHz, R L Ω V DD 3.6 V 0.19% noise P O 200 mW, f kHz, R L Ω V DD 2.5 V 0.20% f 217 Hz, V (RIPPLE) 200 mV pp k SVR Supply ripple rejection ratio V DD 3.6 V -71 dB Inputs ac-grounded with C i µ F SNR Signal-to-noise ratio P O R L Ω V DD V dB No weighting V DD 3.6 f Hz to kHz, V n Output voltage noise µ V RMS Inputs ac-grounded with C i µ F A weighting CMRR Common-mode rejection ratio V IC V pp f 217 Hz V DD 3.6 V -63 dB Z I Input impedance 142 150 158 k Ω Start-up time from shutdown V DD 3.6 V ms

www.ti.com PIN ASSIGNMENTS (A1) (B1) (A4) (C4) (D4) (SIDE VIEW) MicroStar Junior (GQY) PACKAGE (TOP VIEW) NC VDD SHUTDOWN IN+ IN− VO− VDD VO+ (C1) (D1) (B4) GND 8SHUTDOWN NC IN+ IN− VO− GND VDD VO+ 8-PIN QFN (DRB) PACKAGE (TOP VIEW) NC − No internal connection VO− GND VDD VO+ SHUTDOWN NC IN+ IN− 8-PIN MSOP (DGN) PACKAGE (TOP VIEW) 150 kΩ 150 kΩ Deglitch Logic Deglitch Logic Gate Drive Gate Drive VDD Short Circuit Detect Startup & Thermal Protection Logic Ramp Generator Biases and References TTL Input Buffer SD Gain = 2 V/V B4, C4 VDD VO− VO+ GND D1IN− C1IN+ A1SHUTDOWN ² A2, A3, B3, C2, C3, D2, D3 (terminal labels for MicroStar Junior /char????????package) TPA2005D1-Q1 SLOS474 AUGUST 2005 The shaded terminals are used for electrical and thermal connections to the ground plane. All of the shaded terminals must be electrically connected to ground. No connect (NC) terminals still need a pad and trace. The thermal pad of the DRB and DGN packages must be electrically and thermally connected to a ground plane. Terminal Functions TERMINAL I/O

DESCRIPTION

ZQY, GQY DRB, DGN IN- I Negative differential input IN+ I Positive differential input V DD B4, I Power supply V O Positive BTL output A2, A3, B3, GND C2, C3, D2, I High-current ground V O Negative BTL output SHUTDOWN I Shutdown terminal (active low logic) NC No internal connection Thermal Pad Must be soldered to a grounded pad on the PCB. FUNCTIONAL BLOCK DIAGRAM

www.ti.com TYPICAL CHARACTERISTICS Table of Graphs TEST SET-UP FOR GRAPHS TPA2005D1 IN+ IN− OUT+ OUT− VDD GND CI CI RI RI Measurement Output 1 µF+ VDD Load 30 kHz Low Pass Filter Measurement Input TPA2005D1-Q1 SLOS474 AUGUST 2005 FIGURE Efficiency vs Output power P D Power dissipation vs Output power Supply current vs Output power I (Q) Quiescent current vs Supply voltage I (SD) Shutdown current vs Shutdown voltage vs Supply voltage P O Output power vs Load resistance vs Output power 11, THD+N Total harmonic distortion plus noise vs Frequency 13, 14, 15, vs Common-mode input voltage vs Frequency 18, 19, k SVR Supply-voltage rejection ratio vs Common-mode input voltage vs Time GSM power-supply rejection vs Frequency vs Frequency CMRR Common-mode rejection ratio vs Common-mode input voltage C I was shorted for any common-mode input voltage measurement. A 33- µ H inductor was placed in series with the load resistor to emulate a small speaker for efficiency measurements. The 30-kHz low-pass filter is required, even if the analyzer has a low-pass filter. An RC filter (100 Ω nF) is used on each output for the data sheet graphs.

www.ti.com VDD = 2.5 V, RL= 8 Ω , 33 µH VDD = 5 V, RL = 8 Ω , 33 µH Class-AB, V DD = 5 V, RL = 8 Ω PO - Output Power - W Efficiency - % 0.1 0.2 0.3 0.4 0.5 0.6 0.7 - Power Dissipation - WPD PO - Output Power - W Class-AB, VDD = 5 V, RL = 8 Ω Class-AB, V DD = 3.6 V, RL = 8 Ω VDD = 3.6 V, RL = 8 Ω, 33 µH VDD = 5 V, RL = 8 Ω, 33 µH 100 PO - Output Power - W Efficiency - % VDD = 3.6 RL = 32 Ω , 33 µH RL = 16 Ω , 33 µH RL = 8 Ω , 33 µH Class-AB, R L = 8 Ω 100 150 200 250 300 PO - Output Power - W VDD = 2.5 V, RL = 8 Ω , 33 µH VDD = 3.6 V, RL = 8 Ω , 33 µH VDD = 5 V, RL = 8 Ω , 33 µH Supply Current - mA 100 150 200 250 RL = 8 Ω , 33 µH VDD = 3.6 V RL = 32 Ω , 33 µH Supply Current - mA PO - Output Power - W 2.2 2.4 2.6 2.8 3.2 3.4 3.6 3.8 2.5 3 3.5 4 4.5 5 5.5 I(Q) − Quiescent Current − mA VDD − Supply Voltage − V No Load RL = 8 Ω , 33 µH 2.5 3 3.5 4 4.5 5 VDD - Supply Voltage - V - Output Power - WPO RL = 8 Ω f = 1 kHz Gain = 2 V/V THD+N = 1% THD+N = 10% 0.2 0.4 0.6 0.8 1.2 1.4 1.6 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 VDD = 2.5 V VDD = 3.6 V VDD = 5 V Shutdown Voltage - V - Shutdown Current - I(SD) Aµ 0.2 0.4 0.6 0.8 1.2 1.4 8 12 16 20 28 RL - Load Resistance - Ω - Output Power - WPO 3224 f = 1 kHz THD+N = 1% Gain = 2 V/V VDD = 2.5 V VDD = 3.6 V VDD = 5 V TPA2005D1-Q1 SLOS474 AUGUST 2005 EFFICIENCY EFFICIENCY POWER DISSIPATION vs vs vs OUTPUT POWER OUTPUT POWER OUTPUT POWER Figure Figure Figure SUPPLY CURRENT SUPPLY CURRENT QUIESCENT CURRENT vs vs vs OUTPUT POWER OUTPUT POWER SUPPLY VOLTAGE Figure Figure Figure SHUTDOWN CURRENT OUTPUT POWER OUTPUT POWER vs vs vs SHUTDOWN VOLTAGE SUPPLY VOLTAGE LOAD RESISTANCE Figure Figure Figure

www.ti.com 0.2 0.4 0.6 0.8 1.2 1.4 1.6 8 12 16 20 24 28 32 VDD = 5 V VDD = 3.6 V VDD = 2.5 V RL - Load Resistance - Ω - Output Power - WPO f = 1 kHz THD+N = 10% Gain = 2 V/V 0.1 0.2 0.5 0.01 20.1 1 PO − Output Power − W THD+N − Total Harmonic Distortion + Noise − % 5 V 3.6 V 2.5 V RL = 8 Ω , f = 1 kHz, Gain = 2 V/V 0.1 0.2 0.5 0.01 20.1 1 PO − Output Power − W THD+N − Total Harmonic Distortion + Noise − % 5 V 3.6 V 2.5 V RL = 16 Ω , f = 1 kHz, Gain = 2 V/V 0.008 0.02 0.05 0.1 0.2 0.5 20 100 1 k 20 k f − Frequency − Hz THD+N − Total Harmonic Distortion + Noise − % 50 mW 250 mW 1 W VDD = 5 V CI = 2 µF RL = 8 Ω Gain = 2 V/V 0.5 0.2 0.1 0.05 0.02 0.01 20 100 1 k 20 k f − Frequency − Hz THD+N − Total Harmonic Distortion + Noise − % VDD = 3.6 V CI = 2 µF RL = 8 Ω Gain = 2 V/V 500 mW 25 mW 125 mW 0.5 0.2 0.1 0.05 0.02 0.0120 100 1 k 20 k f − Frequency − Hz THD+N − Total Harmonic Distortion + Noise − % 15 mW VDD = 2.5 V CI = 2 µF RL = 8 Ω Gain = 2 V/V 75 mW 200 mW 0.1 0 0.5 1 1.5 2 2.5 3 3.5 VDD = 2.5 V VDD = 3.6 V f = 1 kHz P O = 200 mW VIC - Common Mode Input Voltage - V THD+N - Total Harmonic Distortion + Noise - % −80 −70 −60 −50 −40 −30 −20 −10 20 100 1 k 20 k f − Frequency − Hz − Supply Voltage Rejection Ratio − dBkSVR CI = 2 µF RL = 8 Ω Vp-p = 200 mV Inputs ac-Grounded Gain = 2 V/V VDD = 5 V VDD = 3.6 V VDD =2. 5 V VDD = 3.6 V CI = 2 µF RL = 16 Ω Gain = 2 V/V f − Frequency − Hz THD+N − Total Harmonic Distortion + Noise − % 0.5 0.2 0.1 0.05 0.02 0.01 20 100 1 k 20 k 15 mW 75 mW 200 mW TPA2005D1-Q1 SLOS474 AUGUST 2005 OUTPUT POWER TOTAL HARMONIC DISTORTION TOTAL HARMONIC DISTORTION vs NOISE NOISE LOAD RESISTANCE vs vs OUTPUT POWER OUTPUT POWER Figure 10. Figure 11. Figure 12. TOTAL HARMONIC DISTORTION TOTAL HARMONIC DISTORTION TOTAL HARMONIC DISTORTION NOISE NOISE NOISE vs vs vs FREQUENCY FREQUENCY FREQUENCY Figure 13. Figure 14. Figure 15. TOTAL HARMONIC DISTORTION TOTAL HARMONIC DISTORTION SUPPLY-VOLTAGE REJECTION NOISE NOISE RATIO vs vs vs FREQUENCY COMMON MODE INPUT VOLTAGE FREQUENCY Figure 16. Figure 17. Figure 18.

www.ti.com −80 −70 −60 −50 −40 −30 −20 −10 f − Frequency − Hz − Supply Voltage Rejection Ratio − dBkSVR 20 100 1 k 20 k Gain = 5 V/V C I = 2 µF RL = 8 Ω Vp-p = 200 mV Inputs ac-Grounded VDD = 5 V VDD = 2. 5 V VDD = 3.6 V −100 −90 −80 −70 −60 −50 −40 −30 −20 −10 f − Frequency − Hz − Supply Voltage Rejection Ratio − dBkSVR VDD = 3.6 V CI = 2 µF RL = 8 Ω Inputs Floating Gain = 2 V/V 20 100 1 k 20 k -90 -80 -70 -60 -50 -40 -30 -20 -10 VIC - Common Mode Input Voltage - V f = 217 Hz R L = 8 Ω Gain = 2 V/V VDD = 2.5 V - Supply Voltage Rejection Ratio - dBkSVR VDD = 3.6 V VDD = 5 V C1 - Duty 12.6% C1 - Frequency 216.7448 Hz C1 - Amplitude 512 mV C1 - High 3.544 V Voltage - V t - Time - ms VDD VOUT -150 -100 -50 0 400 800 1200 1600 2000 -150 -100 -50 f - Frequency - Hz - Output Voltage - dBVVO - Supply Voltage - dBVVDD VDD Shown in Figure 22 CI = 2 µF, Inputs ac-grounded Gain = 2V/V −70 −60 −50 −40 −30 −20 −10 f − Frequency − Hz CMRR − Common Mode Rejection Ratio − dB 20 100 1 k 20 k VDD = 2.5 V to 5 V VIC = 1 Vp−p RL = 8 Ω Gain = 2 V/V -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 RL = 8 Ω Gain = 2 V/V VIC - Common Mode Input Voltage - V CMRR - Common Mode Rejection Ratio - dB VDD = 5 V VDD = 2.5 V VDD = 3.6 V TPA2005D1-Q1 SLOS474 AUGUST 2005 SUPPLY-VOLTAGE REJECTION SUPPLY-VOLTAGE REJECTION SUPPLY-VOLTAGE REJECTION RATIO RATIO RATIO vs vs vs FREQUENCY FREQUENCY COMMON-MODE INPUT VOLTAGE Figure 19. Figure 20. Figure 21. GSM POWER-SUPPLY REJECTION GSM POWER-SUPPLY REJECTION vs vs TIME FREQUENCY Figure 22. Figure 23. COMMON-MODE REJECTION RATIO COMMON-MODE REJECTION RATIO vs vs FREQUENCY COMMON-MODE INPUT VOLTAGE Figure 24. Figure 25.

www.ti.com APPLICATION INFORMATION FULLY DIFFERENTIAL AMPLIFIER Advantages of Fully Differential Amplifiers COMPONENT SELECTION IN± IN+ PWM H± Bridge VO+ VO± Internal Oscillator CS To Battery VDD GND Bias Circuitry RI RI Differential Input TPA2005D1 Filter-Free Class D SHUTDOWN TPA2005D1-Q1 SLOS474 AUGUST 2005 The TPA2005D1 is a fully differential amplifier with differential inputs and outputs. The fully differential amplifier consists of a differential amplifier and a common-mode amplifier. The differential amplifier ensures that the amplifier outputs a differential voltage on the output that is equal to the differential input times the gain. The common-mode feedback ensures that the common-mode voltage at the output is biased around V DD /2, regardless of the common-mode voltage at the input. The fully differential TPA2005D1 can still be used with a single-ended input; however, the TPA2005D1 should be used with differential inputs when in a noisy environment, like a wireless handset, to ensure maximum noise rejection. Input-coupling capacitors not required: The fully differential amplifier allows the inputs to be biased at a voltage other than midsupply. For example, if a codec has a midsupply lower than the midsupply of the TPA2005D1, the common-mode feedback circuit adjusts, and the TPA2005D1 outputs still is biased at midsupply of the TPA2005D1. The inputs of the TPA2005D1 can be biased from 0.5 V to V DD 0.8 If the inputs are biased outside of that range, input-coupling capacitors are required. Midsupply bypass capacitor, C (BYPASS) not required: The fully differential amplifier does not require a bypass capacitor. This is because any shift in the midsupply affects both positive and negative channels equally and cancels at the differential output. Better RF-immunity: GSM handsets save power by turning on and shutting off the RF transmitter at a rate of 217 Hz. The transmitted signal is picked-up on input and output traces. The fully differential amplifier cancels the signal much better than the typical audio amplifier. Figure shows the TPA2005D1 typical schematic with differential inputs, and Figure shows the TPA2005D1 with differential inputs and input capacitors, and Figure shows the TPA2005D1 with single-ended inputs. Differential inputs should be used whenever possible, because the single-ended inputs are much more susceptible to noise. Table Typical Component Values REF DES VALUE EIA SIZE MANUFACTURER PART NUMBER R I 150 k Ω 0.5%) 0402 Panasonic ERJ2RHD154V C S µ F (+22%, -80%) 0402 Murata GRP155F50J105Z C I (1) 3.3 nF 10%) 0201 Murata GRP033B10J332K (1) C I is needed only for single-ended input or if V ICM is not between 0.5 V and V DD 0.8 C I 3.3 nF (with R I 150 k Ω gives a high-pass corner frequency of 321 Hz. Figure 26. Typical TPA2005D1 Application Schematic With Differential Input for a Wireless Phone

www.ti.com IN± IN+ PWM H± Bridge VO+ VO± Internal Oscillator CS To Battery VDD GND Bias Circuitry RI RI Differential Input TPA2005D1 Filter-Free Class D SHUTDOWN CI CI IN± IN+ PWM H± Bridge VO+ VO± Internal Oscillator CS To Battery VDD GND Bias Circuitry RI RI Single-ended Input TPA2005D1 Filter-Free Class D SHUTDOWN CI CI TPA2005D1-Q1 SLOS474 AUGUST 2005 Figure 27. TPA2005D1 Application Schematic With Differential Input and Input Capacitors Figure 28. TPA2005D1 Application Schematic With Single-Ended Input

www.ti.com Input Resistors I Gain 2 150 k RI V V (1) Decoupling Capacitor S Input Capacitors I fc 1

2 RICI

(2) CI 1

2 RI fc

(3) SUMMING INPUT SIGNALS WITH THE TPA2005D1 TPA2005D1-Q1 SLOS474 AUGUST 2005 The input resistors I set the gain of the amplifier according to equation Equation Resistor matching is very important in fully differential amplifiers. The balance of the output on the reference voltage depends on matched ratios of the resistors. CMRR, PSRR, and cancellation of the second harmonic distortion diminish if resistor mismatch occurs. Therefore, it is recommended to use tolerance resistors, or better, to keep the performance optimized. Matching is more important than overall tolerance. Resistor arrays with matching can be used with a tolerance greater than 1%. Place the input resistors very close to the TPA2005D1 to limit noise injection on the high-impedance nodes. For optimal performance, the gain should be set to V/V or lower. Lower gain allows the TPA2005D1 to operate at its best and keeps a high voltage at the input, making the inputs less susceptible to noise. The TPA2005D1 is a high-performance class-D audio amplifier that requires adequate power-supply decoupling to ensure the efficiency is high and total harmonic distortion (THD) is low. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically µ placed as close as possible to the device V DD lead, works best. Placing this decoupling capacitor close to the TPA2005D1 is very important for the efficiency of the class-D amplifier, because any resistance or inductance in the trace between the device and the capacitor can cause a loss in efficiency. For filtering lower-frequency noise signals, a 10- µ or greater, capacitor placed near the audio power amplifier also helps, but it is not required in most device. The TPA2005D1 does not require input coupling capacitors if the design uses a differential source that is biased from 0.5 V to V DD 0.8 V (shown in Figure If the input signal is not biased within the recommended common-mode input range, if needing to use the input as a high pass filter (shown in Figure or if using a single-ended source (shown in Figure input coupling capacitors are required. The input capacitors and input resistors form a high-pass filter with the corner frequency, f c determined in Equation The value of the input capacitor is important to consider, as it directly affects the bass (low frequency) performance of the circuit. Speakers in wireless phones usually cannot respond well to low frequencies, so the corner frequency can be set to block low frequencies in this application. Equation is reconfigured to solve for the input coupling capacitance. If the corner frequency is within the audio band, the capacitors should have a tolerance of 10% or better, because any mismatch in capacitance causes an impedance mismatch at the corner frequency and below. For a flat low-frequency response, use large input coupling capacitors µ F). However, in a GSM phone the ground signal is fluctuating at 217 Hz, but the signal from the codec does not have the same 217-Hz fluctuation. The difference between the two signals is amplified, sent to the speaker, and heard as a 217-Hz hum. Most wireless phones or PDAs need to sum signals at the audio power amplifier or just have two signal sources that need separate gain. The TPA2005D1 makes it easy to sum signals or use separate signal sources with different gains. Many phones now use the same speaker for the earpiece and ringer, where the wireless phone would require a much lower gain for the phone earpiece than for the ringer. PDAs and phones that have stereo headphones require summing of the right and left channels to output the stereo signal to the mono speaker.

www.ti.com Summing Two Differential Input Signals Gain 1 VO VI1 2 150 k RI1 V V (4) Gain 2 VO VI2 2 150 k RI2 V V (5) IN± IN+ PWM H± Bridge VO+ VO± Internal Oscillator CS To Battery VDD GND Bias Circuitry RI2 RI2 Differential Input 1 SHUTDOWN RI1 RI1 Differential Input 2 Filter-Free Class D Summing a Differential Input Signal and a Single-Ended Input Signal Gain 1 VO VI1 2 150 k RI1 V V (6) Gain 2 VO VI2 2 150 k RI2 V V (7) CI2 1

2 RI2 fc2

(8) TPA2005D1-Q1 SLOS474 AUGUST 2005 Two extra resistors are needed for summing differential signals total of components). The gain for each input source can be set independently (see Equation and Equation and Figure If summing left and right inputs with a gain of V/V, use R R 300 k Ω If summing a ring tone and a phone signal, set the ring-tone gain to gain V/V, and the phone gain to gain 0.1 V/V. The resistor values are: R M Ω and R 150 k Ω Figure 29. Application Schematic With TPA2005D1 Summing Two Differential Inputs Figure shows how to sum a differential input signal and a single-ended input signal. Ground noise can couple in through IN+ with this method. It is better to use differential inputs. The corner frequency of the single-ended input is set by C shown in Equation To ensure that each input is balanced, the single-ended input must be driven by a low-impedance source even if the input is not in use.

www.ti.com CI2 1 2 150k 20Hz (9) CI2 53pF (10) IN± IN+ PWM H± Bridge VO+ VO± Internal Oscillator CS To Battery VDD GND Bias Circuitry RI2 RI2 Differential Input 1 Filter-Free Class D SHUTDOWN RI1 RI1 Single-Ended Input 2 CI2 CI2 Summing Two Single-Ended Input Signals TPA2005D1-Q1 SLOS474 AUGUST 2005 If summing a ring tone and a phone signal, the phone signal should use a differential input signal while the ring tone might be limited to a single-ended signal. If phone gain is set at gain 0.1 V/V, and the ring-tone gain is set to gain V/V, the resistor values are: R M Ω and R 150 k Ω The high-pass corner frequency of the single-ended input is set by C If the desired corner frequency is less than Hz, then: Figure 30. Application Schematic With TPA2005D1 Summing Differential Input and Single-Ended Input Signals Four resistors and three capacitors are needed for summing single-ended input signals. The gain and corner frequencies and f for each input source can be set independently (see Equation through Equation and Figure Resistor, R P and capacitor, C P are needed on the IN+ terminal to match the impedance on the IN- terminal. The single-ended inputs must be driven by low-impedance sources, even if one of the inputs is not outputting an ac signal.

www.ti.com Gain 1 VO VI1 2 150 k RI1 V V (11) Gain 2 VO VI2 2 150 k RI2 V V (12) CI1 1

2 RI1 fc1

(13) CI2 1 (14) CP CI1 CI2 (15) RP RI1 RI2 RI1 RI2 (16) IN± IN+ PWM H± Bridge VO+ VO± Internal Oscillator CS To Battery VDD GND Bias Circuitry RI2 RP Filter-Free Class D SHUTDOWN RI1 Single-Ended Input 2 CI2 CP Single-Ended Input 1 CI1 EFFICIENCY AND THERMAL INFORMATION JA 1 Derating Factor 1 0.016 62.5° CW (17) TAMax TJMax JAPDmax 150 62.5 (0.2) 137.5° C (18) TPA2005D1-Q1 SLOS474 AUGUST 2005 Figure 31. Application Schematic With TPA2005D1 Summing Two Single-Ended Inputs The maximum ambient temperature depends on the heat-sinking ability of the PCB system. The derating factor for the 2,5-mm x 2,5-mm MicroStar Junior package is shown in the dissipation rating table. Converting this to θ JA Given θ JA of 62.5 C/W, the maximum allowable junction temperature of 150 and the maximum internal dissipation of 0.2 W (worst case 5-V supply), the maximum ambient temperature can be calculated with equation Equation Equation shows that the calculated maximum ambient temperature is 137.5 C at maximum power dissipation with a 5-V supply; however, the maximum ambient temperature of the package is limited to Because of the efficiency of the TPA2005D1, it can be operated under all conditions to an ambient temperature of The TPA2005D1 is designed with thermal protection that turns the device off when the junction temperature surpasses 150 C to prevent damage to the IC. Also, using speakers more resistive than Ω dramatically increases the thermal performance by reducing the output current and increasing the efficiency of the amplifier.

www.ti.com BOARD LAYOUT Component Location Trace Width MicroStar Junior BGA Layout 0,28 mm 0,38 mm 0,25 mm SD NC IN+ IN− GND GND GND GND GND GND GND VDD VDD Vo+ Vo− Solder Mask Paste Mask Copper Trace 8-Pin QFN (DRB) Layout TPA2005D1-Q1 SLOS474 AUGUST 2005 Place all the external components very close to the TPA2005D1. The input resistors need to be very close to the TPA2005D1 input pins so noise does not couple on the high-impedance nodes between the input resistors and the input amplifier of the TPA2005D1. Placing the decoupling capacitor, C S close to the TPA2005D1 is important for the efficiency of the class-D amplifier. Any resistance or inductance in the trace between the device and the capacitor can cause a loss in efficiency. Make the high current traces going to pins VDD, GND, V and V of the TPA2005D1 have a minimum width of 0,7 mm. If these traces are too thin, the TPA2005D1 performance and output power will decrease. The input traces do not need to be wide, but do need to run side-by-side to enable common-mode noise cancellation. Use the following MicroStar Junior BGA ball diameters: 0,25 mm diameter solder mask 0,28 mm diameter solder paste mask/stencil 0,38 mm diameter copper trace Figure shows how to lay out a board for the TPA2005D1 MicroStar Junior BGA. Figure 32. TPA2005D1 MicroStar Junior BGA Board Layout (Top View) Use the following land pattern for board layout with the 8-pin QFN (DRB) package. Note that the solder paste should use a hatch pattern to fill solder paste at 50% to ensure that there is not too much solder paste under the package.

www.ti.com 0,65 mm 0,38 mm Solder Mask: 1,4 mm x 1,85 mm centered in package 0,7 mm 1,4 mm Make solder paste a hatch pattern to fill 50% 3,3 mm 1,95 mm 0,33 mm plugged vias (5 places) ELIMINATING THE OUTPUT FILTER WITH THE TPA2005D1 Effect on Audio Traditional Class-D Modulation Scheme TPA2005D1-Q1 SLOS474 AUGUST 2005 Figure 33. TPA2005D1 8-Pin QFN (DRB) Board Layout (Top View) This section focuses on why the user can eliminate the output filter with the TPA2005D1. The class-D amplifier outputs a pulse-width modulated (PWM) square wave, which is the sum of the switching waveform and the amplified input audio signal. The human ear acts as a band-pass filter such that only the frequencies between approximately Hz and kHz are passed. The switching frequency components are much greater than kHz, so the only signal heard is the amplified input audio signal. The traditional class-D modulation scheme, which is used in the TPA005Dxx family, has a differential output in which each output is 180 degrees out of phase and changes from ground to the supply voltage, V DD Therefore, the differential pre-filtered output varies between positive and negative V DD where filtered 50% duty cycle yields V across the load. The traditional class-D modulation scheme with voltage and current waveforms is shown in Figure Note that, even at an average of V across the load (50% duty cycle), the current to the load is high, causing a high loss and thus causing a high supply current.

www.ti.com 0 V ±5 V +5 V Current OUT+ Differential Voltage Across Load OUT± TPA2005D1 Modulation Scheme 0 V ±5 V +5 V Current OUT+ OUT± Differential Voltage Across Load 0 V ±5 V +5 V Current OUT+ OUT± Differential Voltage Across Load Output = 0 V Output > 0 V TPA2005D1-Q1 SLOS474 AUGUST 2005 Figure 34. Traditional Class-D Modulation Scheme Output Voltage and Current Waveforms Into an Inductive Load With No Input The TPA2005D1 uses a modulation scheme that still has each output switching from to the supply voltage. However, OUT+ and OUT- are now in phase with each other, with no input. The duty cycle of OUT+ is greater than 50% and OUT- is less than 50% for positive voltages. The duty cycle of OUT+ is less than 50% and OUT- is greater than 50% for negative voltages. The voltage across the load remains at V throughout most of the switching period, greatly reducing the switching current, which reduces any I R losses in the load. Figure 35. The TPA2005D1 Output Voltage and Current Waveforms Into an Inductive Load

www.ti.com Efficiency: Why You Must Use a Filter With the Traditional Class-D Modulation Scheme Effects of Applying a Square Wave Into a Speaker PSPKR PSUP±PSUP THEORETICAL (at max output power) (19) PSPKR PSUP POUT PSUP THEORETICAL POUT (at max output power) (20) PSPKR POUT 1 MEASURED THEORETICAL (at max output power) (21) THEORETICAL RL RL 2rDS(on) (at max output power) (22) When to Use an Output Filter TPA2005D1-Q1 SLOS474 AUGUST 2005 The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform results in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple current is large for the traditional modulation scheme because the ripple current is proportional to voltage multiplied by the time at that voltage. The differential voltage swing is V DD and the time at each voltage is one-half the period for the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from each half-cycle for the next half-cycle, while any resistance causes power dissipation. The speaker is both resistive and reactive, whereas an LC filter is almost purely reactive. The TPA2005D1 modulation scheme has very little loss in the load without a filter because the pulses are very short and the change in voltage is V DD instead of V DD As the output power increases, the pulses widen, making the ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for most needed. An LC filter with a cutoff frequency less than the class-D switching frequency allows the switching current to flow through the filter instead of the load. The filter has less resistance than the speaker, resulting in less power dissipation, which increases efficiency. If the amplitude of a square wave is high enough and the frequency of the square wave is within the bandwidth of the speaker, a square wave could cause the voice coil to jump out of the air gap and/or scar the voice coil. A 250-kHz switching frequency, however, is not significant because the speaker cone movement is proportional to 1/f for frequencies beyond the audio band. Therefore, the amount of cone movement at the switching frequency is very small. However, damage could occur to the speaker if the voice coil is not designed to handle the additional power. To size the speaker for added power, the ripple current dissipated in the load must be calculated by subtracting the theoretical supplied power, P SUP THEORETICAL from the actual supply power, P SUP at maximum output power, P OUT The switching power dissipated in the speaker is the inverse of the measured efficiency, η MEASURED minus the theoretical efficiency, η THEORETICAL The maximum efficiency of the TPA2005D1 with a 3.6-V supply and an Ω load is 86% from Equation Using Equation with the efficiency at maximum power (84%), we see that there is an additional mW dissipated in the speaker. The added power dissipated in the speaker is not an issue as long as it is taken into account when choosing the speaker. Design the TPA2005D1 without an output filter if the traces from amplifier to speaker are short. The TPA2005D1 passed FCC and CE radiated emissions with no shielding and with speaker trace wires 100 mm long or less. Wireless handsets and PDAs are great a filter. A ferrite bead filter often can be used if the design is failing radiated emissions without an LC filter, and the frequency-sensitive circuit is greater than MHz. This is good for circuits that just have to pass FCC and CE because FCC and CE only test radiated emissions greater than MHz. If choosing a ferrite bead, choose one with high impedance at high frequencies, but very low impedance at low frequencies. Use an LC output filter if there are low-frequency MHz) EMI-sensitive circuits and/or there are long leads from amplifier to speaker. Figure and Figure show typical ferrite bead and LC output filters.

www.ti.com 1 nF Ferrite Chip Bead OUTP OUTN Ferrite Chip Bead 1 nF 1 µF 1 µF 33 µH 33 µH OUTP OUTN TPA2005D1-Q1 SLOS474 AUGUST 2005 Figure 36. Typical Ferrite Chip Bead Filter (Chip bead example: NEC/Tokin: N2012ZPS121) Figure 37. Typical LC Output Filter, Cut-Off Frequency of kHz

Orderable Device Status(1) Package Type Package Drawing Pins Package Qty Eco Plan(2) Lead/Ball FinishMSL Peak Temp (3) TPA2005D1DRBQ1 ACTIVE SON DRB 8 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR (1)The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2)Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontentfor the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS):TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt):This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br):TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. PACKAGE OPTION ADDENDUM www.ti.com 18-Apr-2006 Addendum-Page 1

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