TPA122D

TPA122 TI | Alldatasheet

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IN1− BYPASS GND VDD VO2 IN2− SHUTDOWN D OR DGN PACKAGE (TOP VIEW) TYPICAL APPLICATION CIRCUIT Audio Input Bias Control VO1 VO2 VDD IN1± BYPASS SHUTDOWN VDD/2 CI RI RF 320 kΩ 320 kΩ CB CS Audio Input CI RI IN2± RF VDD From Shutdown Control Circuit CC CC TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 150-mW STEREO AUDIO POWER AMPLIFIER 150-mW Stereo Output The TPA122 is a stereo audio power amplifier pack- aged in either an 8-pin SOIC, or an 8-pin PC Power Supply Compatible PowerPAD MSOP package capable of delivering Fully Specified for 3.3-V and 5-V Operation 150 mW of continuous RMS power per channel into Operation to 2.5 V Ω loads. Amplifier gain is externally configured by Pop Reduction Circuitry means of two resistors per input channel and does not require external compensation for settings of to Internal Midrail Generation 10. Thermal and Short-Circuit Protection THD+N when driving an Ω load from V is 0.1% at Surface-Mount Packaging kHz, and less than across the audio band of PowerPAD MSOP Hz to kHz. For 32- Ω loads, the THD+N is reduced SOIC to less than 0.06% at kHz, and is less than across the audio band of Hz to kHz. For 10-k Ω Pin Compatible With LM4880 and LM4881 loads, the THD+N performance is 0.01% at kHz, (SOIC) and less than 0.02% across the audio band of Hz to kHz. Please be aware that an important notice concerning availability, standard warranty, and use in critical

applications

sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Copyright 1998 2004, Texas Instruments Incorporated Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

www.ti.com ABSOLUTE MAXIMUM RATINGS DISSIPATION RATING TABLE TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. AVAILABLE OPTIONS PACKAGED DEVICES MSOP T A SMALL OUTLINE (1) MSOP (1) SYMBOLIZATION (D) (DGN) C to C TPA122D TPA122DGN TI AAE (1) The D and DGN packages are available in left-ended tape and reel only (e.g., TPA122DR, TPA122DGNR). Terminal Functions TERMINAL I/O NO. BYPASS I Tap to voltage divider for internal mid-supply bias supply. Connect to a 0.1 µ F to µ F low ESR capacitor for best performance. GND I GND is the ground connection. IN1- I IN1- is the inverting input for channel IN2- I IN2- is the inverting input for channel SHUTDOWN I Puts the device in a low quiescent current mode when held high V DD I V DD is the supply voltage terminal. V O O V O is the audio output for channel V O O V O is the audio output for channel over operating free-air temperature range (unless otherwise noted) (1) UNIT V DD Supply voltage V V I Input voltage 0.3 V to V DD 0.3 V Continuous total power dissipation Internally limited T J Operating junction temperature range C to 150 C T stg Storage temperature range C to 150 C Lead temperature 1,6 mm (1/16 inch) from case for seconds 260 C (1) Stresses beyond those listed under "absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. T A C DERATING FACTOR T A C T A C PACKAGE POWER RATING ABOVE T A C POWER RATING POWER RATING D 725 mW 5.8 mW/ C 464 mW 377 mW DGN 2.14 W (1) 17.1 mW/ C 1.37 W 1.11 W (1) See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD of that document.

www.ti.com RECOMMENDED OPERATING CONDITIONS DC ELECTRICAL CHARACTERISTICS AC OPERATING CHARACTERISTICS DC ELECTRICAL CHARACTERISTICS TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 MIN MAX UNIT V DD Supply voltage 2.5 5.5 V T A Operating free-air temperature C V IH High-level input voltage, (SHUTDOWN) 0.80 V DD V V IL Low-level input voltage, (SHUTDOWN) 0.40 V DD V at T A V DD 3.3 V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V OO Output offset voltage mV PSRR Power supply rejection ratio V DD 3.2 V to 3.4 V dB I DD Supply current V DD 2.5, SHUTDOWN V 1.5 mA I DD(SD) Supply current in SHUTDOWN mode V DD 2.5, SHUTDOWN V DD µ A Z I Input impedance M Ω V DD 3.3 T A R L Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT P O Output power (each channel) THD 0.1% (1) mW THD+N Total harmonic distortion noise P O mW, Hz kHz B OM Maximum output power BW G 10, THD kHz Phase margin Open loop Supply ripple rejection f kHz dB Channel/channel output separation f kHz dB SNR Signal-to-noise ratio P O 100 mW 100 dB V n Noise output voltage 9.5 µ V(rms) (1) Measured at kHz at T A V DD 5.5 V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V OO Output offset voltage mV PSRR Power supply rejection ratio V DD 4.9 V to 5.1 V dB I DD Supply current SHUTDOWN V 1.5 mA I DD(SD) Supply current in SHUTDOWN mode SHUTDOWN V DD 100 µ A IH High-level input current (SHUTDOWN) V DD 5.5 V I V DD µ A IL Low-level input current (SHUTDOWN) V DD 5.5 V I V µ A Z I Input impedance M Ω

www.ti.com AC OPERATING CHARACTERISTICS AC OPERATING CHARACTERISTICS AC OPERATING CHARACTERISTICS TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 V DD T A R L Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT P O Output power (each channel) THD 0.1% (1) mW THD+N Total harmonic distortion noise P O 150 mW, Hz kHz B OM Maximum output power BW G 10, THD kHz Phase margin Open loop Supply ripple rejection ratio f kHz dB Channel/channel output separation f kHz dB SNR Signal-to-noise ratio P O 150 mW 100 dB V n Noise output voltage 9.5 µ V(rms) (1) Measured at kHz V DD 3.3 T A R L Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT P O Output power (each channel) THD 0.1% (1) mW THD+N Total harmonic distortion noise P O mW, Hz kHz 0.5% B OM Maximum output power BW G 10, THD kHz Phase margin Open loop Supply ripple rejection f kHz dB Channel/channel output separation f kHz dB SNR Signal-to-noise ratio P O 100 mW 100 dB V n Noise output voltage 9.5 µ V(rms) (1) Measured at kHz V DD T A R L Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT P O Output power (each channel) THD 0.1% (1) mW THD+N Total harmonic distortion noise P O mW, Hz kHz 0.4% B OM Maximum output power BW G 10, THD kHz Phase margin Open loop Supply ripple rejection f kHz dB Channel/channel output separation f kHz dB SNR Signal-to-noise ratio P O 150 mW 100 dB V n Noise output voltage 9.5 µ V(rms) (1) Measured at kHz

www.ti.com TYPICAL CHARACTERISTICS 0.1 0.01 0.001 20 100 1k 10k 20k AV = −10 V/V AV = −5 V/V THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz AV = −1 V/V VDD = 3.3 V PO = 30 mW CB = 1 µ F RL = 32 Ω 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 3.3 V AV = −1 V/V RL = 32 Ω CB = 1 µF PO = 10 mW THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz PO = 15 mW PO = 30 mW TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 Table of Graphs FIGURE 10, 11, 13, vs Frequency 14, 16, 17, 34, THD+N Total harmonic distortion plus noise vs Output power 12, 15, Supply ripple rejection vs Frequency 19, V n Output noise voltage vs Frequency 21, Crosstalk vs Frequency 23-26, 37, Mute attenuation vs Frequency 27, Open-loop gain and phase margin vs Frequency 29, Output power vs Load resistance 31, Phase vs Frequency 39-44 I DD Supply current vs Supply voltage SNR Signal-to-noise ratio vs Voltage gain Closed-loop gain vs Frequency 39-44 Power dissipation/amplifier vs Output power 45, TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY FREQUENCY Figure Figure

www.ti.com 10 kHz 0.1 0.01 THD+N −Total Harmonic Distortion + Noise − % PO − Output Power − mW 1 10 50 VDD = 3.3 V RL = 32 Ω AV = −1 V/V CB = 1 µF 20 kHz 1 kHz 20 Hz 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 5 V PO = 60 mW RL = 32 Ω CB = 1 µF AV = −10 V/V THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz AV = −5 V/V AV = −1 V/V 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 5 V RL = 32 Ω AV = −1 V/V CB = 1 µF PO = 15 mW THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz PO = 30 mW PO = 60 mW THD+N −Total Harmonic Distortion + Noise − % 20 kHz 0.1 0.01 PO − Output Power − W VDD = 5 V AV = −1 V/V RL = 32 Ω CB = 1 µF 10 kHz 1 kHz 20 Hz 0.002 0.01 0.1 0.2 TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs OUTPUT POWER FREQUENCY Figure Figure TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY OUTPUT POWER Figure Figure

www.ti.com 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 3.3 V RL = 10 kΩ PO = 100 µF CB = 1 µF THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz AV = −5 V/V AV = −2 V/V 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 3.3 V RL = 10 kΩ AV = −1 V/V CB = 1 µF THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz PO = 45 µW PO = 130 µW PO = 90 µW 5 10 100 200 THD+N −Total Harmonic Distortion + Noise − % 20 Hz 0.01 0.001 PO − Output Power − µW 10 kHz 1 kHz 20 Hz 0.1 VDD = 3.3 V RL = 10 kΩ AV = −1 V/V CB = 1 µF 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 5 V RL = 10 kΩ PO = 300 µW CB = 1 µF THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz AV = −1 V/V AV = −2 V/V AV = −5 V/V TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY FREQUENCY Figure Figure TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs OUTPUT POWER FREQUENCY Figure Figure 10.

www.ti.com 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 5 V RL = 10 kΩ AV = −1 V/V CB = 1 µF THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz PO = 300 µW PO = 200 µW PO = 100 µW 0.1 0.01 0.001 5 10 100 500 VDD = 5 V RL = 10 kΩ AV = −1 V/V CB = 1 µ F THD+N −Total Harmonic Distortion + Noise − % 20 Hz 1 kHz 20 kHz 10 kHz PO − Output Power − µW THD+N − Total Harmonic Distortion Plus Noise − % f − Frequency − Hz 0.1 0.01 0.001 100 1k 10k 20k AV = −1 V/V AV = −2 V/V AV = −5 V/V VDD = 3.3 V PO = 75 mW RL = 8 Ω CB = 1 µF 0.1 0.01 0.001 20 100 1k 10k 20k VDD = 3.3 V RL = 8 Ω AV = −1 V/V THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz PO = 75 mW PO = 15 mW PO = 30 mW TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY OUTPUT POWER Figure 11. Figure 12. TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY FREQUENCY Figure 13. Figure 14.

www.ti.com THD+N − Total Harmonic Distortion Plus Noise − % f − Frequency − Hz 0.1 0.01 0.001 100 1k 10k 20k AV = −1 V/V AV = −2 V/V AV = −5 V/V VDD = 5 V PO = 100 mW RL = 8 Ω CB = 1 µF 20 kHz 0.1 0.01 THD+N −Total Harmonic Distortion + Noise − % PO − Output Power − W 10m 0.1 0.3 VDD = 3.3 V RL = 8 Ω AV = −1 V/V 10 kHz 1 kHz 20 Hz 0.1 0.01 0.001 20 100 1k 10k 20k THD+N −Total Harmonic Distortion + Noise − % f − Frequency − Hz VDD = 5 V RL = 8 Ω AV = −1 V/V PO = 30 mW PO = 60 mW PO = 10 mW 20 kHz 0.1 0.01 THD+N −Total Harmonic Distortion + Noise − % PO − Output Power − W 10m 0.1 1 1 kHz 20 Hz 10 kHz VDD = 5 V RL = 8 Ω AV = −1 V/V TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs OUTPUT POWER FREQUENCY Figure 15. Figure 16. TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY OUTPUT POWER Figure 17. Figure 18.

www.ti.com 20 100 20k f − Frequency − Hz −50 −70 −90 −60 −80 −100 VDD = 3.3 V RL = 8 Ω to 10 kΩ −40 −10 −30 −20 10k CB = 0.1 µF CB = 1 µF CB = 2 µF Bypass = 1.65 V Supply Ripple Rejection Ratio − dB 20 100 20k f − Frequency − Hz −50 −70 −90 −60 −80 −100 VDD = 5 V RL = 8 Ω to 10 kΩ −40 −10 −30 −20 10k CB = 0.1 µF CB = 1 µF CB = 2 µF Bypass = 2.5 V Supply Ripple Rejection Ratio − dB f − Frequency − Hz 20 100 1k 10k 20k VDD = 3.3 V BW = 10 Hz to 22 kHz AV = −1 V/V RL = 8 Ω to 10 kΩ − Output Noise Voltage − VµVn f − Frequency − Hz 20 100 1k 10k 20k VDD = 5 V BW = 10 Hz to 22 kHz RL = 8 Ω to 10 kΩ AV = −1 V/V − Output Noise Voltage − VµVn TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 SUPPLY RIPPLE REJECTION RATIO SUPPLY RIPPLE REJECTION RATIO vs vs FREQUENCY FREQUENCY Figure 19. Figure 20. OUTPUT NOISE VOLTAGE OUTPUT NOISE VOLTAGE vs vs FREQUENCY FREQUENCY Figure 21. Figure 22.

www.ti.com 20 100 20k f − Frequency − Hz −85 −95 −105 −90 −100 −110 PO = 25 mW VDD = 3.3 V RL = 32 Ω CB = 1 µF AV = −1 V/V −80 −65 −75 −60 −70 Crosstalk − dB 10k IN2 TO OUT1 IN1 TO OUT2 20 100 20k f − Frequency − Hz −75 −85 −95 −80 −90 −100 PO = 100 mW VDD = 3.3 V RL = 8 Ω CB = 1 µF AV = −1 V/V −70 −55 −65 −50 −60 Crosstalk − dB 10k IN2 TO OUT1 IN1 TO OUT2 20 100 10k f − Frequency − Hz −90 −100 −110 −95 −105 −85 −65 −80 −60 −75 −65 20k VDD = 5 V PO = 25 mW CB = 1 µF RL = 32 Ω AV = −1 V/V Crosstalk − dB IN2 TO OUT1 IN1 TO OUT2 20 100 10k f − Frequency − Hz −80 −90 −100 −85 −95 −75 −55 −70 −50 −65 −60 20k VDD = 5 V PO = 100 mW CB = 1 µF RL = 8 Ω AV = −1 V/V Crosstalk − dB IN2 TO OUT1 IN1 TO OUT2 TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 CROSSTALK CROSSTALK vs vs FREQUENCY FREQUENCY Figure 23. Figure 24. CROSSTALK CROSSTALK vs vs FREQUENCY FREQUENCY Figure 25. Figure 26.

www.ti.com 20 100 20k f − Frequency − Hz −50 −70 −90 −60 −80 −100 VDD = 3.3 V RL = 32 Ω CB = 1 µF −40 −10 −30 −20 Mute Attenuation − dB 10k 20 100 10k f − Frequency − Hz −60 −80 −100 −70 −90 −50 −10 −40 −30 −20 20k VDD = 5 V CB = 1 µF RL = 32 Ω Mute Attenuation − dB −20 −30° VDD = 3.3 V TA = 25° C No Load 100 30° 60° 90° 120° 150° mφ − Phase Margin 100 10k f − Frequency − Hz 1k 100k 10M10 Open-Loop Gain − dB Phase Gain TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 MUTE ATTENUATION MUTE ATTENUATION vs vs FREQUENCY FREQUENCY Figure 27. Figure 28. OPEN-LOOP GAIN AND PHASE MARGIN vs FREQUENCY Figure 29.

www.ti.com f − Frequency − Hz −20 100 1k 10k 10M1M100k Open-Loop Gain − dB 100 VDD = 5 V TA = 25° C No Load −30° 30° 60° 90° 120° 150° Phase Gain mφ − Phase Margin RL − Load Resistance − Ω 100 16 32 24 40 64 120 48 56 THD+N = 1 % VDD = 3.3 V AV = −1 V/V PO − Output Power − mW 250 100 16 32 200 150 24 40 64 300 48 56 THD+N = 1 % VDD = 5 V AV = −1 V/V PO − Output Power − mW RL − Load Resistance − Ω TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 OPEN-LOOP GAIN AND PHASE MARGIN vs FREQUENCY Figure 30. OUTPUT POWER OUTPUT POWER vs vs LOAD RESISTANCE LOAD RESISTANCE Figure 31. Figure 32.

www.ti.com THD+N − Total Harmonic Distortion Plus Noise − % f − Frequency − Hz 0.1 0.01 0.001 100 1k 10k 20k VI = 1 V AV = −1 V/V RL = 10 kΩ CB = 1 µF VDD − Supply Voltage − V 0.6 0.2 3 4 0.8 0.4 3.5 4.5 1.4 2.5 5 5.5 1.2 I DD − Supply Current − mA THD+N − Total Harmonic Distortion Plus Noise − % f − Frequency − Hz 0.1 0.01 0.001 100 1k 10k 20k VDD = 5 V AV = −1 V/V RL = 10 kΩ CB = 1 µF SNR − Signal-to-Noise Ratio − dB AV − Voltage Gain − V/V 104 100 5 7 9 10 102 862 43 VI = 1 V TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 SUPPLY CURRENT TOTAL HARMONIC DISTORTION NOISE vs vs SUPPLY VOLTAGE FREQUENCY Figure 33. Figure 34. SIGNAL-TO-NOISE RATIO TOTAL HARMONIC DISTORTION NOISE vs vs VOLTAGE GAIN FREQUENCY Figure 35. Figure 36.

www.ti.com Crosstalk − dB f − Frequency − Hz −60 −100 −150 100 1k 10k 20k −70 −80 −90 −110 −120 −130 −140 VDD = 3.3 V VO = 1 V RL = 10 kΩ CB = 1 µF IN2 to OUT1 IN1 to OUT2 Crosstalk − dB f − Frequency − Hz −60 −100 −150 100 1k 10k 20k −70 −80 −90 −110 −120 −130 −140 VDD = 5 V VO = 1 V RL = 10 kΩ CB = 1 µF IN1 to OUT2 IN2 to OUT1 Closed-Loop Gain − dB f − Frequency − Hz −10 100 1k 10k 1M 100k 200° 180° 160° 140° 120° 100° 80° Gain Phase Phase VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 CROSSTALK CROSSTALK vs vs FREQUENCY FREQUENCY Figure 37. Figure 38. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 39.

www.ti.com Closed-Loop Gain − dB f − Frequency − Hz −10 100 1k 10k 1M 100k Phase 200° 180° 160° 140° 120° 100° 80° Gain Phase VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V Closed-Loop Gain − dB f − Frequency − Hz −20 100 1k 10k 1M 100k Phase 200° 180° 160° 140° 120° 100° 80° 60° Gain Phase VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 40. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 41.

www.ti.com Closed-Loop Gain − dB f − Frequency − Hz −10 100 1k 10k 1M 100k Phase 200° 180° 160° 140° 120° 100° 80° Gain Phase VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V Closed-Loop Gain − dB f − Frequency − Hz −20 100 1k 10k 1M 100k Phase 200° 180° 160° 140° 120° 100° 80° 60° 40° Gain Phase VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 42. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 43.

www.ti.com f − Frequency − Hz −10 100 1k 10k 1M 100k Phase 200° 180° 160° 140° 120° 100° 80° Gain Phase VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V Closed-Loop Gain − dB Amplifier Power − mW Load Power − mW 80 120 180 200 14010020 6040 160 VDD = 3.3 V 8 Ω 16 Ω 64 Ω 32 Ω Amplifier Power − mW Load Power − mW 180 100 80 120 180 200 120 14010020 6040 160 140 160 VDD = 5 V 8 Ω 16 Ω 64 Ω 32 Ω TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY Figure 44. POWER DISSIPATION/AMPLIFIER POWER DISSIPATION/AMPLIFIER vs vs OUTPUT POWER OUTPUT POWER Figure 45. Figure 46.

www.ti.com APPLICATION INFORMATION GAIN SETTING RESISTORS, R F and R I Gain RF RI (1) Effective Impedance RFRI RF RI (2) fc(lowpass) 1 2RFCF (3) INPUT CAPACITOR C I fc(highpass) 1 2RICI (4) CI 1 2RIfc(highpass) (5) TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 The gain for the TPA122 is set by resistors R F and R I according to Equation Given that the TPA122 is an MOS amplifier, the input impedance is high. Consequently, input leakage currents are not generally a concern, although noise in the circuit increases as the value of R F increases. In addition, a certain range of R F values is required for proper start-up operation of the amplifier. Taken together, it is recommended that the effective impedance seen by the inverting node of the amplifier be set between k Ω and k Ω The effective impedance is calculated in Equation As an example, consider an input resistance of k Ω and a feedback resistor of k Ω The gain of the amplifier would be and the effective impedance at the inverting terminal would be k Ω which is within the recommended range. For high-performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of R F above k Ω the amplifier tends to become unstable due to a pole formed from R F and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately pF should be placed in parallel with R F In effect, this creates a low-pass filter network with the cutoff frequency defined in Equation For example, if R F is 100 k Ω and C F is pF, then f c(lowpass) is 318 kHz, which is well outside the audio range. In the typical application, an input capacitor, C I is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, C I and R I form a high-pass filter with the corner frequency determined in Equation The value of C I is important to consider, as it directly affects the bass (low-frequency) performance of the circuit. Consider the example where R I is k Ω and the specification calls for a flat bass response down to Hz. Equation is reconfigured as Equation In this example, C I is 0.4 µ so one would likely choose a value in the range of 0.47 µ F to µ A further consideration for this capacitor is the leakage path from the input source through the input network I C I and the feedback resistor F to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high-gain 10). For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications, as the dc level there is held at V DD /2, which is likely higher than the source dc level. Note that it is important to confirm the capacitor polarity in the application.

www.ti.com POWER SUPPLY DECOUPLING, C S MIDRAIL BYPASS CAPACITOR, C B CB 160 kΩ CIRI (6) OUTPUT COUPLING CAPACITOR, C C fc 1 2RLCC (7) TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 APPLICATION INFORMATION (continued) The TPA122 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µ placed as close as possible to the device V DD lead, works best. For filtering lower frequency noise signals, a larger aluminum electrolytic capacitor of µ F or greater placed near the power amplifier is recommended. The midrail bypass capacitor, C B serves several important functions. During start-up, C B determines the rate at which the amplifier starts up. This helps to push the start-up pop noise into the subaudible range (so low it can not be heard). The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier. The capacitor is fed from a 160-k Ω source inside the amplifier. To keep the start-up pop as low as possible, the relationship shown in Equation should be maintained. As an example, consider a circuit where C B is µ C I is µ and R I is k Ω Inserting these values into Equation results in: 6.25 which satisfies the rule. Bypass capacitor, C B values of 0.1- µ F to µ F ceramic or tantalum low-ESR capacitors are recommended for the best THD and noise performance. In the typical single-supply, single-ended (SE) configuration, an output coupling capacitor C is required to block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by Equation The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the low-frequency corner higher. Large values of C C are required to pass low frequencies into the load. Consider the example where a C C of µ F is chosen and loads vary from Ω to k Ω Table summarizes the frequency response characteristics of each configuration. Table Common Load Impedances vs Low Frequency Output Characteristics in SE Mode R L C C LOWEST FREQUENCY Ω µ F Hz 10,000 Ω µ F 0.23 Hz 47,000 Ω µ F 0.05 Hz As Table indicates, headphone response is adequate and drive into line level inputs home stereo for example) is good. The output coupling capacitor required in single-supply, SE mode also places additional constraints on the selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following relationship:

www.ti.com CB 160 kΩ CIRI RLCC (8) USING LOW-ESR CAPACITORS 5-V VERSUS 3.3-V OPERATION TPA122 SLOS211E AUGUST 1998 REVISED JUNE 2004 Low-ESR capacitors are recommended throughout this application. A real capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor behaves like an ideal capacitor. The TPA122 was designed for operation over a supply range of 2.5 V to 5.5 This data sheet provides full specifications for 5-V and 3.3-V operation because these are considered to be the two most common standard voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability. The most important consideration is that of output power. Each amplifier in the TPA122 can produce a maximum voltage swing of V DD This means, for 3.3-V operation, clipping starts to occur when V O(PP) 2.3 as opposed to V O(PP) V for 5-V operation. The reduced voltage swing subsequently reduces maximum output power into the load before distortion begins to become significant.

the integrated circuit (IC). at www.ti.com. See Figure 1 for DGN package exposed thermal die pad dimensions. NOTE: All linear dimensions are in millimeters. Figure 1. DGN Package Exposed Thermal Die Pad Dimensions PowerPAD is a trademark of Texas Instruments.

Orderable Device Status(1) Package Type Package Drawing Pins Package Qty Eco Plan(2) Lead/Ball FinishMSL Peak Temp (3) TPA122D ACTIVE SOIC D 8 75 Pb-Free (RoHS) CU NIPDAU Level-2-260C-1YEAR/ Level-1-220C-UNLIM TPA122DGN ACTIVE MSOP- Power PAD DGN 8 80 None CU NIPDAU Level-1-220C-UNLIM TPA122DGNR ACTIVE MSOP- Power PAD DGN 8 2500 None CU NIPDAU Level-1-220C-UNLIM TPA122DGNRG4 ACTIVE MSOP- Power PAD DGN 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPA122DR ACTIVE SOIC D 8 2500 Pb-Free (RoHS) CU NIPDAU Level-2-260C-1YEAR/ Level-1-220C-UNLIM TPA122EVM OBSOLETE 0 None Call TI Call TI (1)The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2)Eco Plan - May not be currently available - please checkhttp://www.ti.com/productcontentfor the latest availability information and additional product content details. None: Not yet available Lead (Pb-Free). Pb-Free (RoHS):TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Green (RoHS & no Sb/Br):TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens, including bromine (Br) or antimony (Sb) above 0.1% of total product weight. (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. PACKAGE OPTION ADDENDUM www.ti.com 21-Feb-2005 Addendum-Page 1

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