TPA0102PWP

TPA0102_16 TI1 | Alldatasheet

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Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. TPA0102 1.5-W STEREO AUDIO POWER AMPLIFIER SLOS166E – MARCH 1997 – REVISED MARCH 2000 1POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 /C0068High Power with PC Power Supply – 1.5 W/Ch at 5 V – 600 mW/Ch at 3 V /C0068Ultra-Low Distortion < 0.05% THD+N at 1.5 W and 4-W Load /C0068Bridge-Tied Load (BTL) or Single Ended (SE) Modes /C0068Stereo Input MUX /C0068Surface-Mount Power Package 24-Pin TSSOP PowerPAD  /C0068Shutdown Control...IDD < 10 mA C B C S Right MUX RLINEIN RHPIN Left MUX LHPIN LLINEIN Bias, Mute, Shutdown, and SE/BTL MUX Control RBYPASS MUTE IN MUTE OUT SHUTDOWN LBYPASS ROUT+ ROUT– RV DD LVDD LOUT+ LOUT– SE/BTL HP/LINE C IR R IR R FRC FR System Control C IL R IL NC NC R FLC FL 100 kW 100 kW VDD VDD C OUTR C OUTL 1 kW 0.1 mF GND/HS NC LOUT+ LLINEIN LHPIN LBYPASS LVDD SHUTDOWN MUTE OUT LOUT– MUTE IN GND/HS GND/HS NC ROUT+ RLINEIN RHPIN RBYPASS RV DD NC HP/LINE ROUT– SE/BTL GND/HS PWP PACKAGE (TOP VIEW) Copyright  2000, Texas Instruments IncorporatedPRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. PowerPAD is a trademark of Texas Instruments Incorporated.

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description

The TPA0102 is a stereo audio power amplifier in a 24-pin TSSOP thermal package capable of delivering greater than 1.5 W of continuous RMS power per channel into 4-W loads. This device functionality provides a very efficient upgrade path from the TPA4860 and TPA4861 mono amplifiers where three separate devices are required for stereo applications: two for speaker drive, plus a third for headphone drive. The TPA0102 simplifies design and frees up board space for other features. Full power distortion levels of less than 0.1% THD+N from a 5-V supply are typical. This provides significant improvement in fidelity for speech and music over the popular TPA4860/61 series. Low-voltage applications are also well served by the TPA0102 providing 600-mW per channel into 4-W loads with a 3.3-V supply voltage. Amplifier gain is externally configured by means of two resistors per input channel and does not require external compensation for settings of 2 to 20 in BTL mode (1 to 10 in SE mode). An internal input MUX allows two sets of stereo inputs to the amplifier. In notebook applications, where internal speakers are driven as BTL and the line (often headphone drive) outputs are required to be SE, the TPA0102 automatically switches into SE mode when the SE/BTL input is activated. Using the TPA0102 to drive line outputs up to 500 mW/channel into external 4 W loads is ideal for small non-powered external speakers in portable multimedia systems. The TPA0102 also features a shutdown function for power sensitive applications, holding the supply current below 5 mA. In speakerphone or other monaural applications, the TPA0102 is configured through the power supply terminals to activate only half of the amplifier which reduces supply current by approximately one-half over stereo applications. The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only in TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB applications. This allows the TPA0102 to operate at full power into 4-W loads at ambient temperature of up to 55°C. Into 8-W loads, the operating ambient temperature increases to 100°C. AVAILABLE OPTIONS PACKAGE TA TSSOP (PWP) 40°C to 85°C TPA0102PWP

1.5-W STEREO AUDIO POWER AMPLIFIER SLOS166E – MARCH 1997 – REVISED MARCH 2000 3POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 Terminal Functions TERMINAL I/O DESCRIPTION NAME NO. I/O DESCRIPTION GND/HS 1, 12, 13, 24 Ground connection for circuitry, directly connected to thermal pad HP/LINE 16 I Input MUX control input, hold high to select L/RHPIN (5, 20), hold low to select L/RLINEIN (4, 21) LBYPASS 6 Tap to voltage divider for left channel internal mid-supply bias LHP IN 5 I Left channel headphone input, selected when HP/LINE terminal (16) is held high LLINE IN 4 I Left channel line input, selected when HP/LINE terminal (16) is held low LOUT+ 3 O Left channel + output in BTL mode, + output in SE mode LOUT– 10 O Left channel – output in BTL mode, high-impedance state in SE mode LVDD 7 I Supply voltage input for left channel and for primary bias circuits MUTE IN 11 I Mute all amplifiers, hold low for normal operation, hold high to mute MUTE OUT 9 O Follows MUTE IN terminal (11), provides buffered output NC 2, 17, 23 No internal connection RBYPASS 19 Tap to voltage divider for right channel internal mid–supply bias RHP IN 20 I Right channel headphone input, selected when HP/LINE terminal (16) is held high RLINE IN 21 I Right channel line input, selected when HP/LINE terminal (16) is held low ROUT+ 22 O Right channel + output in BTL mode, + output in SE mode ROUT– 15 O Right channel – output in BTL mode, high impedance state in SE mode RV DD 18 I Supply voltage input for right channel SE/BTL 14 I Hold low for BTL mode, hold high for SE mode SHUTDOWN 8 I Places entire IC in shutdown mode when held high, IDD < I mA

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absolute maximum ratings over operating free-air temperature range (unless otherwise noted)† † Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATING TABLE PACKAGE TA ≤ 25°C DERATING FACTOR TA = 70°C TA = 85°C PWP 2.7 W‡ 21.8 mW/°C 1.7 W 1.4 W ‡ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (literature number SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD on page 33 of the before mentioned document. recommended operating conditions MIN NOM MAX UNIT Supply voltage, VDD 3 5 5.5 V O perating free air temperature TA VDD = 5 V, 4- W stereo BTL drive, 250 mW/ch average power, With proper PCB design –40 85 °COperating free-air temperature, TA VDD = 5 V, 4- W stereo BTL drive,

1.5 W/ch average power, With proper PCB design –40 55

Common mode input voltage VICM VDD = 5 V 1.25 4.5 VCommon mode input voltage, VICM VDD = 3.3 V 1.25 2.7 V dc electrical characteristics, TA = 25°C PARAMETER TEST CONDITIONS TYP † MAX UNIT Stereo BTL 19 25 mA VDD =5V Stereo SE 9 15 mA VDD = 5 V Mono BTL 9 15 mA IDD Supply current Mono SE 3 10 mA IDD Supply current Stereo BTL 13 20 mA VDD =33V Stereo SE 3 10 mA VDD = 3.3 V Mono BTL 3 10 mA Mono SE 3 10 mA VOO Output offset voltage (measured differentially) VDD = 5 V Gain = 2, See Note 1 5 25 mV IDD(MUTE) Supply current in mute mode VDD = 5 V 800 mA IDD(SD) IDD in shutdown VDD = 5 V 5 15 mA NOTE 1: At 3 V < VDD < 5 V the dc output voltage is approximately VDD /2.

1.5-W STEREO AUDIO POWER AMPLIFIER SLOS166E – MARCH 1997 – REVISED MARCH 2000 5POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 ac operating characteristics, VDD = 5 V, TA = 25°C, RL = 4 W PARAMETER TEST CONDITIONS MIN TYP MAX UNIT THD = 0.2%, BTL 1.25 W PO Outputpower (each channel) see Note 2 THD = 1%, BTL 1.5 W PO O utput pow er (each channel) see Note 2 THD = 0.2%, SE 500 mW THD = 1%, SE 600 mW THD+N Total harmonic distortion plus noise Po = 1 W, f = 20 to 20 kHz 200 m% BOM Maximum output power bandwidth G = 10, THD < 5 % >20 kHz BTL 72° Phase margin Open Load 71° SE 52° Power supply ripple rejection f = 1 kHz 75 dBPow er supply ripple rejection f = 20 – 20 kHz, 60 dB Mute attenuation 85 dB Channel-to-channel output separation f = 1 kHz 65 dB Line/HP input separation 100 dB BTL attenuation in SE mode 100 dB ZI Input impedance 2 M W Signal-to-noise ratio Po = 500 mW, BTL 95 dB Vn Output noise voltage 25 mV(rms) NOTE 2: Output power is measured at the output terminals of the IC at 1 kHz. ac operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 4 W PARAMETER TEST CONDITIONS MIN TYP MAX UNIT THD = 0.2% BTL 600 PO Outputpower (each channel) see Note 2 THD = 1% BTL 750 mWPO O utput pow er (each channel) see Note 2 THD = 0.2%, SE 200 mW THD = 1%, SE 250 THD+N Total harmonic distortion plus noise Po = 600 mW, f = 20 to 20 kHz 250 m% BOM Maximum output power bandwidth G = 10, THD < 5 % >20 kHz BTL 92° Phase margin Open Load 70° SE 57° Power supply ripple rejection f = 1 kHz 70 dBPow er supply ripple rejection f = 20 – 20 kHz 55 dB Mute attenuation 85 dB Channel-to-channel output separation f = 1 kHz 65 dB Line/HP input separation 100 dB BTL attenuation in SE mode 100 dB ZI Input impedance 2 M W Signal-to-noise ratio Po = 500 mW, BTL 95 dB Vn Output noise voltage 25 mV(rms) NOTE 2 Output power is measured at the output terminals of the IC at 1 kHz.

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Figure 1. BTL Test Circuit Figure 2. SE Test Circuit

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PO = 1.5 W PO = 0.25 W VDD = 5 V R L = 4 W AV = –2 V/V BTL 0.1 0.01 20 100 1 k 10 k 20 k THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY PO = 0.75 W Figure 6 f = 20 kHz f = 1 kHz f = 20 Hz 0.1 0.01 0.01 0.1 11 0 PO – Output Power – W VDD = 5 V R L = 4 W BTL THD+N –Total Harmonic Distortion + Noise – % TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER Figure 7 0.1 0.01 20 100 1 k 10 k 20 k THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY PO = 1 W VDD = 5 V R L = 8 W AV = –2 V/V BTL PO = 0.25 W PO = 0.5 W Figure 8 0.1 0.01 20 100 1 k 10 k 20 k THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY VDD = 5 V PO = 1 W R L = 8 W BTL AV = –2 V/V AV = –20 V/VAV = –10 V/V

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0.1 0.01 0.01 11 00.1 f = 20 kHz f = 1 kHz f = 20 Hz PO – Output Power – W VDD = 3.3 V R L = 4 W AV = –2 V/V BTL THD+N –Total Harmonic Distortion + Noise – % TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER Figure 14 0.1 0.01 20 100 1 k 10 k 20 k THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY AV = –20 V/V AV = –10 V/V AV = –2 V/V VDD = 3.3 V PO = 0.4 W R L = 8 W BTL Figure 15 PO = 0.4 W PO = 0.25 W PO = 0.1 W VDD = 3.3 V R L = 8 W AV = –2 V/V BTL 0.1 0.01 20 100 1 k 10 k 20 k THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY Figure 16 0.1 0.01 0.01 0.1 11 0 f = 20 kHz f = 1 kHz f = 20 Hz PO – Output Power – W VDD = 3.3 V R L = 8 W AV = –2 V/V BTL THD+N –Total Harmonic Distortion + Noise – % TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER

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0.1 0.01 20 100 1 k 10 k 20 k PO = 0.25 W PO = 0.05 W PO = 0.1 W THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY VDD = 5 V R L = 8 W SE Figure 22 0.1 0.01 0.001 0.1 PO – Output Power – W VDD = 5 V R L = 8 W AV = –2 V/V SE THD+N –Total Harmonic Distortion + Noise – % TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 0.01 f = 20 kHz f = 1 kHz f = 100 Hz Figure 23 0.1 0.01 20 100 1 k 10 k 20 k AV = –10 V/V AV = –5 V/V AV = –1 V/V THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY VDD = 5 V PO = 0.075 W R L = 32 W SE Figure 24 0.1 0.01 20 100 1 k 10 k 20 k PO = 75 mW PO = 25 mW PO = 50 mW THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY VDD = 5 V R L = 32 W SE

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AV = –10 V/V AV = –5 V/V AV = –1 V/V VDD = 3.3 V PO = 100 mW R L = 8 W SE 0.1 0.01 20 100 1 k 10 k 20 k THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY Figure 30 0.1 0.01 20 100 1 k 10 k 20 k PO = 25 mW PO = 50 mW PO = 100 mW THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY VDD = 3.3 V R L = 8 W SE Figure 31 VDD = 3.3 V R L = 8 W SE 0.1 0.01 0.001 0.1 PO – Output Power – W THD+N –Total Harmonic Distortion + Noise – % TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER f = 20 kHz f = 1 kHz f = 100 Hz 0.01 Figure 32 0.1 0.01 20 100 1 k 10 k 20 k AV = –10 V/V AV = –5 V/V AV = –1 V/V THD+N –Total Harmonic Distortion + Noise – % f – Frequency – Hz TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY VDD = 3.3 V PO = 30 mW R L = 32 W SE

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–50 –60 –80 –100 20 100 1 k Supply Ripple Rejection Ratio – dB –30 –20 f – Frequency – Hz SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY 10 k 20 k –10 –40 –70 –90 VDD = 5 V VDD = 3.3 V R L = 4 W C B = 4.7 mF BTL Figure 38 –50 –60 –80 –100 20 100 1 k –30 –20 f – Frequency – Hz SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY 10 k 20 k –10 –40 –70 –90 VDD = 5 V VDD = 3.3 V R L = 4 W C B = 4.7 mF SE Supply Ripple Rejection Ratio – dB Figure 39 Left to Right Right to Left –80 –90 –110 –120 –60 –50 –40 –70 –100 20 100 1 k Crosstalk – dB f – Frequency – Hz CROSSTALK vs FREQUENCY 10 k 20 k VDD = 5 V PO = 1.5 W R L = 4 W BTL Figure 40 –80 –90 –110 –120 –60 –50 –40 –70 –100 20 100 1 k Crosstalk – dB f – Frequency – Hz CROSSTALK vs FREQUENCY 10 k 20 k VDD = 3.3 V PO = 0.75 W R L = 4 W BTL Left to Right Right to Left

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–20 –40 80 180° 90° –90° –180° 0.01 Gain – dB f – Frequency – kHz OPEN LOOP RESPONSE 0.1 1 10 100 1000 10000 VDD = 3.3 V BTL Gain Figure 44 Phase Phase 20 100 1 k 10 k Gain – dB f – Frequency – Hz CLOSED LOOP RESPONSE 100 k 200 k –45° –90° –135° –180° –225° –270° Phase Phase Gain VDD = 5 V AV = –2 V/V PO = 1.5 W BTL Figure 45

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–10 20 100 1 k 10 k Gain – dB f – Frequency – Hz CLOSED LOOP RESPONSE 100 k 200 k –45° –90° –135° –180° –225° –270° Phase VDD = 3.3V AV = –1 V/V PO = 0.25 W SE Phase Gain Figure 48 Figure 49 Stereo BTL SUPPLY CURRENT vs SUPPLY VOLTAGE 46 5 VDD – Supply Voltage – V Stereo SE – Supply Current – mA ÁÁ ÁÁ IDD Figure 50 1.5 0.5 2.5 3 3.5 4 4.5 5 2.5 5.5 6 R L = 4 W R L = 8 W – Output Power – WPO OUTPUT POWER vs SUPPLY VOLTAGE VDD – Supply Voltage – V THD+N = 1% BTL Each Channel

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0.3 0.2 0.1 0 0.25 0.5 – Power Dissipation – W 0.4 0.5 POWER DISSIPATION vs OUTPUT POWER 0.6 0.75 1 PO – Output Power – W PD R L = 4 W R L = 8 W VDD = 3.3 V BTL Each Channel Figure 56 0.4 0.2 0 0.1 0.2 0.3 0.6 0.8 0.4 0.5 0.6 – Power Dissipation – W POWER DISSIPATION vs OUTPUT POWER PO – Output Power – W PD R L = 4 W R L = 8 W VDD = 5 V SE Each Channel R L = 32W Figure 57 0.2 0 0.05 0.1 0.15 0.4 0.6 0.2 0.25 – Power Dissipation – W POWER DISSIPATION vs OUTPUT POWER PO – Output Power – W PD R L = 4 W R L = 8 W VDD = 3.3V SE Each Channel R L = 32W

to provide an effective thermal contact between the IC and the PWB. severely limits the usable range of many high-performance analog circuits. performance comparable to much larger power packages. Figure 58. Views of Thermally Enhanced PWP Package

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APPLICATION INFORMATION

bridged-tied load versus single-ended mode Figure 59 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA0102 BTL amplifier consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive configuration but initially consider power to the load. The differential drive to the speaker means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the load as compared to a ground referenced load. Plugging 2 × V O(PP) into the power equation, where voltage is squared, yields 4× the output power from the same supply rail and load impedance (see equation 1). Power /C0043 V (rms) R L (1) V (rms)/C0043 V O(PP) 22/C0504 R L 2x VO(PP) VO(PP) –VO(PP) VDD VDD Figure 59. Bridge-Tied Load Configuration which is loudness that can be heard. In addition to increased power there are frequency response concerns. created with the speaker impedance and the coupling capacitance and is calculated with equation 2.

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Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different. Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform. The following equations are the basis for calculating amplifier efficiency. I DD rms /C0043 2V P /C0112R L P SUP /C0043V DD IDD rms /C0043 V DD 2V P /C0112R L Efficiency/C0043 P L P SUP Efficiency of a BTL Configuration/C0043 /C0112V P 2V DD /C0043 /C0112/C0466 PLR L 2 /C0467 1/C03242 2V DD (3) Where: (4) P L /C0043 V Lrms2 R L /C0043 V p 2R L V Lrms /C0043 V P 2/C0504 Table 1 employs equation 4 to calculate efficiencies for four different output power levels. Note that the efficiency of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at full output power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper power supply design. For a stereo 1-W audio system with 8-W loads and a 5-V supply, the maximum draw on the power supply is almost 3.25 W. Table 1. Efficiency Vs Output Power in 5-V 8-W BTL Systems † High peak voltages cause the THD to increase. indicates that as VDD goes down, efficiency goes up.

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Bias, Mute, Shutdown, and SE/BTL MUX Control RBYPASS MUTE IN MUTE OUT SHUTDOWN LBYPASS ROUT+ ROUT– RV DD LVDD LOUT+ LOUT– SE/BTL HP/LINE C IRLINE R IRLINE R FRLINEC FRLINE System Control C ILLINE R ILLINE R FLLINEC FLLINE 100 kW 100 kW VDD C OUTR C OUTL 1 kW C IRHP R IRHP R FRHP See Note A C BL C ILHP R ILHP R FLHP C SR VDD 0.1 mF NOTE A: This connection is for ultralow current in shutdown mode. Figure 63. TPA0102 Full Configuration Application Circuit The gain for each audio input of the TPA0102 is set by resistors RF and RI according to equation 5 for BTL mode.

1.5-W STEREO AUDIO POWER AMPLIFIER SLOS166E – MARCH 1997 – REVISED MARCH 2000 29POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 gain setting resistors, RF and RI (continued) BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the voltage swing across the load. Given that the TPA0102 is a MOS amplifier, the input impedance is very high, consequently input leakage currents are not generally a concern although noise in the circuit increases as the value of RF increases. In addition, a certain range of RF values are required for proper startup operation of the amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5 kW and 20 kW . The effective impedance is calculated in equation 6. (6)Effective Impedance/C0043 R FR I R F /C0041R I As an example consider an input resistance of 10 kW and a feedback resistor of 50 kW . The BTL gain of the amplifier would be –10 and the effective impedance at the inverting terminal would be 8.3 kW , which is well within the recommended range. For high performance applications metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of RF above 50 kW the amplifier tends to become unstable due to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than 50 kW . This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7. (7)fc(lowpass)/C00431 2/C0112R F C F –3 dB fc For example, if RF is 100 kW and Cf is 5 pF then fc is 318 kHz, which is well outside of the audio range. input capacitor, CI In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency determined in equation 8. (8)fc(highpass)/C00431 2/C0112R IC I –3 dB fc

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input capacitor, CI (continued) The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit. Consider the example where RI is 10 kW and the specification calls for a flat bass response down to 40 Hz. Equation 8 is reconfigured as equation 9. (9)C I /C00431 2/C0112R Ifc In this example, CI is 0.40 mF so one would likely choose a value in the range of 0.47 mF to 1 mF. A further consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications as the dc level there is held at V DD /2, which is likely higher than the source dc level. Please note that it is important to confirm the capacitor polarity in the application. power supply decoupling, CS The TPA0102 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 mF placed as close as possible to the device V DD lead works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 mF or greater placed near the audio power amplifier is recommended. midrail bypass capacitor, CB The midrail bypass capacitor, CB, serves several important functions. During startup or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier. The capacitor is fed from a 25-kW source inside the amplifier. To keep the start-up pop as low as possible, the relationship shown in equation 10 should be maintained. (10) /C0466C B /C003225 kW /C0467 /C01181 /C0466C IR I/C0467 As an example, consider a circuit where CB is 0.1 mF, CI is 0.22 mF and RI is 10 kW . Inserting these values into the equation 10 we get 400 ≤ 454 which satisfies the rule. Bypass capacitor, CB, values of 0.1 mF to 1 mF ceramic or tantalum low-ESR capacitors are recommended for the best THD and noise performance. In Figure 63, the full feature configuration, two bypass capacitors are used. This provides the maximum separation between right and left drive circuits. When absolute minimum cost and/or component space is required, one bypass capacitor can be used as shown in Figure 62. It is critical that terminals 6 and 19 be tied together in this configuration.

1.5-W STEREO AUDIO POWER AMPLIFIER SLOS166E – MARCH 1997 – REVISED MARCH 2000 31POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 In SE mode (see Figure 59 and Figure 60), the load is driven from the primary amplifier output for each channel (OUT+, terminals 22 and 3). In SE mode the gain is set by the RF and RI resistors and is shown in equation 11. Since the inverting amplifier is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included. (11)SE Gain /C0043/C0042/C0466 R F R I /C0467 The output coupling capacitor required in single-supply SE mode also places additional constraints on the selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of the following relationship: (12) /C0466C B /C003225 kW /C0467 /C01181 /C0466C IR I/C0467 /C04161 R LC C output coupling capacitor, CC In the typical single-supply SE configuration, an output coupling capacitor (CC ) is required to block the dc bias at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by equation 13. (13)f c(high)/C00431 2/C0112R LC C –3 dB fc The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives the low-frequency corner higher degrading the bass response. Large values of CC are required to pass low frequencies into the load. Consider the example where a CC of 330 mF is chosen and loads vary from 4 W , 8 W , 32 W, to 47 kW . Table 2 summarizes the frequency response characteristics of each configuration. Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode

4 W 330 mF 120 Hz

8 W 330 mF 60 Hz

32 W 330 mF 15 Hz

headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

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The ability of the TPA0102 to easily switch between BTL and SE modes is one of its most important cost saving features. This feature eliminates the requirement for an additional headphone amplifier in applications where internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated. Internal to the TPA0102, two separate amplifiers drive OUT+ and OUT–. The SE/BTL input (terminal 14) controls the operation of the follower amplifier that drives LOUT– and ROUT– (terminals 10 and 15). When SE/BTL is held low, the amplifier is on and the TPA0102 is in the BTL mode. When SE/BTL is held high, the OUT– amplifiers are in a high output impedance state, which configures the TPA0102 as an SE driver from LOUT+ and ROUT+ (terminals 3 and 22). I DD is reduced by approximately one-half in SE mode. Control of the SE/BTL input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in Figure 64. MUX RLINE IN RHP IN + Bypass SE/BTL HP/LINE R m2 100 kW R m1 100 kW VDD C OUTR R m3 1 kW – ROUT– 15 ROUT+ 22 Left Channel 0.1 mF Figure 64. TPA0102 Resistor Divider Network Circuit O ) into the headphone jack. benefits of doing this are described in the following input MUX operation section.

1.5-W STEREO AUDIO POWER AMPLIFIER SLOS166E – MARCH 1997 – REVISED MARCH 2000 33POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 Working in concert with the SE/BTL feature, the HP/LINE MUX feature gives the audio designer the flexibility of a multichip design in a single IC (see Figure 65). The primary function of the MUX is to allow different gain settings for BTL versus SE mode. Speakers typically require approximately a factor of 10 more gain for similar volume listening levels as compared to headphones. To achieve headphone and speaker listening parity, the resistor values would need to be set as follows: (14)SE Gain (HP) /C0043/C0042/C0466 R F(HP) R I(HP) /C0467 If, for example RI(HP) = 20 kW and RF(HP) = 20 kW then SE Gain(HP) = –1 (15)BTL Gain(LINE) /C0043/C00422 /C0466 R F(LINE) R I(LINE) /C0467 If, for example RI(LINE) = 20 kW and RF(LINE) = 100 kW then BTL Gain(LINE) = –10 ROUT+ ROUT– C IRLINE R IRLINE C IRHP R IRHP R FRHP MUX RLINE IN RHP IN SE/BTL HP/LINE VDD Left Channel Right ChannelMID R FRLINE 0.1 mF Figure 65. TPA0102 Example Input MUX Circuit operation section for a description of the headphone jack control circuit.

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The TPA0102 employs both a mute and a shutdown mode of operation designed to reduce supply current, IDD , to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal should be held low during normal operation when the amplifier is in use. Pulling SHUTDOWN high causes the outputs to mute and the amplifier to enter a low-current state, I DD < 1 mA. SHUTDOWN or MUTE IN should never be left unconnected because amplifier operation would be unpredictable. Mute mode alone reduces I DD < 1 mA. Table 3. Shutdown and Mute Mode Functions † Inputs should never be left unconnected. resistance the more the real capacitor behaves like an ideal capacitor. special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability goes. before distortion becomes significant. approximately two-thirds the supply power for a given output-power level than operation from 5-V supplies. in battery-powered applications.

1.5-W STEREO AUDIO POWER AMPLIFIER SLOS166E – MARCH 1997 – REVISED MARCH 2000 35POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 headroom and thermal considerations Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions. A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion as compared with the average power output. From the TPA0102 data sheet, one can see that when the TPA0102 is operating from a 5-V supply into a 4-W speaker that 1.5 W peaks are available. Converting watts to dB: P dB /C004310Log /C0466 P W P ref /C0467 /C004310Log /C04661.5 1 /C0467 /C00431.76 dB Subtracting the headroom restriction to obtain the average listening level without distortion yields: 1.76 dB/C004215 dB /C0043/C004213.24 dB (15 dB headroom) 1.76 dB/C004212 dB /C0043/C004210.24 dB (12 dB headroom) 1.76 dB/C00429d B /C0043/C00427.24 dB (9 dB headroom) 1.76 dB/C00426d B /C0043/C00424.24 dB (6 dB headroom) 1.76 dB/C00423d B /C0043/C00421.24 dB (3 dB headroom) Converting dB back into watts: P W /C004310PdB /C032410 /C0032P ref /C004347 mW (15 dB headroom) /C004394 mW (12 dB headroom) /C0043188 mW (9 dB headroom) /C0043376 mW (6 dB headroom) /C0043752 mW (3 dB headroom) This is valuable information to consider when attempting to estimate the heat dissipation requirements for the amplifier system. Comparing the absolute worst case, which is 1.5 W of continuous power output with 0 dB of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the system. Using the power dissipation curves for a 5-V, 4-W system, the internal dissipation in the TPA0102 and maximum ambient temperatures is shown in Table 4. Table 4. TPA0102 Power Rating, 5-V, 4-W , Stereo

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headroom and thermal considerations (continued) DISSIPATION RATING TABLE PACKAGE TA ≤ 25°C ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ DERATING FACTOR TA = 70°C TA = 85°C ÁÁÁÁÁÁ ÁÁÁÁÁÁ PWP † ÁÁÁÁÁÁ ÁÁÁÁÁÁ 2.7 W ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ 21.8 mW/°C ÁÁÁÁÁÁ ÁÁÁÁÁÁ 1.7 W ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ 1.4 W PWP ‡ 2.8 W ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ 22.1 mW/°C 1.8 W 1.4 W † This parameter is measured with the recommended copper heat sink pattern on a 1-layer PCB, 4 in2 5-in × 5-in PCB, 1 oz. copper, 2-in × 2-in coverage. ‡ This parameter is measured with the recommended copper heat sink pattern on an 8-layer PCB, 6.9 in2 1.5-in × 2-in PCB, 1 oz. copper with layers 1, 2, 4, 5, 7, and 8 at 5% coverage (0.9 in2) and layers 3 and 6 at 100% coverage (6 in2). The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM and 300 CFM data from the dissipation rating table, the derating factor for the PWP package with 6.9 in2 of copper area on a multilayer PCB is 22 mW/°C and 54 mW/°C respectively. Converting this to Q JA: Q JA /C00431 Derating /C00431 0.022 /C004345°C /C0324W To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are per channel so the dissipated heat needs to be doubled for two channel operation. Given Q JA, the maximum allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be calculated with the following equation. The maximum recommended junction temperature for the TPA0102 is 150 °C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs. T A Max /C0043TJ Max /C0042Q JA P D /C0043150 /C004245(0.4/C00322) /C0043114°C (15 dB headroom, 0 CFM) NOTE: Internal dissipation of 0.4 W is estimated for a 1.5-W system with 15 dB headroom per channel. Table 4 shows that for most applications no airflow is required to keep junction temperatures in the specified range. The TPA0102 is designed with thermal protection that turns the device off when the junction temperature surpasses 150°C to prevent damage to the IC. Table 4 was calculated for maximum listening volume without distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-W speakers dramatically increases the thermal performance by increasing amplifier efficiency.

www.ti.com 24-Jan-2013 Addendum-Page 1 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3) Op Temp (°C) Top-Side Markings (4) Samples TPA0102PWP ACTIVE HTSSOP PWP 24 60 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA0102 TPA0102PWPG4 ACTIVE HTSSOP PWP 24 60 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA0102 TPA0102PWPLE OBSOLETE HTSSOP PWP 24 TBD Call TI Call TI TPA0102PWPR ACTIVE HTSSOP PWP 24 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA0102 TPA0102PWPRG4 ACTIVE HTSSOP PWP 24 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA0102 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) Only one of markings shown within the brackets will appear on the physical device. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

www.ti.com 24-Jan-2013 Addendum-Page 2 In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

*All dimensions are nominal Device Package Type Package Drawing Pins SPQ Reel Diameter (mm) Reel Width W1 (mm) (mm) (mm) (mm) (mm) W (mm) Pin1 Quadrant PACKAGE MATERIALS INFORMATION www.ti.com 14-Jul-2012 Pack Materials-Page 1

*All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) TPA0102PWPR HTSSOP PWP 24 2000 367.0 367.0 38.0 PACKAGE MATERIALS INFORMATION www.ti.com 14-Jul-2012 Pack Materials-Page 2

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