SPB03N60S5

AN75 DIODES | Alldatasheet

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Issue 1 – January 2011 2 www.diodes.com © Diodes Incorporated 2010 sense signal and the switching adjusts to be the sum of the on- and off-time. This change allows the converter to work with duty cycles greater than 50%. Design Guide – High power factor offline LED driver In this section the design procedure is outlined acco rding to the schematic shown in Figure 1. First, the guideline for selecting the comp onents for valley fill power factor correction stag e and fixed off- time buck converter is shown. The power inductor calculation is then demonstrated and finally, the power losses within MOSFET and free-wheel diode are assessed. The specifications for the system are: VAC = 230Vac VAC(min) = 85Vac VAC(max)= 264Vac ILED(nom) = 240mA VLED(nom) = 54V VLED(min) = 42V VLED(max) = 59V POUT = 12.96W fswi(nom) = 55kHz Passive factor correction stage design The purpose of the valley fill circuit (see Figure 2) is to allow the buc k converter to pull power directly off the AC line when the line voltage is greater than 50% of its peak voltage. Figure 2 Valley-fill PFC stage and operating waveforms (Green: VIN to LED driver; Orange: AL9910’s gate voltage) The maximum bus voltage at the input of the buck converter is, V373Vac6422V2V ac(max)IN(max) =×=×= During this time, capacitors within the valley fill circuit (C1 and C2) are in series and charged via D2 and R1. If the capacitors have identical capacit ance value, the peak voltage across C1 and C2

Issue 1 – January 2011 3 www.diodes.com © Diodes Incorporated 2010 is V1862VIN(max) = . Often a 20% difference in capaci tance could be observed between like capacitors. Therefore a voltage rating margin of 25% should be considered. Once the line drops below 50% of its peak voltag e, the two capacitors are essentially placed in parallel. The bus voltage V IN(min) is the lowest voltage value at the input of the buck converter. VIN(min) at the minimum AC line voltage Vac(min) is, V602Vac8522V2V ac(min)IN(min) =×=×= At 60Hz, the total time of a half AC line cycle is 8.33ms. The power to the buck converter is derived from the valley-fill capacitors when the AC line voltage is equal to or less than 50% of its peak voltage. The hold up time for the capacitors equates to ms77.2ms33.831tHOLD =×= . The valley-fill capacitor value can then be calculated, Fμ30V20 ms77.2V60W96.12 V tV P C DROOP HOLD(min)IN out TOTAL = Therefore, Fμ152C1C == . VDROOP is the voltage droop on the capacito rs when they are delivering full power to the buck c onverter. Ideally VDROOP should be set to less than (max)LED(min)INDROOP VVV −= in order to ensure continuous LED conducti on at low line voltage. Nevertheless, V DROOP is set to be 20V in the design example to avoid the need for very large valley-fill electrolytic capacitor. A 20V V DROOP implies that the bus voltage V IN at the input of buck converter will drop to 40V during part of the AC line cycle. As the buck regulator requires V IN to be greater than the LED stack voltage (VLED(max)=59V) for regulation, the LED will be off during part of the AC line cycle. This has the effect of reducing the actual output LED current at low AC input voltage. In the design example, the LED current drops by approximately 20% from its nominal value at 85Vac (see Figure 4). Setting the fixed off-time and switching frequency range For fixed off-time operation, the switching frequency will vary subjected to the actual input voltage and output LED conditions. A nominal switching frequency f swi(nom) should be chosen. A high nominal switching frequency will result in smaller inductor size, but could lead to increased switching losses in the circuit. A good design practice is to choose a nominal switching frequency knowing that the switching frequency will decrease as the line voltage drops and increases as the line voltage increases. The fixed off-time tOFF can be computed as, sμ9.1355kHz 230V 54V-1 f V V-1 t swi(nom) ac(nom) LED(nom) off === The off-time is programmed by timing resistor RT as shown in Figure 1. The value of RT is given by, () () Ω=−×=−×=Ω k32622259.132225sμtkR OFFT A 330k Ω is selected for R T. Next, the two extremes of the variable switching frequency can be approximated as, kHz10sμ9.13 V69V591 t VV1f OFF (min)IN(max)LED swi(min) =−=−= kHz8.63sμ9.13 V373V421 t VV1f OFF (max)IN(min)LED swi(max) =−=−=

Issue 1 – January 2011 4 www.diodes.com © Diodes Incorporated 2010 It is advisable to keep below the maximum switching frequency f swi(max) below 150kHz to avoid excessive switching loss. Inductor selection and setting the LED current The fixed off-time architecture of the AL9910 regulates the average current through the inductor LBUCK. The value of LBUCK depends on the desirable peak-to-peak ripple ΔIL in the output LED current. L BUCK can be set with the following equation, mH6.6mA115 sμ9.13V54 I tVL L OFF)nom(LED BUCK =×=Δ Due to diameter limitation of the T8 tube, LBUCK is made up of L3 and L4 as shown in Figure 1. The AL9910 constant off-time control loop regulates the peak inductor current I pk. As the average inductor current equals the average LED curren t, the average LED current can be regulated by controlling Ipk. Given a fixed inductor value, the change in the inductor current over time is proportional to the voltage applied across the inductor. During the off-time, t he voltage seen by the inductor is the LED stack voltage. So, the peak inductor current should be regulated to, mA297mH6.6 sμ9.13V545.0mA240L tV5.0II BUCK OFF)nom(LED )nom(LEDpk =××+=××+= The peak current is constant and set by the sense resistor R SENSE. If the LD pin is tied to the VDD pin, the value of RSENSE can be easily calculated because the voltage threshold on the CS pin is 0.25V, Ω== 84.0mA297 25.0RSENSE In the circuit shown in Figure 1, RSENSE consists of R5, R6 and R7. The peak current rating of the L BUCK should be greater than I pk and the RMS current rating of the inductor should be at least 110% of ILED(nom). Although the described solution, working in fixed o ff-time and Continuous Conduction Mode (CCM), works as a constant current source, a limitation to the output LED current accuracy is its dependency on the number of LEDs and overall LED chain voltage. The best result can be achieved using a fixed number of LEDs. A variable number of LEDs results in reduced current precision. The two extremes of the output LED current can be approximated as, mA234mH6.6 sμ9.13V595.0-mA972L tV5.0-II BUCK OFF(max)LED pkLED(min) =××=××= mA253mH6.6 sμ9.13V425.0-mA972L tV5.0-II BUCK OFF(min)LED pkLED(max) =××=××= The above equation shows that the precision of t he LED current also depends on the tolerance of practical inductor L BUCK. Inductor with tolerance rating equal or less than 10% should be chosen to ensure good LED current precision at mass production. Power MOSFET calculation The power MOSFET is chosen based on maximum voltage stress, peak MOSFET current, total power losses, maximum allowable working temperature and the gate driver capability of the AL9910.

Issue 1 – January 2011 5 www.diodes.com © Diodes Incorporated 2010 Maximum drain-source voltage stress on the power MO SFET for this converter is equal to the input voltage. However, a typical voltage safety margin for the MOSFET defines the maximum reverse voltage as follows, V485V3733.1V3.1V (max)INDSS =×=×= which implies that a common 500V MOSFET is suitable. The power MOSFET losses will be dominated by sw itching loss. The switching loss depends on the switching time, frequency, MOSFET drain current a nd drain-source voltage. The switching rise time tRISE and fall time tFALL is a function of the MOSFET’s gate capacitance, the gate driver capability of the AL9910 and layout design. The worse case switching power losses occurs at V LED(min) and V IN(max). The switching loss is approximately, 455mW 63.8kHz65ns373V 63.8kHz65ns88mA297mA373V ftIV ftL tVIV P swi(max)FALLpkIN(max) swi(max)RISE BUCK OFFLED(min) pkIN(max) SW ×××+ ××⎟⎟ ⎛ −× where the switching time t RISE and t FALL are measured to be 65ns with the 600V MOSFET SPB03N60S5 as the power MOSFET. As shown in Figure 1, R10 is a series gate resistor that slows down the MOSFET switching and reduces EMI emission. The RMS current through the MOSFET at VLED(min) and VIN(max) is given by, mA89 mH6.6sμ9.13V42mA240V373 V42 LtVIV VI BUCKOFF(min)LED )nom(LED (max)IN (min)LED D(RMS) ⎛ ×+×= ⎛ ×+×= The power MOSFET conduction loss depends on its static drain-source resistance R DS(ON) at the MOSFET working temperature. It is possible to calculate the continuous conduction loss: () mW195.2mA89RIP 2 DS(ON) D(RMS)COND =Ω×=×= The total power MOSFET loss is: mW474mW19mW455PPP CONDSWTOT =+=+= Total MOSFET power loss is dissipated from the SM D package into the PC Board. So it is possible to calculate the MOSFET working junction temperat ure can be calculated if the package junction-to- ambient thermal resistance RthJA is known. The calculated MOSFET junction temperature, TJ, must be lower then the maximum allowable junction temperature TJ(MAX): C4.109C80WC62mW474TθPT AMBthJATOTJ ooo =+×=+×= The internal ambient temperature within the LED converter, T AMB, is assumed to be 80ºC. θthJA = WC62 o is the thermal resistance for TO-263 with minimum copper area. For practical design, it is recommended to keep the junction temperature below 110ºC to avoid temperature stress on the device.

Issue 1 – January 2011 6 www.diodes.com © Diodes Incorporated 2010 Free-wheel diode calculation The free-wheel diode D F shown in Figure 1 is chosen based on its maximum stress voltage and total power loss. The maximum stress voltage rating of the free-wheel diode is the same as the MOSFET. It is advisable to use ultra-low reverse recovery time T RR (<35ns) diode as D F to reduce the MOSFET’s switching ON loss. In the design example, 1A 600V rectifier, MUR160, is selected. The worst case average current through the diode occurs at VLED(max) and VIN(min). mA202V373 V421mA240V V1II (max)IN (min)LED )nom(LEDD(avg) =⎟ −×= Assuming a constant forward voltage drop V F across the diode, the conduction power loss can be calculated, mW222V1.1mA202VIP F)avg(DCOND_D =×=×= Finally, the diode junction temperature without using the heat sink can be calculated from, C87C80WC32mW222TθPT AMBthJACOND_Dj ooo =+×=+×= The internal ambient temperature within the LED converter, T AMB, is assumed to be 80ºC. θthJA = WC32 o is the thermal resistance for DO-201 package. For practical design, it is recommended to keep the junction temperature below 110ºC to avoid temperature stress on the device.

Issue 1 – January 2011 7 www.diodes.com © Diodes Incorporated 2010 The BOM in table 1 and the PCB layout in Figure 2 complete the tools needed to design a high power factor LED driver using the AL9910. Figure 3 shows the picture of the completed LED driver designed with a footprint to fit inside the T8 LED Fluorescent replacement lamp tube. Table 1 BOM Ref. Descriptions Part number Package Mfr. U1 Universal high brightness LED driver AL9910 SO8 Diodes Inc. D1, D2, D3 1A, 1kV diode t RR = 1.8μs S1M-13-F SMA Diodes Inc. D4 Ultra-fast-recovery diode 1A, 600V, tRR = 35ns MUR160 DO201AD Diodes Inc. DB1 1A, 600V bridge rectifier DF06S DF-S Diodes Inc. C1, C2 15 μF, 450V electrolytic capacitor +/-20% 1000hrs @ 105ºC EEUED2W150 400KXW27M10X30 UCY2G150MPD 5mm pitch Panasonic Rubicon Nichicon C4 4.7 μF, 50V electrolytic capacitor +/-20% 1000hrs @ 105ºC ECE-A1HKG4R7 1.5mm pitch Panasonic C5 10 μF 450V electrolytic capacitor +/-20% 1000hrs @ 105ºC, 10mm diameter EEUEE2W100U 5mm pitch Panasonic CX1, CX2, CX3, CX4 100nF, 275VAC, Film, X2 ECQU2A104ML 17.5mm pitch Panasonic F1 10Ohm 1W fusible resistor +/-200ppm NFR0100001009JR500 Through-hole axial Vishay L1 6.8mH inductor +/-10% 290mA radial 19R685C 5mm pitch Murata L2 30mH common-mode inductor, 8mm height B82791G2301N001 10mm pitch EPCOS L3, L4 3.3mH inductor +/-10% 420mA radial 19R335C 6mm pitch Murata MOV1 275V, 21J, 9mm, Radial B72207S0271K101 5mm pitch EPCOS Q1 N-ch MOSFET 600V, 3.2A, Qg(max) = 16nC SPB03N60S5 TO263 Infineon R1 10R 3W wire wound resistor, 50ppm/ºC, +/-1% UB3C-10RF1 Through-hole axial Riedon R2 3k 0.25W resistor +/-5% Any 1206 Any R5 1R2 0.25W +/-1% Any 1206 Any R6 2R7 0.25W +/-1% Any 1206 Any R7 100R 0.25W +/-1% Any 1206 Any RT 330k 0.125W resistor +/- Any 1206 Any R10 10R 0.25W +/-5% Any 1206 Any

Issue 1 – January 2011 8 www.diodes.com © Diodes Incorporated 2010 Figure 2 Top layer and bottom layer layout

Issue 1 – January 2011 9 www.diodes.com © Diodes Incorporated 2010 Figure 3 Picture of the LED T8 Fluorescent replacement lamp driver

Issue 1 – January 2011 11 www.diodes.com © Diodes Incorporated 2010 Figure 6 LED driver power factor Conclusion This application note provides a simple tool to design an offline LED driver using the AL9910 high voltage LED controller. It provides a high level of effi ciency as well as LED current control over a wide range of input voltages. Moreover the document expl ains how to design a system with passive power factor correction to achieve PF greater than 0.7, allowing compliant with emergent international solid state lighting standards.

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