OP260

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; DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER FEATURES PIN CONNECTIONS © -3dB Bandwidth (Ay=#1) ovecsssneenreecenee SOMHZ Typ ve © Bandwidth Independent of Gain 2 outa Lf orn * Unity-Gain Stable ouras youre ma ch A Aine * Available in Die Form AINA2 6 -INB vf FF) sme + +A 3) 5 NB ORDERING INFORMATION 2 EPOXY MINI-DIP Tyss2sc_ PACKAGE OPERATING (ease) suns) VosMAX CERDIP tcc TEMPERATURE “mv) «8 PIN TO-99 PLASTIC _20-CONTACT _ RANGE TO-99 (2-Suffix)

35 OP260AZ OP260A - © OP260ARC83 ML (Suffix) 6 5 a4

35 OP260EZ OP260EJ - - XIND 23zs2

OP260FZ J - - EN) EN} ESE) 50 OP2SOFZ OP26OF) cp : nD OY pao xe fa Payne. 70 - -___op2socstt - XIND. +INA a> [75] nc. -watEh DA -zjoure For devices processed in total compliance to MIL-STD-883, add /883 after part ne. G} [a] nc. Nc. [S| ig: ne. number. Consult factory for 883 data sheet. va Fa) ve +a F} fis] 8 t Burn-in is available on commercial and industrial temperature range parts in aH Fa] nc. ne. fy fe] nc. CerDIP, plastic DIP, and TO-can packages. For ordering information, see Nc 7 olor PMfs Data Book, Section 2. +weCel fii} nc. -eLebels ‘For availability and burn-in information on SO and PLCC packages, contact Ne rt f> Fa) our 8 gigzg your local sales office. Ne. qe fh ne. + 20-CONTACT GENERAL DESCRIPTION “eownn) Hee outtay iw The dual OP-260 represents anew conceptin monolithic operational amplifiers. Built on PMI's high-speed bipolar process, the OP-260 continued SIMPLIFIED SCHEMATIC (One of Two Ampifiers) Ov r) © Vour | : | © 33" 208 ov 5/90, Rev. C1

PMI) OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER employs current feedback to provide consistently wideband opera- Operating Temperature Range tion regardless of gain. The OP-260's —3dB bandwidth of 90MHz at OP-260A, (J, RC, Z) snvsveersntsctteeeneenennB5°C t0+125°C™ A,=+1 combines with a slew rate of 1000V/us for extremely high- OP-260E/F (J, Z) eevsseccssssseetseseesssssssnenessssessee 40°C to +85°C i 0 0 TheOP 200iseasytodesignithsincemostothectoutessume- Lead Tamperature (Soler. 10860). 400°C tions for voltage feedback amplifiers can also be used for current ' feedback amplifiers. The two independent amplifiers of the OP-260 PACKAGE TYPE @j_ (Note 2) % UNITS allow two channel amplification with matched AC performance. It is ee also ideal for high-speed instrumentation amplifiers. Other applica- § 108) MS tions for the OP-260 include ultrasound and sonar systems, video _&PinHermeticDIP@ oO 1H# amplifiers and high-speed data acquisition systems. 8-Pin Plastic DIP (P) 96 v7 °C 20-Contact LCC (RC) 88 33 CW. ABSOLUTE MAXIMUM RATINGS (Note 1) “ePmsoLg) ea ee ce #20MA Peak in'socket for TO, P-DIP, and LCC packages; ©, is specified for device soldered to seseessssnnsneeceennnncneee printed erouit board for SOL ‘ ELECTRICAL CHARACTERISTICS at V, = £15V, Vo,, = OV, R, = 2.5k0, T, = +25°C, unless otherwise noted. OP-260A/E OP-260F OP-260G PARAMETER SYMBOL —_ CONDITIONS MIN TYP) MAX MIN TYP) MAX MIN TYP) = MAX UNITS Teleoe Mos - 1 35 - 2 5 - 3 7 mW Input Bias le, Noninverting Input - 02 1 - 03 2 - 05 3 pA Current ed Inverting Input - 3 8 - 4 10 - 5 18 s Input Bias j Mode Roecion CMRRI,_ ‘en input - 004 04 - 006 02 - 04 05 pAV Ratio Input Bias inverting Input - 002 O14 - 006 02 - 005 08 Gurrent Power PSRRIa——Noninverting Input = 0.002 0.02 - 0.006 0.04 - 001 of bAV Supt Rejection PSARIS Yay wteV oem? CMR Voq=tttV a a 50 60 B rroeciey PSR Vg =28V 10 818V 6 72K 6 668 6 66 8-8 Open-Loop R, = tka Transimpedance Vg =#10V 5 7 4 5 4 5 > Ma Pango WR Fan) - - at - - an - - v Output Voltage sy R= 1k #12 4126 - #12 412.6 - #124126 - v ‘Swing ° lou =220mA #10 £115 - 211 £115 - 2110 $11.5 - Supply No Load, Curent Igy Both Amplifiers - 9 105 - 9 105 - 9 105 = mA Ay = 41, VQ =t10V, Ria tkQ,festatV, assy ~ 1000 - = 1000 - ~ 1000 - Ry = 1k0, Test at Vq = 45V 375550 7 soo 880 ~ soo 880 - . Slew Rate sR A, =#10, Ve =H10V, Vins = 1kQ, Testat Vq = 45V ohn Hermetic DIP (2) 300 - - 800 - - - - - Package ‘ 2 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER ELECTRICAL CHARACTERISTICS at V, = +15V, Voy, = OV, Ry = 2.5kQ, T, = 25°C, unless otherwise noted. Continued nn nc eS EEE EEE SORES we OP-260A/E OP-260F OP-260G PARAMETER | SYMBOL CONDITIONS MIN TYP MAX MN TYP MAX MN TYP) MAX UNITS -3d8 348 point Av Bw Avett - 90 - - 9 - - 90 - MHz Bandwidth R, = 5000 Ay = +10 - 40 - - 40 - - 40 - I Setting Time ts 100 stop, 0.1% 250 250 250 ns ON RO I Input Noninverting Capacitance cw and Inverting inputs - 45 - - 45 - - 45 - pF fo = 100kHz, Channel 2 * Separation CS wore” - 100 - - 100 - - 100 = 4B I ELECTRICAL CHARACTERISTICS at V, = +15V, V,,, = 0V, Ry = 2.5k2, -55°C < T, < +125°C, for the OP-260A, unless other- wise noted. a OP-260A PARAMETER ‘SYMBOL CONDITIONS wan Te MAX UNITS Input Offset Voltage Vos - 18 6 mv Average Input Ottset v on TCViog - 8 - rc le Noninverting Input - 03 2 Input Bias Current ia Inverting Input - 4 12 A Input Bias u Current Common Inverting Input Mode Pej CMRRI,_ Veuntit¥ - 0.05 02 WAV Rat Input Bias , Inverting Input Current Power PSRRI,_ Noni ret - 0.03 02 wv ‘Supply Rejection PSRRI,, Vee "Ov ioe av - 0.003 0.05 Ratio s Common Mode cMR Voy=tt1V 82 58 - 8 Rejection ‘ower Supply PSR Vg =49V to +18V 2 70 - eB Rejection Open-Loop Ry = tka, Transimpedance Rr Vo =#10¥ 3 48 - Ma Input Voltage VR a - - v Cup Vohiage y, Rota 215 #24 - y Swing ° lour=#20mA #05 a4 - ‘Supply Current ley cage - 9 115 mA a a BION Rav 41

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER ELECTRICAL CHARACTERISTICS at Vg = £15V, Voy = OV, Re = 2.5kQ, -40°C < T, < +85°C for the OP-260E/F/G, unless other- wise noted. OP-260E OP-260F OP-260G PARAMETER © SYMBOL CONDITIONS MIN: TYP MAX MIN TYP MAX MIN TYP MAX UNITS Input Offset Wattage Vios - 14 6 - 25 8 - 37 10 mv Average Input Offset Voltage TCVigg - 6 - - 8 - - 10 - avec Drift Input Bias lb, Noninverting Input - 03 2 - 04 3 - 08 5 A Current lp Inverting input - 4 12 - 5 15 - 7 20 La Input Bias Current Common Inverting Input Rejection MARI. ety - 005 02 - 07 04 - 015 10 Av Ratio Input Bias Current Power PSRRIg__Inverting Input - 003 02 - 005 04 - 04 10 gy Supply Rejection PSRRI,, — Noninverting Input - 0.003 0.05 = 0.005 0.1 - 001 02 Ratio ‘Common Mode Reh cMR Voy =2t1V 52 60 - 5058 - 50 5B GB Power Supply _ . . ojocton PSR Vg =#9V to +15V 62 70 60 64 60 64 3B Open-Loop RL = 1ka, . _ . Transimpedance Vo =#10V 5. 8 2 ‘ 2 ‘ Ma Input Voltage , WR wt - - tt - - int - Range v \\ Output Voltage R= tka #115 £125 — HS 4125 - ans 25 V ‘Swing ° Nour = t20mA £10.50 £171 - 210.5 411.1 - 210.5, oan - Supply No Load, _ . _ Goront ley Both amolfirs 9 15 9 15 9 11.5 mA CHANNEL SEPARATION TEST CIRCUIT BURN-IN CIRCUIT 250 ery Pee * => “ne PN | S) vem e>> an 10K a as b come sional) : & N 4 5/90, Rev. C1

}¥-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER « DICE CHARACTERISTICS — = ae ee 1, OUT A oe IS 2-INA rE.-- = a he 3.+IN A Wa._f-=i, ave ar Souls] fi iitgee ~5y 5. +INB b= he SS 6.-INB Teal a ae 7. OUT B ='5 i | a= av Weal A BL bey ai == = = D Le sestaa laesiaattse! ES , |e) For additional DICE ordering information, — = —— refer to PMI's Data Book, Section 2. DIE SIZE 0.089 x 0.086 inch, 7,654 sq. mils (2.26 x 2.18 mm, 4.93 sq. mm) WAFER TEST LIMITS at V, = +15V, V.,,= OV, Ry = 2.5kO, T, = 25°C, unless otherwise noted. eee OP-260GBC PARAMETER SYMBOL ‘CONDITIONS LIMITS: UNITS: Input Oftset Voltage Vos 5 mV MAX Impwcofiservonage Mos le, Noninverting input 2 ; Input Bias Current iad Inverting Input 10 HAMAX IO Go <= Current Common inverting input Mode Rej CMRRI, Voyettt¥ 02 pA MAX Ratio Input Bias Inverting Input ‘Current Power PSARIy_ Neninvertegh 02 Input HAW MAX Suppl Rejection PSARI,, Vg=40V10218V 0.04 Rat ‘Common Mode CMR Vow=tt Ww 50 dB MIN Rejection Power Supply PSR Vg=29Vt018V 60 <BMIN Rejection Open-Loop R= 1k, Transimpedance * Vo =#10V ‘ MaMIN _Trameimpadance OT Input Voltage vA H vMIN Output Voltage R, =1kQ +12 ‘Swing Yo lour=220mA an yun Swing tor OMA No Load, ‘Supply Current yy Both 105 mAMAX NOTE: Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing. a _ 5 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER EMp (260 unt, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER TYPICAL ELECTRICAL CHARACTERISTICS, SLEW RATE vs GAIN vs FREQUENCY SLEW RATE vs GAIN 0 ‘SUPPLY VOLTAGE » Ay=-1 sae LT TTT) . . tm bese tt TT oo | esto | to] Wren ail vecsesy TTT TH Weoa max sven LA | 5 sare row woo KTH = Lf Li PASS ct oe A) woo 3 rT es O\\\\\\\\ en, | 7] =x \\\\ | wo | ” s7W LH St aaa Rie i\\\\ os | ° | | | 7 | - LUTTE TW can soure urmervocnoewvaes rmeavecr PHASE SHIFT vs FREQUENCY SETTLING TIME vs OUTPUT STEP GAIN vs FREQUENCY Ay=-1 . Ay=-t Ay =-10 s ATP] “AYE re ao <= LT Soe, om alll i e PTS en ee rt || sla Cero SS es a Yon EE AM i a ee 0 mo NY SOON ° {eee Bia aN \\\\| eg a | aU TP yr “ NN LTT i) 7 LV ETT VT » NN : o o = cs ina SH : ‘e rmecume ane coms rat smeourcr ot SETTLING TIME PHASE vs FREQUENCY vs OUTPUT STEP GAIN vs FREQUENCY Av=-10 Ay=-10 Ayset LLIN TTT of TAL Te eebmel AT] "Tada | NM -SS0 Cn COE LES NU =| SNe SCONE Tes mannii aD cw | TTI SANG on : VAN =H SN Cg, ET ANY j= tT NN atypia Bint”: ce | — TTT TTT i [a ei “ ma AN LUT ® See a eal: ww Meee te a-T YT TT ~ Hl | I il | Ro iM eee TTI : 5 * i a c a mec Torso mat smear ee 6 5/90. Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER #7 TYPICAL ELECTRICAL CHARACTERISTICS Continued SMALL-SIGNAL ve SUPPLY VOLTAGE PHASE SHIFT vs FREQUENCY GAIN vs FREQUENCY Ay=4 Ay=+1 s Avy=+2 bed 5 . Ce Am = T Til | wo | Y8tder0 ° CAT JCS ee SCOT Ag SoS ; I ew {Il POTTY TT ee eZ ll mill ao || : CT) sii TN og DEN (nn = NI of. Nea 4\\\\ ey ee ee} SSA COPE = eA | CCCCCeeeee pee Hu all iM Cee BERS INS Ar) £2: 24 £6 £8 £10212 £14216 £18220 1 pnsqutcy ons) wo 1 emzquency pay 100, SUPPLY VOLTAGE (VOLTS) PHASE SHIFT vs FREQUENCY GAIN vs FREQUENCY PHASE SHIFT vs FREQUENCY = Ay=+5 Ay=42 “ Ay=+5 : Ae) eT CART anes, eer “ere CTT [OOS oS ; ¢ =| TSS 6 (2COTr SaA to = PN * ‘ie = Pe NT Cr tN . lees * | 100 v es (| oe > eo r my NESE - ‘i eg A, 4 IS SW PCH iy id p' 0 YT NAN § 2 [| ! 0 il CW =n oo og l NI CCE rrr sec mcrae rnin GAIN vs FREQUENCY PHASE SHIFT vs FREQUENCY GAIN ve FREQUENCY Ay=+10 Av=+10 ° v= ‘ee O-PS) EE LO os § lapdeauzen | | sh iyare To Ig SSS eer TT ° a - \\ pas ll oS) be coo ima;;.| - il SAN Mis "TIS = EN Sl i” nme ZN TTI “NN ‘4 = CAM ” amNY LLU TW Commi | ANT G ' prequency amie “ FREQUENCY (ty FREQUENCY (mt) 7 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER TYPICAL ELECTRICAL CHARACTERISTICS Continued ( PHASE SHIFT vs FREQUENCY GAIN vs FREQUENCY PHASE SHIFT vs FREQUENCY AS Qe] Aes > thease — er — SS = Rie 4 a: Si “rinl =o = OSSET i pees TN * rT WTI i eg A || ae LL 7 ATT i “te NSS ee | od ie mB Ul “NL | | ||I pe LT e7 NIIIl Bon a | os Tae oo | [TTA peg |, =I FCoE oon a 0 = AN\\ LTT TTT : Se “ ~. ‘o ' . rs NONINVERTING INPUT INVERTING INPUT INPUT VOLTAGE NOISE CURRENT NOISE DENSITY CURRENT NOISE DENSITY DENSITY vs FREQUENCY vs FREQUENCY ‘00 vs FREQUENCY 10 10 eee — ea sss oo ee oe | ES EAH, ama fl FF a 2a Seeseii eas iH g Cr ye are TT a a a i RT be EH tl ot | OE ee a gs SSS ENS He RNS 5 ooo H a a | a a 8 | 3 | | 0 | 3 a | a on So Cn So To COUT) I) TT moons rmecuencr PmcauEne OUTPUT IMPEDANCE MAXIMUM OUTPUT SWING TOTAL SUPPLY CURRENT vs FREQUENCY vs FREQUENCY « vs SUPPLY VOLTAGE x 2 - TT TU NT . | __] ee itty SC AW we | -+— = TT! ri ee att i’ - | 6 /\\ || | [CUT ame WITT : ee eet ee a E ad a | : og i. Coit sLrrszmolll | TINA TUTTI g “tial ares OTIS CTT : me * Cette (ll CIES TO Fo} ff chee TTT Cat CoE A . 1 0 00 100K ™ 101m oom o s #10 15 2 rmsouency ets rescuer 9 ‘ur vornae vers a 8 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER r TYPICAL ELECTRICAL CHARACTERISTICS Continued OUTPUT SWING vs POWER SUPPLY REJECTION COMMON-MODE REJECTION OUTPUT CURRENT @ 1kHz te vs FREQUENCY vs FREQUENCY “ « es) ee) ee eS PET) Pee Se - IN BEET TIN f y of BS Z piot | |) hUPEE TS it | tt . ee ° oo 0 «0 9% 1k 10% 100% ™ 10M 100k CTR CtOk SMSC Ser ma) [FREQUENCY (He) FREQUENCY (He) APPLICATIONS INFORMATION CURRENT VERSUS VOLTAGE FEEDBACK AMPLIFIERS between the noninverting and inverting inputs. The inverting The dual OP-260 employs a unique circuit topology that sets it “input’ is in reality a low impedance output. Current can flow into apart from conventional op amps. By using a transimpedance or out of the inverting input. A transimpedance stage follows the amplifier configuration, the OP-260 provides substantial im- input buffer that converts the buffer output current into a linearly provements in bandwidth and slew rate over voltage feedback proportional amplifier output voltage. op amps. Figure 1 compares models of these two different The current feedback amplifier loop works in the following fash- amplifier configurations. ion (Figure 1b). As the noninverting input voltage rises, the A voltage feedback op amp multiplies the differential voltage at inverting input follows and the buffer sources current through its inputs by its open-loop gain. The feedback loop forces the R,. This current, multiplied by the transimpedance stage, output to a voltage that, when divided by R, and R,, equalizes causes the amplifier’s output voltage to rise until the current the input voltages. Unlike a voltage feedback op amp, which has flowing into R, from the amplifier’s output equalizes the current high impedance inputs, the current feedback amplifier has a through R,, replacing the buffer’s output current. At steady high and a low impedance input. The current feedback state, only a very small buffer output current must flow to sustain amplifier's input stage consists of a unity-gain voltage buffer the proper output voltage. The ratio (1 + R/R,) determines the CONVENTIONAL OP AMP CURRENT FEEDBACK OP AMP ° wo O OVour i O OVour A, Ry = Ayer ge = Ayre ge VOLTAGE CONTROLLED ‘CURRENT CONTROLLED "VOLTAGE SOURCE ‘VOLTAGE SOURCE 4 : 5) FIGURE 1: The conventional op amp (a) can be modelled as a voltage-controlled voltage source. In contrast, the current feedback op amp (b), resembles a current-controlled voltage source. . 9 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER closed-loop gain of the circuit. The result is that when designing with current feedback amplifiers the familiar op amp assump- ¢ tions can still be used for circuit analysis: 1. The voltage across the inputs equals zero. ‘wo |] 2. The current into the inputs equals zero. \\/ ov. BANDWIDTH VERSUS GAIN > ~ Aunique feature of the current feedback amplifier design is that the closed-loop bandwidth remains relatively constant as a function of closed-loop gain. Voltage feedback op amps suffer from a bandwidth reduction as closed-loop gain increases, as ‘ quantified by the gain-bandwidth product (GBWP). Thisis illus- Ry trated in Figure 2 which shows the frequency response of the OP-260 for various closed-loop gains and the frequency re- = Ry = SMALL SIGNAL TRANSIMPEDANCE sponse of a voltage feedback op amp with a gain-bandwidth fey mPUT DUFFER OUTPUT RESISTANCE product of 30MHz. The bandwidth of the OP-260 is much less dependent upon closed-loop gain than the voltage feedback op FIGURE 3: Simple frequency response model of the current amp. feedback amplifier. The model shown in Figure 3 can be used to determine the fre- Cy quency response of a current feedback amplifier. With this aL | model, the frequency response dependency on the value of the e a ‘ | I feedback resistance is easily seen. Ba Sai ut | From the model of Figure 3, nodal equations may be written for [ HIS V, and V,. » SONY 1 2 PLT ITN

3 Ih SIN Vin ( Fe) Vout

. en Vy2— Ri) ( rT sae “CT _ l L il Ve=—Rt__| “Soo ™ 108 00M eT TesRTCo | FREQUENCY (He) Vin - V1 1,1)_V where |1 =VIN=Vt oy, (141) Vout =V2. FIGURE 2: Frequency response of the OP-260 when con- ren Rinv ' (é * a) Ro’ and Vout = Va. nected in various closed-loop gains with R, = 2.5kQ and R, = an _ aac. 100Q. Note that the frequency response of the OP-260 does Combining these equations yields: not follow the asymptotic roll-off characteristic of a voltage Vin ( Ro }+Vour feedback op-amp. Vour= Ri ( [a }¥ Rr 1+R2,Re |\\Ri Rel Re | 1+sRrCc FEEDBACK RESISTANCE AND BANDWIDTH Ri Rinv The closed-loop frequency response of the OP-260 shown in Figure Ifthe transimpedance of the amplifier, R,, is » R, and R,,,,, then 2 applies for a fixed feedback resistor of 2.5kQ. The frequency re- the transfer function may be simplified to: sponse ofa current feedback amplifier is primarily dependent on the value of the feedback resistor. The design of the OP-260 has been 14Fe optimized for a feedback resistance of 2.5kQ. By holding the feed- Vour, Rs back resistor value constant, the -3dB frequency pointwillalsoremain Vin tes[Rea(1 +f) Paw] Co constant within a moderate range of closed-loop gain. Ri a 10 5/90, Rev. C1

a ‘OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER The transfer function shows that the dominant closed-loop pole For higher gains, the -3dB frequency is determined by R, plus b is mainly dependent onthe value of the | feedback resistance, R,, the outont resistance of wre buffer, Frwy (sypically hear which quency. ' ? causing the bandwidth to decrease. However, the closed-loop bandwidth of a current feedback amplifier still far exceeds that Vour ., __1___ of a voltage feedback op amp for moderate values of gain. Va 1+ aRaGe Figure 4 shows the effect of the feedback resistance on the fadB= —_1_ bandwidth of the OP-260 for various closed-loop gains. 2xR2Co where R, » Riyy- GAIN “ FREQUENCY GAIN Ma FREQUENCY es iF = ee : eet al . Hee A CCTM TANT Aaa —4.\\ , HAIN ORES @ OPT eI 4, LIN aL UTTIN GT “UIE ANTM aes NUM A “LOM UN LLU TT e ’ een 100 1 onus 100 GAIN vs FREQUENCY GAIN vs FREQUENCY ° Rr = 5kQ. . Re = 10kQ r . pe TN . pe PRUE CERI Bean Pa ass el ASS ial @ 40) USS o 4, a eal TTT eN NT = -LOTTINEST wT TANI ot Tes UM LUT TN A, 1 “en 100 1 am 100 FIGURE 4: Bandwidth will vary with feedback resistance. Peaking increases as the feedback resistance is decreased. R, = 2.5kQ is the recommended value. All graphs are normalized to OaB. eee " 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER ST _o,__ TRL HIGH SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER Lo | - on Ci 7 3 t. oN ©Vour ° Q L 3 Oye 7 7 ov FIGURE 5: Simplified schematic of the OP-260 ‘showing the three stages of the amplifier. SLEW RATE AND GAIN l The simplified schematic in Figure 5 shows the three stages of SLEW RATE vs GAIN the OP-260. The input stage consists of a unity-gain emitter- wo — follower amplifier. Q, and Q, form a class AB output stage at the wee Ht aa inverting input which can source or sink current. The current we Pe litey rie ol flowing through the inverting input is sensed by the top current soo KT mirror, formed by Q,, Q,, and Q,,, or the bottom current mirror, PW oo formed by Q,, Q, ,, and Q,,. When the buffer sources current to E 0 ESN a a load, current flows out of the inverting input, increasing Q,'s = [CP SSR TT collector current and causing more current to flow through Q, i head a | and Q,,. This increases the base drive to the output transistor eo | — TTT So} Q,,. Simultaneously, the increased current in Q, drives Q,, PT Sol which reduces base drive to the complementary output transis~ ao | — fT Ts tor Q,, This push-pull action produces a very fast output slew K-AH tate. Forasmall voltage step, the OP-260's slew rate is. depend- y 0 00 ‘ent on the available current from the two current sources (I, and GAM (ABSOLUTE) |,) that drive Q, and Q, To increase the slew rate, transistors Q, and Q, have been FIGURE 6: Slew rate of the OP-260 is highest in gains added to boost the base drive to Q, and Q,. In closed-loop gains below £10. below 10, a large input step will turn on Q, or Q, increasing the slew rate dramatically as illustrated in Figure 6. For the noninverting amplifier, the equivalent input voltage AMPLIFIER NOISE PERFORMANCE noise, referred to the input, is: ‘Simplified noise models of the OP-260 in the Noninverting and inverting amplifier configurations are shown in Figure 7. All re- En=*/(Reinn?? + en? + (Re ini/AvoL sistors are assumed to be noiseless.

12 RION Raw A4

FMD Cora DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER . Ps Q ° =O Oy, ) ) ™ 25K, Q, Qa OQ» Oe gn, Om Ow Gum FIGURE 7: Simplified noise models for the OP-260 in noninverting (a) and inverting (b) gain. where: 0.1pF bypass capacitor are recommended for each supply, as —, = total input referred noise shown in Figure 8, and will provide adequate high-frequency e, = amplifier voltage noise bypassing in most applications. The bypass capacitors should in = noninverting input current noise be placed at the supply pins of the OP-260. As with all high fre- igi = inverting input current noise quency amplifiers, circuit layout is a critical factor in obtaining Ay = source resistance optimum performance from the OP-260. Proper high frequency Avci= closed loop gain=1+R/R, layout reduces unwanted signal coupling in the circuit. When breadboarding a high frequency circuit, use direct point-to-point For the inverting amplifier, the equivalent input voltage noise, wiring, keeping all lead lengths as short as possible. Donotuse referred to the input, is: wire-wrap boards or “plug-in” prototyping boards. 5 2 (1+lAvewl) , (Re ini) During PC board layout, keep all lead lengths and traces as N= / On’ ( | Aveul | *Tavetl short as possible to minimize inductance. The feedback and ing Re « R. closed | ain =-R,/R gain-setting resistors should be as close as possible to the in- é assuming Rg “ R,. Ayo, = closed loop gain = -A,/R,. verting input to reduce stray capacitance at that point. To fur- Typical values @ 1kHz for the noise parameters of the OP-260 ther reduce stray capacitance, remove the ground plane from are: the area around the inputs of the OP-260. Elsewhere, the use of e, = 5.0nV/VHz asolid unbroken ground plane will insure a good high-frequency ign = 3.0pA/V Hz. ground. i, = 20.0pA/VHz SHORT CIRCUIT PERFORMANCE ve To avoid sacrificing bandwidth and slew rate performance the ? OP-260's output is not short circuit protected. Do not short the Your ote amplifier's output to ground or to the supplies. Also, the buffer e r rt output current should not exceed a value of +20mA peak or ° ~ ~ +7mA continuous. ° POWER SUPPLY BYPASSING AND LAYOUT ° 4 CONSIDERATIONS Proper power supply bypassing is critical in all high-frequency b — circuit applications. For stable operation of the OP-260, the io zor or power supplies must maintain a low impedance-to-ground over = = an extremely wide bandwidth. This is most critical when driving alow resistance or large capacitance, since the current required FIGURE 8: Proper power supplying bypassing is required to to drive the load comes from the power supplies. A 10uF and obtain optimum performance with the OP-260. ee 13 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER —— Neem, HIGHSPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER. APPLICATIONS tor, R, ‘eee with his aay capacitance creating a zeroin ne NOP closed-loop response. For large noninverting gains, R, is small, The oeeen ING AMPLIFIER ' / Creating a very high frequency open-loop pole which has limited ef- Mi can be used as a voltage-follower or noninverting " . an . paar fect on the closed-loop response. As the noninverting gain is de- amplifier as shown in Figure 9. A current feedback amplifier in this . configuration yields the same transfer function as a volta ge feed- creased, R, becomes larger and the stray zero becomes lower in back op amp: frequency, havingamuch greater effecton the closed-loop’ response. Pp amp: To reduce peaking at low noninverting gains, place a series resis- Your T= 4 +Be tor, R,, in series with the noninverting input as shown in Figure 9. 'N M1 an This resistor combines with the stray capacitance atthe noninverting Remember to use a 2.5k2 feedback resistor in vottage-follower inputto form a low-pass filter that will reduce the peaking. The value application. of R, should be determined experimentally in the actual PCB lay- In noninverting applications, stray capacitance at the inverting in- out. Less peaking will occur in inverting gain configurations since put of a current feedback amplifier will cause peaking which will in- the inverting input is a virtual ground which forces a constant volt- crease as the closed-loop gain decreases. The gain setting resis- age across the stray capacitance. Acommon practice to stabilize voltage feedback op ampsis to use capacitor across the feedback resistance. This creates a zero in “ the voltage feedback amplifier response to offset the loss of phase ste margin due to a parasitic pole. In current feedback amplifiers, this + Fy technique will cause the amplifier to become unstable because the = closed-loop bandwidth will increase beyond the stable operating bey al frequency. For the same reason, current feedback amplifiers will Re ae Not be stable in integrator applications. Yo = * 1, oy INVERTING AMPLIFIER ‘eereT ‘or The OP-260 is also capable of operation as an inverting amplifier aad (see Figure 10). The transfer function of this circuit is identical to that using a voltage feedback op amp: exo Vout . Re. " ie Vin Ri { * Fl An optional offset voltage trim is shown in Figure 11. Oe FR AUTOMATIC GAIN CONTROL AMPLIFIER = wt BE > One of the shortcomings of using voltage feedback op amps in an “ ™ iV Automatic-Gain-Control ampliifieris thatits bandwidth drops off rapidly as gain increases, limiting the useful bandwidth. However, for cur- FIGURE 9: The OP-260asa voltage follower or noninverting rentfeedback amplifiers, bandwidth is relatively independentof gain, amplifier. 18 5 10uF 10HF Ae AF a R

2 GN 7 vot |

© Vout oo in © Vout er, 1080 d > fe -18v0 500. aioe = = our Onur Y 8, 4 2 , ~1v \\ FIGURE 10: The OP-260 as an inverting amplifier. FIGURE 11: Optional offset voltage trim circuit for the OP-260. “ 5/90. Rev. C1

PMI) OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER & no 1 200K, our oe.2 *

1089 Ni hs D2

VCR2N, SZ nos FIGURE 12: The OP-260 eliminates the problem of variable bandwidth in AGC amplifiers using voltage feedback op amps. eliminating this problem. Figure 12 shows a simple AGC amplifier Q, on harder, reducing the channel resistance and increasing the design using the OP-260. Amplifier A, , is used as the gain stage. gain. Figure 13 shows the pulse response of the AGC amplifier. Its output is rectified by the second amplifier A,,,. If the output volt- The AGC loop maintains constant peak outputamplitude forasquare age swings more negative, diode D, forward biases and D, reverse wave input signal range of +20mV, to 46.0V,.. ° i ‘ech... pet biases, closing the loop on amplifier A, APositive voltage appears | ow PHASE ERROR AMPLIFIER onthe anode of D, ; but, ifthe output voltage swings positive, D, reverse ‘ th . m Saas biases and D, forward biases, keeping the loop closedon ,,.. This The simple amplifier depicted in Figure 14 utilizes the monolithic prevents the amplifier from saturating to the negative rail. Phe re- dual OP-260 anda few resistors to substantially reduce phase error 6 sult is an accurate positive rectification of the output signal. over a wide frequency range compared to conventional amplifier The output of the rectifier is then compared with a reference current setup by the 604kQ resistor which is biased to—15V. The output of the error amplifier A, will drive the FET (Q,) to the proper voltage “N necessary to achieve a zero voltage at the inverting input of A,. It sour there is insufficient signal, the error amplifier will detect an imbal- + FL ance. This causes the error amp to drive more positive, turning FET + rd — eR de ® Wa \\ 20, — Vu O 4 q = tn, © Vor o 'T . \\ Oar ——— AL o @ 220s] av & FIGURE 13: Pulse response of the AGC amplifier at (a) low level FIGURE 14: Active feedback allows cancellation of the dominant rd 15 5/90, Rev. C1

OP-260 DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER A DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER designs. This technique relies on the matched frequency charac- voltage of the instrumentation amplifier is determined by the Vios teristics of the two current feedback amplifiers in the OP-260. Re- matching of the OP-260, which is typically under 0.5mV. ferring to the circuit, notice that each amplifier has the same feed- Figure 17 shows the relationship between gain and bandwidth for back resistor network, corresponding toa gain of 100. Sincethese —_the instrumentation amplifier. Reducing resistors R, to R, to2.5kQ two amplifiers are set at equal gain and are matched due to the —_ increases the bandwidth but makes circuit performance more de- monolithic construction of the OP-260, they will have an identical pendent on board layout. frequency response. A pole in the feedback loop of an amplifier becomes a zero in the closed loop response. With one amplifier in the feedback loop of the other, the pole and zero are at the same sv frequency, thus cancelling and reducing low phase error. Figure 15 *9 shows that the low phase error amplifier at a gain of 100 exhibits 1° pn of phase error up to a frequency of 1MHz. For a single voltage = FL feedback op amp to match this performance, it would require a gain- = bandwidth product exceeding 10GHz! vw 3 7 , SEAR Bocca _ MABPTLAfeasirteae - Lo ovo ‘Coco Het, | 9 2

40 P—}- aoa x0

od a aS =, ed a ", FL is |} faa ST sa one = s}—_| [| TTT NY + wm, 1m, = ao} —| | TU Yor BW eg a a AN ‘on prequency oe) FIGURE 16: High Speed Instrumentation Amplifier ¢ FIGURE 15: Phase response of the ultra-low phase error ampli- fier compared to that of a single current feedback amplifier. Note 10 -— — that there is only one degree of phase error over a 1MHz band- ——aoe eo width at a gain of 100. | HIGH-SPEED INSTRUMENTATION AMPLIFIER tN The circuit of Figure 16 is a high-speed instrumentation amplifier i LNT constructed with a single OP-260. Gain of the amplifier is set by resistor R, according to the following formula: a ll Vout 10k2 | 5. Vin Re The advantages of the two op amp instrumentation amplifier is that the errors in the individual amplifiers tend to cancel one another. yy 0 100 Common-mode rejection is limited by the matching of resistors R, Am (a8) to R,. For the best CMR performance, these resistors should be matched to 0.01% ora CMR trim can be performed on R,. ACMRR FIGURE 17: Bandwidth versus gain for the high speed instrumen- of 90dB (measured at 60Hz) is achievable at all gains. Input offset tation amplifier. eeeeeSSSSSSSSeSSSSSSSSFSSMMMeee 6 5/90. Rev. C1

Summary

DUAL, HIGH-SPEED, CURRENT FEEDBACK, OPERATIONAL AMPLIFIER

Description