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Provision for Standby mode operation Partial Resonance Power Supply IC Module MR2900 Series Application Note Input voltage autosensing Tentative 2002/03/01
Shindengen Electric MFG.CO.,LTD - 2 - 1. The circuit diagrams and parts tables provided for reference purposes in this document are for th e use of persons with basic circuit design knowledge to aid in understanding the product. As such they do not constitute a guarantee of output, temperature, or other characteristics, or characteristics or safety as determined by the relevant authorities. 2. The products noted in this document are semiconductor components for use in general electronic equipment and for general industrial use. Consideration has been given to ensure safety and reliability as appropriate for the importance of the systems used by the customer. Please contact Shindengen's sales section if any points are unclear. 3. Fail-safe design and safety requirements must be considered in applications in which particularl y high levels of reliability and safety are required (eg nuclear power control, aerospace, traffic equipment, medical equipment used in life-support, combustion control equipment, various type s of safety equipment). Please contact our sales department if anything is unclear. 4. Shindengen takes no responsibility for losses or damage incurred, or infringements of patents or other rights, as a result of the use of the circuit diagrams and parts tables provided for reference purposes in this document. 5. The circuit diagrams and parts tables provided for reference purposes in this document do not guarantee or authorize execution of intellectual property rights, or any other rights, of Shindengen or third parties. 6. Systems using Shindengen products noted in this document and which are strategic materials a s defined in the Foreign Exchange and Foreign Trade Control Law or the Export and Trade Control Law require export permission under the relevant legislation prior to export. Inquiries: Functional Devices Division, Device Sales Department, Device Sales Section Ph 03-5951-8131 Fax 03-5951-8089 Thank you July 1st, 1995 Cautions When Using This Document MR2900 Application Note
Shindengen Electric MFG.CO.,LTD - 3 - 1. Outline
1.1 Introduction … 4
1.2 Characteristics … 4
1.3 Applications … 4
1.4 Absolute Maximum Ratings
and Reference Output Capacities … 4
1.5 Equivalent Circuit and Dimensions … 4
- Block Diagram
2.1 Block Diagram … 5
2.2 Pin Function Description … 5
- Operation Description
3.1 Start-up Circuit … 6
3.2 On-trigger Circuit … 7
3.3 Partial Resonance … 7
3.4 Standby Mode Control … 8
3.5 Output Voltage Control … 9
3.6 Soft Drive Circuit … 9
3.7 Circuit for Load Shorts … 10
3.8 Collector Pin (pin 7) … 10
3.9 Thermal Shut-down Circuit (TSD) … 10
3.10 Over-voltage Protection Circuit (OVP) … 10
3.11 Malfunction Prevention Circuit
(patent applied for) … 11
3.12 Over-current Protection Circuit … 11
- Standard Circuit … 12 5. Design Procedures
5.1 Design Flow Chart … 13
5.2 Main Transformer Design Procedure … 13
5.3 Main Transformer Design Examples … 15
5.4 Selection of Constants for
Peripheral Components … 18 6. Cooling Design
6.1 Junction Temperature and Power Losses … 19
6.2 Junction Temperature
and Thermal Resistance … 19
6.3 Cautions for Cooling Design … 19
The values presented in this document are based on tentative specifications as of June 29th, 2001, and may change in future
Contents
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1.1 Introduction
The MR2900 Series IC modules are designed for both 200V and autosensing input with a burst-mode switching function at micro- loads. These modules are of the partial resonance type, and are comprised of a switching device optimized for both 200V and autosensing power supply input, and a control IC. They are designed to provide the following power supply characteristics.
1.2 Characteristics
- An ultra high-speed IGBT with 900V resistance ensures high efficiency and low noise at partial resonance. 2. An ultra high-speed IGBT with 900V resistance simplifies design for autosensing power supply input. 3. Very low power consumption at micro-loads (in burst mode). 4. Onboard start-up circuit eliminates the need for start-up resistors. 5. Soft drive circuit achieves low noise levels. 6. Excess current protection function (ton limit, primary current limit). 7. Excess voltage protection and thermal shut-down function. 8. Power supply circuits may be constructed with a minimum of external components. 9. The use of a full mold package provides benefits in insulation design.
1.3 Applications
TVs, displays, printers, VTR, DVD, STB, air-conditioners, refrigerators, and other electrical appliances, and office equipment.
1.4 Absolute Maximum Ratings and Reference Output Capacities
Absolute maximum ratings Maximum output capacity P o［W］ Peak input voltage Peak input current Input voltage range Model Vin［V］ I in［A］ 90V to 276VAC 180V to 276VAC MR2920 7 100 150 MR2940 900 10 150 225 Maximum output capacity and input voltage range differ with design conditions.
1.5 Equivalent Circuit and Dimensions
5 Vin
4.5±0.5 4.2±0.5 4.2 5.0±0.2 2.4±0.2 3.0±0.2 16.7±0.3 12.0 φ3.2 +0.2 -0.1 20.0±0.2 11.8±0.5 0.7 +0.3 -0.12.54±0.2 8.0 7.6±0.5 2.7±0.2 0.7±0.2 1. Outline MR2900 Application Note
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2.1 Block Diagram
2.2 Pin Function Description
Pin number Abbreviation Description 1 Z/C Trigger input pin Zero detection voltage: 0.35V Standby: Up to 4.5V in standby mode.
2 F/B Feedback signal input pin
ton(min) to ton(max): 1.5V to 4.5V/0μs to 25μs Standby: Oscillation stopped at up to 0.8V. Standby: Oscillation started at 1.8V or higher.
3 GND GND pin
4 V cc IC power supply pin
Oscillation start voltage: Vcc≧14V Oscillation stop voltage: Vcc≦8.5V Excess voltage latching voltage:Vcc=20V
5 V in Start pin
Current supplied Vin→Vcc at start-up Start-up circuit OFF:Vcc≧14V Start-up circuit ON: Vcc≦7.6V
6 Emitter
/OCL Main switching device emitter and current detection pin Excess current detection threshold:0.6V Excess current detection threshold at standby: 50mV
7 Collecter Main switching device collector pin
- Block Diagram MR2900 Application Note Collector Emitter/OCL Vcc Z/C F/B Thermal Shutdown circuit Soft drive circuit Start-up circuit R S Q VOVP OVP comparator VUL Unlatch comparator VCC(start) /VCC(stop) UVLO comp VCC(startup off) /VCC(startup on) Start-up circuit UVLO comparator Vin On-dead timer ON range timer S R Q Standby circuit Restart timer Vref IF/B VTH(OCL) Excess current detection comparatorVTH(burst limit) Burst current limit comparator GND Zero current detection circuit
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3.1 Start-up Circuit
In conventional start-up circuits employing a start-up resistor, current continues to flow following power supply start-up, thus wasting power and reducing efficiency, particularly during standby. See Fig.3.1 Comparison of Start-up Circuits - Conventional Start-up Circuit. In the MR2000 Series start-up circuit the start-up current is supplied from the input voltage at power supply start-up, and is shut-off when the power supply is in operation. The start-up circuit supplies a current of 12mA (typical) from the IC internal constant current source until the voltage at the V cc pin reaches 14V (typical). This current is consumed internally in the IC as well as being used as the charging current for the condenser connected externally between the V cc pin and GND. This design allows a stable start-up only minimally dependent upon input voltage. When the voltage at the Vcc pin reaches 14V (typical) the start-up circuit is disconnected, the start-up current no longer flows and oscillation begins simultaneously. The current consumed in the IC is then supplied from the control coil. See Fig.3.1 Comparison of Start-up Circuits - MR2000 Start-up Circuit. In the case of an instantaneous power failure or a load short, oscillation is stopped when the voltage at the V cc pin reaches 8.5V, and when this voltage drops to 7.6V the start-up circuit operates again and the voltage at the Vcc pin then begins rising. See Fig.3.2. Incorporation of the functions described above improve efficiency, particularly during standby, and reduces the number of start-up resistors required, thus reducing the overall number of components. Fig.3.1 Comparison of Start-up Circuits Fig.3.2 Start-up Circuit Operation Sequence 3. Operation Description MR2900 Application Note Load shortInstantaneous power failure 【VOUT】 VCC(startup on) =7.6V 【VCE】 VCC(stop) =8.5V VCC(startup off) =VCC(Start) =14.0V 【VCC】 【Vin】 Vcc pin Vin pin5 Vcc(startup off) /Vcc(startup on) 14.5V/7.2V 【MR2000 Start-up Circuit】 Control coil Start-up current switched off following start-up, thus eliminating the need for start-up resistors. 【Conventional Start-up Circuit】 IC Start-up current Start-up current flows even during steady-state operation, resulting in losses.
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3.2 On-trigger Circuit
The MR2000 Series employs current-critical operation to detect energy bursts at the secondary side of the main transformer and switch on the main switching device. Energy discharge timing is detected at the negative edge of the control coil voltage waveform (0.2V in the diagram at right), and the main switching device switched on for current-critical operation. The on-trigger detection voltage (0.2V) incorporates a 50mV hystersis to increase noise resistance.
3.3 Partial Resonance
In current-critical switching power supplies (RCC), damping begins at the resonance frequency (determined by the primary inductance L P of the main transformer and the resonating condenser C) when the secondary current in the circuit formed by connecting the resonating condenser between the collector and GND of the main switching device reaches 0A. The discharge current of the resonating condenser flows through the primary coil and returns energy to the input. Adjustment of the CR time constant applied to the Z/C pin (see diagram at right) allows the main switching device to be turned on at the trough of the damping voltage waveform, thus permitting a reduction in turn-on losses. In a circuit using partial resonance, the energy stored in the resonating condenser during the OFF period of the main switching device is returned to the input, thus permitting a reduction in turn-on losses. This allows the connection of a large-capacity condenser between the collector and GND of the main switching device, and thus permits a reduction in noise. The use of partial resonance is effective in permitting a simple circuit configuration with improved efficiency and noise reduction. Fig.3.3 On-trigger Operation Sequence Fig.3.4 Partial Resonance 【VZ/C】 0.2V 【IC】 【Secondary rectification diode current】 【VCE】 【Control coil voltage】 【IC】 【Secondary rectification diode current】 【VCE】 Turn-on delay Damping begins at the resonance frequency determined by LP and C. Z/C pin Collector pin GND pin C RResonating condenser On timing delayed with CR time constant. Emitter/OCL pin
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3.4 Standby Mode Control (patent applied for)
The MR2000 Series is able to switch between two methods of output voltage control - normal operation and the standby mode, in a single power supply. The standby mode supported by this IC employs the burst method for intermittent operation under light loads to reduce oscillation frequency and switching losses, and is effective in reducing the standby input voltage under micro-loads. A unique characteristic of this IC is the use of the burst mode for intermittent operation without stopping IC control, and thus minimizing output ripple. The Z/C pin is clamped to a voltage of 4.5V (typical) or less by an external signal to allow selection of standby mode control. The standby mode is cleared (ie restored to the normal mode) by clearing the clamp voltage on the Z/C pin, and applying a voltage of 4.5V (typical) or higher. In normal operation the ON range of the main switching device is controlled in a linear manner in relation to voltage variation at the F/B pin, while in standby mode operation the Emitter/OCL pin current detection threshold value is switched from 0.6V for the normal mode to 0.05V for the standby mode. The collector current is fixed at a peak value by the current detection threshold value, and the burst mode is selected. Burst mode control is such that oscillation occurs when the voltage at the F/B pin is 1.8V (typical) or higher, and is stopped when this voltage is 0.8V (typical) or lower. As output voltage control in the standby mode fixes the drain current peak value for each oscillation cycle, the duty ratio of the oscillating and non-oscillating intervals is varied to ensure a constant voltage. Fig.3.5 Standby Mode Control Fig.3.6 Standby Mode Control Sequence Fig.3.7 Standby Signal Receive Sequence Z/C pin Drain pin Emitter/OCL pin Output voltage error detection feedback signal 4.5V (TYP) Standby signal (external signal) F/B pin Switched from 0.6V to 0.05V VF/B(burst start) =1.8V 【VF/B】 【IC】 【VOUT ripple】 VF/B(burst stop) =0.8V 【VZ/C】 【IC】 【IOUT】 Standby modeNormal operation Normal operation 0.2V 4.5V(TYP) Standby mode start Standby mode clear
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3.5 Output Voltage Control (normal operation)
The MR2000 Series controls output voltage with the ON range proportional to the voltage at the F/B pin. When the voltage at the F/B pin is 1.5V the ON range is 0µs, and is controlled in a linear manner so that when the voltage is 4.5V the ON range is 25 µs. A current of 200µA=IF/B (typical) flows at the F/B pin, and the impedance of the photocoupler transistor connected externally between the F/B pin and GND is varied with the control signal from the secondary output detection circuit, thus controlling the ON range of the main switching device to produce a constant voltage. The maximum ON range is adjusted by setting the maximum value for the voltage at the F/B pin using a resistor connected externally between the F/B pin and GND.
3.6 Soft Drive Circuit (patent applied for)
The MR2000 Series supplies the main switching device gate drive voltage from two separate drive circuits. A voltage exceeding the threshold value for the main switching device is supplied from the first drive circuit at the leading edge of the drive voltage waveform to switch on the main switching device with the optimum timing. The drive voltage is then supplied gradually by the second drive circuit (see Fig.3.9). Supply of drive voltage in this manner reduces drive losses, as well as reducing noise due to gate charge current and discharge current when the resonating condenser is switched on. Fig.3.8 Output Voltage Control Fig.3.9 Comparison of Drive Circuits F/B pin 2 Output voltage controlled by varying impedance of photocoupler. 200μA 5Vref Droop resistor Output voltage error detection feedback signal 1.5 4.5 ON range ton［μs］ Feedback voltage VF/B［V］ 【VGE】 【IG】 【IC】 Gate voltage supply matched to collector current. Reactive charge reduced under light load. Gate charge spikes reduced. Damping of resonating condenser discharge current. 【MR2000 drive circuit】 Large resonating condenser discharge current. 【Conventional drive circuit】 Gate charge remains unchanged even when collector current is small.
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3.7 Circuit for Load Shorts
The MR2000 Series is designed so that when droop occurs under excessive load, output voltage drops, and control coil voltage drops in proportion. When the control coil voltage falls below 4.5V (typical) the standby mode is selected and the Emitter/OCL pin threshold voltage changes from 0.6V to 0.05V, thus limiting the collector current to approximately 1/10th of its previous value. This design permits a reduction in the stress on the MR2000 Series IC in the case of a load short, and control of the short-circuit current in the secondary diode and load circuit.
3.8 Collector Pin (pin 7)
The collector pin on the main switching device. The transformer is designed, and the resonating condenser adjusted, to ensure that V CE(max) is less than 900V. Depending upon input conditions, the collector pin may be subjected to reverse bias for a period during partial resonance. This IC employs an ultra high-speed IGBT in the main switching device. This device differs from MOSFET devices in that it has no body diode structure, thus requiring connection of an external high-speed diode between the Collector and Emitter/OCL pins.
3.9 Thermal Shut-down Circuit (TSD)
The MR2000 Series incorporates a thermal shut-down circuit. The onboard IC is latched at 150°C (typical) and oscillation is then stopped. Unlatch is achieved by momentarily dropping the voltage at the V cc pin to VUL (unlatch voltage) or lower.
3.10 Over-voltage Protection Circuit (OVP)
The MR2000 Series incorporates an over-voltage protection circuit (OVP). Latching occurs when the control coil voltage exceeds 20V (typical), and secondary output over-voltage protection then operates indirectly. Unlatch is achieved in the same manner as for the overheat protection circuit. Fig.3.10 Circuit for Load Shorts 【VZ/C】 【IC】 【VCC】 Load short 【VOUT】 4.5V(TYP) ICP limited when VZ/C falls below 4.5V (typical).
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3.11 Malfunction Prevention Circuit (patent applied for)
The use of current-critical operation in the MR2000 Series ensures that the main transformer does not become saturated provided the droop setting is optimized. On the other hand, at start-up, and in the case of a load short, the output voltage is very much less than the set voltage. As the control coil voltage is proportional to the output voltage it also reaches an extremely small value, and the on-trigger timing may be incorrectly detected due to the ringing voltage while the device is OFF and switched on before the current-critical point. To counter this problem, the MR2000 Series incorporates a circuit to prevent on-trigger malfunction at start-up, and in the case of a load short. This function disables the on-trigger for a period of 2.7μs (typical) after the main switching device in the IC is switched OFF (on-dead time). This prevents incorrect detection due to the ringing voltage while the device is OFF. This design permits detection of the point at which the transformer secondary current is 0A at start-up, and in the case of a load short. The main switching device is then switched on at this point, allowing abnormal oscillation to be controlled.
3.12 Over-current Protection Circuit
A current detection resistor is connected between the Emitter/OCL pin and GND to detect current between the emitter of the main switching device and the emitter current detection pin. During stable operation the main switching device current is limited by pulse-by-pulse operation with the 0.6V threshold value. The leading edge clamp function prevents malfunctioning and thus prevents incorrect detection at turn-on. During standby, the 50mV threshold value is selected and the oscillation noise from the transformer due to burst oscillation is reduced. Fig.3.11 Comparison of Drive Circuits Fig.3.12 Current Detection Resistor 【IC】 【Secondary rectification diode】 【VCE】 【VOUT】 【VZ/C】 Enlarged view 【IC】 【VZ/C】 On-trigger disabled during this period. 2.5μs0.2V 【Secondary rectification diode】
7 Collector pin
Shindengen Electric MFG.CO.,LTD - 12 - T101 C104 C103 D101 C105 L101 C101 F101 C106 R101 L N D103R103 PC101 2 1 R105 R106 D105 D106 C111 PC102 C109 C107 R104 IC101 C102 D104 C201-1 L201 C202 C201-2 R205 C203 D201 VO GND R202 R204 R203 R201 R206 C204 IC201 R207 PC102 TR201 R208 SW201 R209 R210 C108 R102 D102 PC101 4. Standard Circuit Standard circuit/Parts listMR2900 Application Note
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5.1 Design Flow Chart
5.2 Main Transformer Design Procedure
This design procedure provides an example of an electrical design procedure. Ensure that design of insulation materials, insulation configuration, and structure are in accordance with the necessary safety standards as determined by the relevant authorities.
5.2.1 Standard Design Conditions
Abbreviation Unit Reference value Minimum input voltage V AC(min) V ― Rated output voltage Vo V ― Rated output current Io A ― Maximum output current Io(max) A ― Efficiency η 0.80～0.85 Minimum oscillation frequency f(min) kHz 25k～50kHz Duty ratio D 0.50～0.70 Control coil voltage VNC V 15～17V Effective cross-sectional area of transformer core Ae mm Magnetic flux density variation ΔB mT 250～320mT Coil current density α A/mm 4～6A/mm Note that the above values are for reference only, and should be adjusted to suit load conditions. 5. Design Procedures MR2900 Application Note Refer to:
5.3 Main Transformer Design Examples
Selection of primary circuit components Main transformer design Specifications determined Cooling design Trial manufacture Refer to:
5.4 Selection of Constants for Peripheral Components
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5.2.2 Standard Design Calculations
1 Minimum DC input voltage AC(min)DC(min) V1.2V ×= [V]
2 Maximum DC input voltage AC(max)DC(max) V2V ×= [V]
3 Oscillation cycle
(min) (max) f 1T = [s]
4 Maximum ON period
(min) on(max) f Dt = [s]
5 Maximum OFF period q
ON(max)DC(min)S1 off(max) t)V(VN tVNt ++× ××= [s]
6 Resonance period
CL2πt qP q ×= [s]
7 Maximum load power O(max)OO(max) IVP ×= [W]
8 Maximum output power
(reference value) O(max)L P1.3P ×= [W]
9 Peak collector current DVη
DC(min) L CP ×= [A]
10 Primary coil inductance
on(max)DC(min) P I tVL ×= [H]
11 Number of turns in primary coil
e 9on(max)DC(min) P AΔB 10tVN × ××= [Turn]
12 Core gap
P 2Pe10 g L NA104πl ×××= [mm] The gap Ig is the center gap value. Review transformer core size and oscillation frequency and redesign if Ig is 1mm or greater.
13 Number of turns
on(max)DC(min) qon(max) (min) PF1O1 tV )t-t-f 1(N)V(V N × ××+ = [Turn]
14 Number of turns
in non-control output coil F1O1 F2O2 S1S2 VV VVNN + +×= [Turn]
15 Number of turns in control coil
+×= [Turn] Consider the secondary diode forward voltage for each output when determining the number of turns in an output coil. VFNC is the control coil voltage rectification diode forward voltage. The reference value for determining the control coil voltage VNC(min) is 15V to 17V. If the VNC(min) value is too small, start-up characteristics may deteriorate and start-up may become difficult. If the VNC(min) value is too large, the over-voltage latch stop voltage VOP is able to be reached easily. Check the VNC(min) voltage in an actual circuit during the design process to determine its optimum value.
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16 Primary coil size
(min)on(max)DC(min) O NP ftVη3α PD2A ××= [mm
17 Secondary coil size
(min)qoff(max) O(min)q NS f)t(t3α I)f(tD12A ×−×× ANC=0.2mm dia. is recommended for the NC coil for ease of calculation.
5.3.1 Initial Setup
Input voltage AC90 ～276V V O1：DC135V, 0.45A Efficiency 85% V O2：DC35V,0.40A Rated output VO3：DC16V,0.40A Total output 81.2W Oscillation frequency at droop 29.6kHz Droop output 110.36W (rated output x 1.36) Duty ratio T ON/T=0.655
5.3.2 Primary Inductance (LP) Calculations
on(max) is 29μs or less. Droop output (rated total output x 1.36) calculated as P L
5.3.3 Calculation of Number of Turns in Primary Coil (N
P), and Gap (Ig) The maximum rating for ΔB for PC40 at 100°C is 390mT. ΔB has been derated to 310mT in this example. The number of turns has been rounded to the nearest integer, however this value may be adjusted as necessary. Primary inductance (L P) calculated using equations 1, 4, 9, and 10 in 5.2.2. 108901.2V1.2V AC(min)DC(min) =×=×= ［V］ 22.131029.6 0.655 f Dt 3(min) on(max) =×== ［μs］ 3.670.6551080.85 110.362 DVη P2I DC(min) L CP =×× ×=×× ×= ［A］ 651.243.67 1022.13108 I tVL CP on(max)DC(min) Primary inductance LP =0.65mH. The number of turns in the primary coil is calculated using equation 11 in 5.2.2. Specifications require the use of PC40 EER39L steel in the transformer core. Substitute A e=130mm and ΔB=310mT in equation 11. 5959.3130310 101022.13108 AΔB 10tVN 9-6 e 9on(max)DC(min) P ≅=× ×××=× ××= ［Turn］ The gap (Ig) is calculated using equation 12 in 5.2.2. 0.87100.65 59130103.144 L NA104πl 3- 210 P 2Pe-10 g =× ［mm］ The number of turns in the primary coil is NP=59, and the gap Ig=0.87mm. The gap (Ig) calculated above is a reference value. During trial manufacture, adjust the gap (Ig) in relation to the value found in the calculations, and ensure that it is appropriate to the primary inductance value. Substitute
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5.3.4 Calculation of Number of Turns in Secondary Coil (NS1)
ç Calculation assumes tq =2.5μs.
5.3.5 Verification of Resonance Time (t
differs, change t q and recalculate.
5.3.6 Calculation of Number of Turns in Secondary Coils (N
S2, NS3) ç ç
5.3.7 Calculation of Number of Turns in Control Coil (N
ç For ease of handling, a 0.2mm dia. wire is recommended for the control coil. ç ç ç ç ç ç ç ç ç ç ç The number of turns in the secondary coil is calculated using equation 13 in 5.2.2. on(max)DC(min) qon(max) (min) PF1O1 tV )t-t-f 1(N)V(V N × ××+ 3130.731022.13108 )102.51022.13 1029.6 1(591)(135 ≅=×× ×−×− ××+ = − ［Turn］ The number of turns in the secondary coil is therefore NS1=31. The calculation above assumes a resonance period (tq) of 2.5μs. This calculation verifies the effectiveness of this value in terms of LP and the resonance condenser Cq (C108) as previously calculated. 2.532 101000100.652π CL2πt 123gP q =×××=×= ［μs］ Conditions are therefore satisfied. Note that the calculation assumes a resonance condenser Cq of 1000pF. The numbers of turns NS2 and NS3 in the secondary coils are calculated using equation 14 in 5.2.2. 88.201135 13531VV VVNN F1O1 F2O2 S1S2 ≅=+ +×=+ +×= ［Turn］ 43.781135 0.61631VV VVNN F1O1 F3O3 S1S3 ≅=+ +×=+ +×= ［Turn］ The numbers of turns in the secondary coils are NS2=8 and NS3=4. A value of between 15V and 17V is optimum for Vcc. This design assumes Vcc=16V, and the number of turns in the control coil is calculated using equation 15 in 5.2.2. 43.881135 11631VV VVNN F1O1 FNCNC S1C ≅=+ +×=+ +×= ［Turn］ç
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5.3.8 Calculation of Wire Size for Primary Coil (NP)
ç ç Adjust current density in accordance with conditions of use and structure of the transformer.
5.3.9 Calculation of Wire Size for Secondary Coils (N
S1, NS2, NS3) ç Coil size is calculated using the rated output power. Cross-sectional area of the primary coil is calculated using equation 16 in 5.2.2. With current density（α）set at 6A/mm (min)on(max)DC(min) O NP ftVη3α PD2A ××= 0.210 1029.61022.131080.8536 81.20.6552 ××= ［mm A diameter of 0.50mm is therefore appropriate for the wire size of the primary coil. toff(max) is first calculated using equation 5 in 5.2.2. 11.73102.51)(13559 1022.1310831t)V(VN tVNt 6-6 q F1O1P ON(max)DC(min)S1 off(max) =×++× ×××=++× 0.165 1029.6)102.510(11.7336 0.45)1029.610(2.50.65512 f)t(t3α I)f(tD12A 36 6(min)qoff(max) O1(min)q NS1 = − ［mm 0.146 1029.6)102.510(11.7336 0.40)1029.610(2.50.65512 f)t(t3α I)f(tD12A 36 6(min)qoff(max) O2(min)q NS2 = − ［mm 0.146 1029.6)102.510(11.7336 0.40)1029.610(2.50.65512 f)t(t3α I)f(tD12A 36 6(min)qoff(max) O3(min)q NS3 = − ［mm The wire sizes for the secondary coils are therefore as follows. N S1: 0.32mm dia. x 2 wires NS2: 0.29mm dia. x 2 wires NS3: 0.29mm dia. x 2 wires The structure of the transformer requires that all turns in coil N S1 be in a single layer. Fig.5.1 Transformer Specifications and Coil Structure NP1 NP2 NC NS1 NP1=37［Turn］ 0.50mmφ NP2=22［Turn］ 0.50mmφ NS2 NC=4［Turn］ 0.20mmφ NS3=4［Turn］ 0.30mmφ×2wires NP2 NP1 NS2 NS1 NC Spacer Spacer NS3 NS3 NS1=31［Turn］ 0.30mmφ×2wires NS2=8［Turn］ 0.30mmφ×2wires Primary inductance (LP): 0.65mH (between transformer pins ① and ③) Gap Ig: 0.87mm
Shindengen Electric MFG.CO.,LTD - 18 - 5.4.1 Values of Constants for MR2900 Peripheral Components (see 4. Standard Circuit on P12) Component Constant C107 This is the power supply voltage rectification condenser. If this value is small operation at start-up readily becomes intermittent, and if it is too large start-up time becomes excessive. A value of between 47μF and 100μF is appropriate. C108 This condenser determines the resonance frequency. Select the value on the basis of noise and efficiency etc. A value of between 820pF and 2200pF is appropriate for autosensing power supplies of between 75W and 150W capacity. C109 This condenser is incorporated to deal with noise at pin 2. A value of approximately 4700pF is appropriate. Also beneficial in gain phase adjustment, however frequency response deteriorates if the value is too large. C111 This is the partial resonance adjustment condenser. Adjust so that turn-on occurs at the resonance trough. Turn-on occurs earlier if this value is small, and later if it is large. A value of between 10pF and 33pF is appropriate. R102 This is the current limiting damper resistor for C108. A value up to a few ohms is appropriate. Select the value on the basis of noise and efficiency etc. R103 This is the over-current detection resistor. It determines the droop point. Calculate the resistance value as follows. [0.60 (over-current threshold voltage) / Droop point collector current at minimum input] R104 Adjust on the basis of droop characteristics. Set to a value slightly higher than the droop point set with R103. A value of a few tens of kohms is appropriate. R105 This resistor compensates for droop due to input voltage. Adjust on the basis of droop characteristics. A value of approximately 50kohms is appropriate. R106 This resistor limits current at the Z/C pin. A value of approximately 20kohms is appropriate. D102 This corresponds to the body diode for the main switching device (ultra high-speed IGBT). Select a high-speed diode in the 900V, 1A class. D106 This is a Zener diode to compensate for droop due to input voltage. Select a diode for a Zener voltage at least equal to that found with the following equation. Zener voltage P C N N1501.3 ××= (assume an initial compensation voltage of 150V) R105 and D106 are additional components for autosensing input specifications.
Shindengen Electric MFG.CO.,LTD - 19 - Tj(max) for the MR Series is 150°C. As operation of the MR Series is accompanied by an increase in temperature associated with power losses, it is necessary to consider the type of heat sink to be used. While a design which ensures that T j(max) is not exceeded is of absolute importance, the overheat protection function (TSD=150°C (typical)) must be also considered in any design. The extent to which Tj is derated in a design is therefore extremely important in improving reliability. The majority of power losses during operation of the MR Series are associated with the internal MOSFET. If the majority of power losses are considered as ON losses, they may be expressed by the following equation. PD =VDS ×ID The temperature increase (ΔTj) due to power losses (PD) is expressed as, ΔTj +Ta ≦Tj(max) and if TSD=150°C (typical) and TSD(min)=120°C are assumed, PD is limited so that the following equation is satisfied. ç ΔTj+Ta≦TSD(min) Tj may be calculated as follows using the thermal resistance θja. Tj =( PD ×θja) +Ta θja is the thermal resistance in the vicinity of the junction, and is expressed as follows. θja =θjc +θcf +θfa Thermal shutdown (TSD) is a protective function which stops and latches operation at 150°C in the event of abnormal heating of the MR1520. Circuit design therefore requires a cooling design in which temperature has been sufficiently derated. Shindengen recommends that cooling design be such that case temperature does not exceed 100°C.
6.1 Junction Temperature and Power Losses
6.2 Junction Temperature and Thermal Resistance
Thermal resistance between junction and vicinity. θja ℃/W Thermal resistance between junction and case. θjc ℃/W Thermal resistance between case and fins (contact thermal resistance). θcf ℃/W Thermal resistance between case and fins (contact thermal resistance). θfa ℃/W
6.3 Cautions for Cooling Design
- Cooling Design MR2900 Application Note