MIC3232YMM

MIC3230 MICREL | Alldatasheet

Document overview

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Technical content

Features

• 6V to 45V input supply range
• Capable of driving up to 70W
• Ultra low EMI via dithering on the MIC3231
• Programmable LED drive current
• Feedback voltage = 250mV ±3%
• Programmable switching frequency (MIC3230/1) or 400kHz fixed frequency operation (MIC3232)
• PWM Dimming and separate enable shutdown
• Frequency synchronization with other MIC3230s
• Protection features: Over Voltage Protection (OVP) Over temperature protection Under-voltage Lock-out (UVLO)
• Packages: IADJ IS 65

1 VIN

10 VDD

1 VDD

1 N/C

16 N/C

• –40°C to +125°C junction temperature range

Applications

• Street Lighting
• Solid State Lighting
• General Illumination
• Architectural Lighting
• Constant Current Power Supplies January 2009 M9999-011409-A (408) 955-1690

Micrel, Inc. MIC3230/1/2 January 2009 2 M9999-011409-A Typical Application L 47µH D1 100k 4.33k COUT 4.7µF 100V 100k RADJ 1/4W RFS 16.5k CCOMP 10nF CIN 4.7µF/50v RSLC VFB = 0.25V RCS 1/2W Analog ground Power ground VOUTVIN PWMD ENABLE Synch to other MIC3230 ILED Return LED 1 LED XQ1 COMP PWMD VDD AGND PGND EPAD IS OVP DRV VIN EN IADJ MIC3230/31 SYNC FS 10µF 10V Figure 1. Typical Application of the MIC3230 LED Driver

Ordering Information

Part Number Temperature Range Package Lead Finish MIC3230YTSE –40° to +125°C EPAD TSSOP-16 Pb-Free MIC3230YML –40° to +125°C 3mm x 3mm MLF®-12L Pb-Free MIC3231YTSE –40° to +125°C EPAD TSSOP-16 Pb-Free MIC3231YML –40° to +125°C 3mm x 3mm MLF®-12L Pb-Free MIC3232YMM –40° to +125°C MSOP-10 Pb-Free

Micrel, Inc. MIC3230/1/2 January 2009 3 M9999-011409-A Pin Configuration IADJ IS 65 MSOP-10 (MM) MIC3232 3mmx3mmMLF®-12L (ML) MIC3230, MIC3231 See Product Option Matrix for selection TSSOP-16 (TSE) MIC3230, MIC3231 See Product Option Matrix for selection Pin Description Pin Number 3x3MLF Pin Number TSSOP-16L Pin Number MSOP-10L Pin Name Pin Function -- 1 -- NC No Connect 1 2 1 VIN Input Voltage (power) 6V to 45V 2 3 2 EN Enable Control (Input). Logic High (≥1.5V) enables the regulator. Logic Low (≤0.4V) shuts down the regulator. Connect a 100kΩ resistor from EN to VIN. 3 4 3 PWMD PWM input. High signal terminates the output power. Low Signal starts up the output power. 4 5 4 COMP Compensation (output): for external compensation 5 6 5 IADJ Feedback (input) 6 7 -- FS Frequency Select (input). Connected to a Resistor to determine the operating frequency -- 8 -- AGND Analog Ground -- 9 -- NC No Connect 7 10 -- SYNC Sync (output). Connect to another MIC3230 to synchronize multiple converters. 8 11 6 IS Current Sense (input). Connected to external current sense resistor which in turn is connected to the source of the external FET as well as an external slope compensation resistor 9 12 7 OVP OVP divider connection (output). Connect the top of the divider string to the output. If the load is disconnected, the output voltage will rise until OVP reaches 1.25V and then will regulate around this point 10 13 8 PGND Power Ground 11 14 9 DRV Drive Output: connect to the gate of external FET (output) 12 15 10 VDD VDD Filter for internal power rail. Do not connect an external load to this pin. Connect 10µF to GND. -- 16 -- NC No Connect -- -- -- EPAD Connect to AGND

Micrel, Inc. MIC3230/1/2 January 2009 4 M9999-011409-A Absolute Maximum Ratings(1) Operating Ratings(2) Junction Thermal Resistance Electrical Characteristics(4) VIN = 12V; VEN = 3.6V; L = 47µH; C = 4.7µF; TJ = 25°C, Bold values indicate –40°C≤ TJ ≤ +125°C, unless noted. Symbol Parameter Condition Min Typ Max Units VIN Supply Voltage Range 6 45 V UVLO Under Voltage Lockout 3.5 4.9 5.5 V IVIN Quiescent Current VFB > 275mV (to ensure device is not switching) 3.2 10 mA ISD Shutdown Current VEN = 0V 30 µA Room temperature (3%) 242.5 250 257.5 mV VIADJ Feedback Voltage (at IADJ) IADJ Feedback Input Current VFB = 250mV 1.2 3 µA Line Regulation VIN = 12V to 24V 2 % Load Regulation VOUT to 2 × VOUT 2 % DMAX Maximum Duty Cycle MIC3230 & MIC3232 MIC3231 VEN Enable Threshold Turn ON Turn OFF 1.5 1.15 1.1 0.4 V V IEN Enable Pin Current VEN = 3.3V REN = 100kΩ 17 30 µA VPWM PWMD Threshold Turn ON Turn OFF 1.5 0.75 0.7 0.4 V V fPWMD PWMD Frequency Range Note 5 (L = 47µH; C = 4.7µF) 0 500 Hz fSW Programmable Oscillator Frequency RFREQ = 82.5kΩ RFREQ = 21kΩ RFREQ = 8.25kΩ 360 109 400 950 440 kHz kHz kHz fSW Fixed Frequency Option (MIC3232YMM) 360 400 440 kHz FDITHER Low EMI (MIC3231) Frequency dither shift from nominal ±12 % VSENS Current Limit Threshold Voltage RSENSE = 390Ω 0.315 0.45 0.585 V ISENSE I SENSE peak current out RSENSE = 390Ω 250 µA VOVP Over Voltage Protection 1.203 1.24 1.277 V Driver Impedance SINK SOURCE 2.4 3.5 Ω Ω

Micrel, Inc. MIC3230/1/2 January 2009 5 M9999-011409-A VDRH Driver Voltage High VIN = 12V 7 9 11 V TJ Over-Temperature Threshold Shutdown 150 °C Thermal Shutdown Hysteresis 5 °C Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 4. Specification for packaged product only. 5. Guaranteed by design

Micrel, Inc. MIC3230/1/2 January 2009 6 M9999-011409-A Typical Characteristics

Micrel, Inc. MIC3230/1/2 January 2009 7 M9999-011409-A 11.8 11.85 11.9 11.95 12.05 12.1 12.15 12.2 0 25 50 75 100 125 150 OUTPUT VOLTAGE (V) LOAD (mA) Load Regulation VIN = 3.6V

the typical application schematic is shown in Figure 1. above or below the input voltage. impedance helps provide higher operating efficiency. Figure 2. MIC3230 Functional Block Diagram

Micrel, Inc. MIC3230/1/2 January 2009 9 M9999-011409-A Power Topology Constant Output Current Controller The MIC323x family are peak current mode boost controllers designed to drive high power LEDs. Unlike a standard constant output volt age controller, the MIC323x family has been designed to provide a constant output current. The MIC323x family is designed for a wide input voltage range, from 6V to 45V. In the boost configuration, the output can be set from V IN up to 100V. As a peak current mode controller, the MIC323x family provides the benefits of superior line transient response as well as an easier to design compensation. This family of LED drivers features a built-in soft-start circuitry in order to prevent start-up surges. Other protection features include:

• Current Limit (I LIMIT) - Current sensing for over current and overload protection
• Over Voltage Protection (OVP) - Output over voltage protection to prevent operation above a safe upper limit
• Under Voltage Lockout (UVLO) – UVLO designed to prevent operation at very low input voltages Setting the LED Current The current through the LED string is set via the value chosen for the current sense resistor, RADJ. This value can be calculated using Equation 1: Eq. (1) ADJ LED R VI 25 . 0= Another important parameter to be aware of in the boost controller design, is the ripple current. The amount of ripple current through the LED string is equal to the output ripple voltage divided by the LED AC resistance (R LED – provided by the LED manufact urer) plus the current sense resistor (RADJ). The amount of allowable ripple through the LED string is dependent upon the application and is left to the designer’s discretion. This equation is shown in Equation 2: Eq. (2) )( ADJLED OUT LED R R V I RIPPLE +≈ Δ Where OUT LED OUT C T D IV RIPPLE × ×= Reference Voltage The voltage feedback loop of the MIC323x uses an internal reference voltage of 0.25V with an accuracy of ±3%. The feedback voltage is the voltage drop across the current setting resistor (RADJ) as shown in Figure 1. When in regulation the voltage at IADJ will equal 0.25V. Output Over Voltage Protection (OVP) The MIC323x provides an OVP circuitry in order to help protect the system from an overvoltage fault condition. This OVP point can be programmed through the use of external resistors (R8 and R9 in Figure 1). A reference value of 1.245V is used for the OVP. Equation 3 can be used to calculate the resistor value for R9 to set the OVP point. Eq. (3) 1 ) 245 . 1 / ( 89 −= OVPV RR LED Dimming The MIC323x family of LED drivers can control the brightness of the LED string via the use of pulse width modulated (PWM) dimming. A PWM input signal of up to 500Hz can be applied to the PWM DIM pin (see Figure 1) to pulse the LED string ON and OFF. It is recommended to use PWM dimming signals above 120Hz to avoid any recognizable flicker by the human eye. PWM dimming is the preferred way to dim a LED in order to prevent color/wavelength shifting, as occurs with analog dimming. The output current level remains constant during each PWMD pulse. Oscillator and Switching Frequency Selection The MIC323x family features an internal oscillator that synchronizes all of the switching circuits internal to the IC. This frequency is adjustable on the MIC3230 and MIC3231 and fixed at 400kHz in the MIC3232. In the MIC3230/1, the switching frequency can be set by choosing the appropriate value for the resistor, R according to Equation 4: Eq. (4) 035 . 1 ) ( 7526) ( ⎟⎟ ⎛= Ω kHz Fk R SW FS SYNC (MIC3230 Only) Multiple MIC3230 ICs can be synchronized by connecting their SYNC pins together. When synchronized, the MIC3230 with the highest frequency (master) will override the other MIC3230s (slaves). The internal oscillator of the master IC will override the os cillator of the slave part(s) and all MIC3230 will be synchronized to the same master switching frequency. The SYNC pin is designed to be used only by other MIC3230s and is available on the MIC3230 only. If the SYNC pin is being unused, it is to be left floating (open). In the MIC3231, the SYNC pin is to be left floating (open).

emission by approximately 10dB. is the bias supply for the inter nal circuitry of the MIC323x. should not be utilized for operation. Figure 3. Slope compensation waveforms

Figure 6. IS pin and VRCS (Ch1 = Switch Node, Ch2 = IS pin, Table 2. Design example parameters

Figure 7. Design Example Schematic to drive six LEDs at 350mA with a ripple of about 12%. resistor must be chosen using Equation 3. Use the closest standard value resistor of 16.5kΩ. closest value from a resistor manufacture. operating conditions listed in the design example table.

Micrel, Inc. MIC3230/1/2 January 2009 14 M9999-011409-A rms AV eff IVI nom IN nomOUTnomOUT nomRMS IN _ 78 . 0 _ _ =× rms AV eff IVI IN OUTOUT RMS IN _ 48 . 0 max _ min _min _ min _ _ =× Iout is the same as ILED Selecting the inductor current (peak-to-peak), IL_PP, to be between 20% to 50% of IIN_RMS_nom, in this case 40%, we obtain: P Pnom rms innom PP in AII −=== 31 . 0 78 . 0 * 4 . 04 . 0 _ __ _ (see the current waveforms in Figure 5). It can be difficult to find large inductor values with high saturation currents in a surface mount package. Due to this, the percentage of the ripple current may be limited by the available inductor. It is recommended to operate in the continuous conduction mode. The selection of L described here is for continuous conduction mode. Eq. (13) PP in IN I T D VL × ×= Using the nominal values, we get: HA sVL μμ 4331 . 0 2 56 . 0 12=× ×= Select the next higher standard inductor value of 47µH. Going back and calculating the actual ripple current gives: Eq. (13a) PP nomnom IN PP in Auh usv L T DVI 29 . 047 2 56 . 0 12_ The average input current is different than the RMS input current because of the ripple current. If the ripple current is low, then the average input current nearly equals the RMS input current. In the case where the average input current is different than the RMS, Equation 10 shows the following: Eq. (13b) () ( ) max _ _max _ _ PP IN RMS INAVE IN III −= () () AI AVE IN 64 . 1 12 / 29 . 0 64 . 122 max _ _ ≈−= The Maximum Peak input current I L_PK can found using equation 11: AIII PP LAVE INPK L 78 . 15 . 0 max _ _max _ _max _ _ =×+= The saturation current (I SAT) at the highest operating temperature of the inductor must be rated higher than this. The power dissipated in the inductor is: Eq. (13c) DCRIP RMS inINDUCTOR ×= 2 max _ _ Current Limit and Slope Compensation Having calculated the IL_pk above, We can set the current limit 20% above this maximum value: A AI Limitpk L 9 . 1 6 . 1 2 . 1_ = × = The internal current limit comparator reference is set at 0.45V, therefore when , the IC enters current limit. 45 . 0_ =PIN ISV Eq. (14) ( )PKA Vcs V PK +=45 . 0 Where is the peak of the waveform and is the peak of the Vcs waveform PKAV AV PKVcs Eq. (14a) CSpk LSLCRAMP R I D R I Limit ×+× ×= _45 . 0 To calculate the value of the slope compensation resistance, RSLC, we can use Equation 5: ( ) SW CSINOUT SLC F A L R VV R MINMAX × × × − = μ250 First we must calculate RCS, which is given below in Equation 15: Eq. (15) Limitpk L SW MINMAX CS IF L DVINVOUTR _max 45 . 0 × −= Therefore; () ( ) Ω = × −= m AkHzH v vRCS 179 9 . 150047 50 . 0 8 28 45 . 0 μ Using a standard value 150mΩ resistor for RCS, we obtain the following for RSLC: ( ) Ω =× × Ω × −= 51150025047 150 8 28 kHz A H mRSLC μ μ Use the next higher standard value if this not a standard value. In this example 511Ω is a standard value. Check: Because we must use a standard value for Rcs and RSLC; may be set at a different level (if the calculated value isn’t a standard value) and we must calculate the actual value (remember is the same as ). Limitpk LI _ Limitpk LI _ Limitpk in_Limitpk LI _ I Rearranging Equation 14a to solve for : Limitpk LI _ CS SLCRAMP pk in R D R II Limit )45 . 0 ( × × −= AuaI Limitactual in 34 . 2150 . ) 75 . 0 511250 45 . 0 ( _ =× × −= This is higher than the initial limit because we have to use standard values for R CS AI PK L 9 . 12 . 1 max _ _=×

Micrel, Inc. MIC3230/1/2 January 2009 15 M9999-011409-A and for RSLC. If is too high than use a higher value for R CS. The calculated value of R CS for a 1.9A current limit was 179m Ω. In this example, we have chosen a lower value which results in a higher current limit. If we use a higher standard value the current limit will have a lower value. The designer does not have the same choices for small valued resistors as with larger valued resistors. The choices differ from resistor manufacturers. If too large a current sense resistor is selected, the maximum output power may not be able to be achieved at low input line voltage levels. Make sure the inductor will not saturate at the actual current limit Limitactual inI _ Limitactual inI _ V PIN IS250_ = Perform a check at IIN=2.34Apk. Vm A 45 . 0 150 34 . 2= Ω × +μ Maximum Power dissipated in RCS is; Eq. (17) CSR RMS R RI P CSCS Eq. (18) +== _max _ RMSFETI RMS 12 max _ _max __ PP L AVE IN I I D CSRI rms A RMS _ 44 . 112 26 . 064 . 1 78 . 0 22 =⎟ +=I CSR _ wattP CSR 31 . 0 15 . 25 . 12 = × = Use a 1/2 Watt resistor for RCS. Output Capacitor In this LED driver application, the ILED ripple current is a more important factor compared to that of the output ripple voltage (although the two are directly related). To find the C OUT for a required ILED ripple use the following calculation: For an output ripple 20% of =rippleILED nomILED mAILEDripple 70 35 . 0 2 . 0= × = Eq. (19) )( * * * _ total LEDadjripple nomnom out R RILED T DILEDC += Find the equivalent ac resistance from the datasheet of the LED. This is the inverse slope of the ILED vs. VF curve i.e.: ac LEDR _ Eq. (20) ILED VR F ac LED Δ Δ=_ In this example use for each LED. Ω =1 . 0_ ac LEDR If the LEDs are connected in series, multiply s, we obtain the following: Ω =1 . 0_ ac LEDR by the total number of LEDs. In this example of 6 LED Ω= Ω × =6 . 0 1 . 0 6_ total LEDR uFR RILED T DILEDC total LEDadjripple nomnom out 1 . 4)( * * * =+= Use the next highest standard value, which is 4.7uF. e is shown in Figure 5. For superior There is a trade off between the output ripple and th rising edge of the PWMD pulse. This is because between PWM dimming pulses, the converter stops pulsing and COUT will start to discharge. The amount that COUT will discharge depends on the time between PWM Dimming pluses. At the next PWMD pulse C OUT has to be charged up to the full output voltage V OUT before the desired LED current flows. Input Capacitor The input current performance, ceramic capacitors should be used because of their low equivalent series resistance (ESR). The input ripple current is equal to the ripple in the inductor plus the ripple voltage across the input capacitor, which is the ESR of C IN times the inductor ripple. The input capacito r will also bypass the EMI generated by the converter as well as any voltage spikes generated by the inductance of the input line. For a required V IN_RIPPLE: Eq. (21) () FkHzmV A FV I C = SWRIPPLEIN PP IN IN μ4 . 150050 8 28 . 0 8 _ This is the minimum value that should be used. The e, the FET has to hold off an output input capacitor should also be rated for the maximum RMS input current. To protect the IC from inductive spikes or any overshoot, a larger value of input capacitance may be required and it is recommended that ceramic capacitors be used. In this design example a value of 4.7µF ceramic capacitor was selected. MOSFET Selection In this design exampl voltage maximum of 30V. It is recommended to use an 80% de-rating value on switching FETs, so a minimum of a 38V FET should be selected. In this design example, a 75V FET has been selected. The switching FET power losses are the sum of the conduction loss and the switching loss: Eq. (22) FETCONDFETFET PP P __ + SWITCH= The conduct ion loss of the FET is when the FET is , where turned on. The conduction power loss of the FET is found by the following equation: Eq. (23) FETCONDFET IP = __ DSONRMS R×2

Micrel, Inc. MIC3230/1/2 January 2009 16 M9999-011409-A += 12 _ 2 PP L AVE INRMS FET II DI The switching losses occur during the switching transitions of the FET. The transition times, ttransition, are ere are the times when the FET is turning off and on. Th two transition times per period, T. It is important not to confuse T (the period) with the transition time, ttransition. Eq. (24) FswT 1= Eq. (25) SWtransitionOUTAVEFETSWITCHFET FtVIP ×××= max _max _max _ _max __ max _transitiont : To find Eq. (26) Igatedrv Qgttransition ≈max _ is the total gate c rge of the external MOSFET provided by the MOSFET manufacturer and charge at a where Qg ha the Qg ould chosen at a V GS≈10V. This is not an exact value, but is more of an estimate of max _transitiont . The FET manufacturers’ provide a gate specified V GS voltage: sh GS G In QC _ =FET V@ This is the FET’s input capacitance. Select a FET with RDS(on) and Q G such that the external power is below harge=68nC (typical) The perature. As the erature at 125°C is: ation 23: about 0.7W for a SO-8 or about 1W for a PowerPak (FET package). The Vishay Siliconix Si7148DP in a PowerPak SO-8 package is one good choice. The internal gate driver in the MIC3230/1/2 is 2A. From the Si7148DP data sheet: R DS(on)_25°C=0.0145Ω Total gate C ) is a function of tem(( tempR on DS) in the FET increases so does the temp RDS(on). To find ) () (tempR on DS use Equation 27, or simply o odouble the ) for ) 125 () ( CR on DS . Eq. (27) (on DSR 25 () ( CR on DS )) ( oo −×= Temp on DS CR temp The (RDS ) () tempon Ω ≈−∗ mCRDSon 30 ) 7) 125 ( ) 25 125 (oo × =00 . 1 ( 0145 . 0 From Equ mWmP CONDFET 6230 64 . 12 _ = Ω × = From Equation 26: nsA nC Igatedrv Qgttransition 342 68 = =≈ AAVE 64 . 1max _ _= IFET VVOUT 28max _= From Equation 25: WattskHzns V A 78 . 0500 34 28 64 . 1P SWITCHFET max __ =×× ×= From Equation 22 WWmW 84 . 0 78 . 0 62= +PFET = This about the limit for a part on a circuit board without having to use any additional heat sinks. is best used here because of the lower d the low reverse recovery time. The Rectifier Diode A Schottky Diode forward voltage an voltage stress on the diode is the max V OUT and therefore a diode with a higher rating than max V OUT should be used. An 80% de-rating is recommended here as well. Eq (28) ⎞⎛ 2 _diode I I +− = ) 1 ( _2 max _ _max _ PP L AVE INRMS I D Eq. (29) WP I V P diode RMSdiodeSCHOTTKYdiode 81 . 0 max _ _ MIC3230 power losses IC3230are: is the total gate charge of the The power losses in the M Eq.(30) F V QP gategateMIC × × =3230 Vin IQ × + where gatQ external e r ati MOSFE gateV is the gate drive voltage of the MIC3230. F is the switching frequency. QI is the quiescent cu rent of the MIC3230 found in th lectrical characteriz on table. mA IQ 2 . 3= . VIN is the voltage at the VIN pin of the MIC3230. From Eq.(30) mAkHznFPMIC 14 2 . 3 500 12 683230 e e W45 . 0=×+××=

Micrel, Inc. MIC3230/1/2 January 2009 17 M9999-011409-A OVP-Over voltage protection 2. Even though the RRC is very short (tens of nanoseconds) the peak currents are high (multiple amperes). The high RRC causes a voltage drop on the ground trace of the PCB and if the converter control IC is referenced to this voltage drop, the output regulation will suffer. Set OVP higher than the maximum output voltage by at least one volt. To find the resistor divider values for OVP use Equation 3 and set the OVP=30V and R8=100kΩ: Ω =− × Ω= kkR 33 . 4245 . 1 30 245 . 1 1009 3. It is important to connec t the IC’s reference to the same point as the output capacitors to avoid the voltage drop caused by RRC. This is also called a star connection or single point grounding. PCB Layout 1. All typologies of DC-to- DC converters have a reverse recovery current (RRC) of the flyback or (freewheeling) diode. Even a Schottky diode, which is advertised as having zero RRC, it really is not zero. The RRC of the freewheeling diode in a boost converter is even greater than in the Buck converter. This is because the output voltage is higher than the input voltage and the diode has to charge up to –V OUT during each on-time pulse and then discharge to VF during the off-time. 4. Feedback trace: The high impedance traces of the FB should be short.

Micrel, Inc. MIC3230/1/2 January 2009 18 M9999-011409-A

Package Information

10-Pin MSOP (MM)

Micrel, Inc. MIC3230/1/2 January 2009 19 M9999-011409-A 12-Pin 3mm × 3mm MLF® (ML)

Micrel, Inc. MIC3230/1/2 January 2009 20 M9999-011409-A 16-Pin Exposed Pad TSSOP (TSE) MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical impla into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. can nt © 2009 Micrel, Incorporated.