MIC3223YTSE

MIC3223 MICREL | Alldatasheet

Document overview

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• PDF pages: 24

Technical content

Features

• 4.5V to 20V supply voltage
• 200mV feedback voltage with an accuracy of ±5%
• Step-up output voltage (boost) conversion up to 37V
• 1MHz switching frequency
• 100m Ω/3.5A internal power FET switch
• LEDs can be dimmed using a PWM signal
• User settable LED current (through external resistor)
• Externally programmable soft-start
• Protection features that include: – Output over-voltage protection (OVP) – Under-voltage lockout (UVLO) – Over temperature protection
• Junction temperature range: -40°C to +125°C
• Available in a exposed pad 16-pin TSSOP package

Applications

• Architectural lighting
• Industrial lighting
• Signage
• Landscape lighting (garden/pathway)
• Under cabinet lighting
• MR-16 bulbs Typical Application

Micrel, Inc. MIC3223 January 2010 2 M9999-011510-A

Ordering Information

Part Number Junction Temp. Range Package Lead Finish MIC3223YTSE –40° to +125°C 16-pin ePad TSSOP PB- free Pin Configuration 16-Pin ePad TSSOP (TSE) Pin Description Pin Number Pin Name Pin Function 1 EN Enable (Input): Logic high enables and logic low disables operation. 2 SS Soft Start (Input resistance of 30k). Connect a capacitor to GND for soft-start. Clamp the pin to a known voltage to control the internal reference voltage and hence the output current. 3 COMP Compensation Pin (Input): Add external R and C-to-GND to stabilize the converter.

4 FB Negative Input to Error Amp

5 OVP Connect to the centre tap of an external resistor divider, the top of which is tied to Vout

and bottom-to-ground.

6 PGND Power Ground

7,8,9,10 SW Switch Node (Input): Internal NMOS switch Drain Pin

11 VIN Input Supply

12 DRVVDD For 4.5V < VIN < 6V, connect DRVVDD to VIN. DRVVDD is the input voltage supply for the converter’s internal power FET gate driver. For VIN > 6V, connect this pin to VDD. 13 VDD For 4.5V < VIN < 6V, this pin becomes the input voltage supply for the converter’s internal circuit. For VIN > 6V, this pin is an output of the internal 5.5V regulator that supplies internal circuits. User must add 10µF decoupling capacitor from VDD-to-AGND. 14 DIM_IN PWM input to control LED dimming. 15 DIM_OUT Output driver to drive external FET for LED dimming.

16 AGND Analog Ground

17 EP Connect to Power Ground

Micrel, Inc. MIC3223 January 2010 3 M9999-011510-A Absolute Maximum Ratings(1) Operating Ratings(2) Junction Thermal Resistance Electrical Characteristics(4) VIN = VEN = 12V; L = 22µH, CIN =4.7µF, COUT =2x4.7µF; TA = 25°C, BOLD values indicate –40°C≤ TJ ≤ +125°C, unless otherwise noted. Symbol Parameter Condition Min Typ Max Units VIN Voltage Supply Range 4.5 20 V VUVLO Under Voltage Lockout Monitoring for V DD 3 3.7 4.4 V VOVP Over Voltage Protec tion 1.216 1.28 1.344 V IVIN Quiescent Current V FB=250mV 2.1 5 mA ISD Shutdown Current V EN =0V 10 µA Room Temperature 190 200 210 mV VFB Feedback Voltage Over Temperature 184 216 mV IFB Feedback Input Current V FB=200mV -450 nA VDD Internal Voltage Regulator 5.3 V DMAX Maximum Duty Cycle 85 90 95 % V DD Line Regulation V LED=18V, VIN=8V to 16V, ILED=350mA 0.5 % ISW Switch Current Limit 3.5 9 10.5 A RSW Switch R DSON plus RCS 100 m Ω ISW Switch Leakage Current V EN=0, VSW=37V 0.01 10 µA Turn On 1.5 V VEN Enable Threshold Turn Off 0.4 V IEN Enable Pin Current 20 40 µA VDIM_TH_H DIM_IN Threshold High Logic High 1.5 V VDIM_TH_L DIM_IN Threshold Low Logic Low 0.4 V Hys DIM_IN Hysteresis 500 mV IDIM_IN DIM_IN Pin Current V DIM_IN = 5V 1 µA TDR Dim Delay (Rising) DIM_IN Rising 40 ns TDF Dim Delay (Falling DIM_IN Falling 30 ns

Micrel, Inc. MIC3223 January 2010 4 M9999-011510-A Symbol Parameter Condition Min Typ Max Units 0.7 1.3 µs DIM MIN Minimum Dimming Pulse DIM_IN =1µs CDIM_OUT = 1.25nF DIM_OUT measured from 4V rising to 2.5 falling 0.5 1.5 µs RDO DIM_OUT Resistance High DIM_OUT pull up resistance IDIM_OUT = +2mA 70 Ω RDO DIM_OUT Resistance Low Dim Out pull down resistance IDIM_OUT = -2mA 40 Ω FSW Oscillator Frequency 0.7 1 1.3 MHz RSS Soft Start Resistance 30 46 62 k Ω Temperature rising 165 °C TSD Over Temperature Threshold Shutdown Hysteresis 10 °C Notes 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 4. Specification for packaged product only. Test Circuit

Micrel, Inc. MIC3223 January 2010 5 M9999-011510-A Typical Characteristics Feedback Voltage v s. Input Voltage 0.190 0.192 0.194 0.196 0.198 0.200 0.202 0.204 0.206 0.208 0.210 4 9 14 19 INPUT VOLTAGE (V) REFERENCE VOTLAGE (V) VOUT = 30V IOUT =0.36A Switching Frequency v s. Input Voltage 0.9 1.0 1.0 1.1 1.1 1.2 49 1 4 1 9 INPUT VOLTAGE (V) SWITCHING FREQUENCY (MHz) T = 25°C VDD = VIN VIN = 4.5V to 6V VOUT = 30V IOUT = 0.36A RSW_NODE vs. Temperature 0.10 0.11 0.12 0.13 0.14 0.15 0.16 0.17 0.18 -40 -20 0 20 40 60 80 100 120 TEMPERA TURE (°C) RSW_NODE ( Ω) VIN = 12V VOUT = 36V ISW = 1.3A Switching Frequency vs. Temperature 0.80 0.85 0.90 0.95 1.00 1.05 1.10 1.15 1.20 -40 -20 0 20 40 60 80 100 120 TEMPERA TURE (°C) SWITCHING FREQUENCY (MHz) VIN = 12V VOUT = 26V IOUT = 0.36A Efficiency vs. Output Current 0 0.5 1 1.5 OUTPUT CURRENT (A ) EFFICIENCY (%) VOUT = 25V 10V 12V Efficiency vs. Output Current 00 . 511 . 5 OUTPUT CURRENT (A ) EFFICIENCY (%) 14V 16V VOUT = 25V Efficiency vs. Output Current 0 0.5 1 1.5 OUTPUT CURRENT (A ) EFFICIENCY (%) 18V 20V VOUT = 25V Efficiency vs. Input Voltage 51 0 1 5 2 0 INPUT VOLTAGE (V) EFFICIENCY (%) VOUT = 25V IOUT = 0.5A T = 25°C VDD Voltage v s. Input Voltage 5.00 5.05 5.10 5.15 5.20 5.25 5.30 5.35 5.40 5.45 5.50 51 0 1 5 2 0 INPUT VOLTAGE (V) VDD VOLTAGE (V) T = 25°C VOUT = 25V IOUT = 0.5A Current Limit v s. Input Voltage 7.0 7.5 8.0 8.5 9.0 9.5 4 9 14 19 INPUT VOLTAGE (V) CURRENT LIMIT (A) T = 25°C VIN = 4.5V to 6V Feedback Voltage vs. Temperature 0.200 0.202 0.204 0.206 0.208 0.210 0.212 0.214 0.216 0.218 0.220 -40 -20 0 20 40 60 80 100 120 TEMPERA TURE (°C) FEEDBACK VOLTAGE (V) VIN = 12V VOUT = 26V IOUT = 0.36A Current Limit vs. Temperature 6.0 6.5 7.0 7.5 8.0 8.5 9.0 9.5 10.0 10.5 11.0 -40 -20 0 20 40 60 80 100 120 TEMPERA TURE (°C) CURRENT LIMIT (A) VIN = 12V

Micrel, Inc. MIC3223 January 2010 6 M9999-011510-A Typical Characteristics (continued) Efficiency vs. Output Current 0 0.5 1 1.5 OUTPUT CURRENT (A ) EFFICIENCY (%) VOUT = 25V 10V Efficiency vs. Output Current 0 0.5 1 1.5 OUTPUT CURRENT (A ) EFFICIENCY (%) VOUT = 25V 12V

Micrel, Inc. MIC3223 January 2010 7 M9999-011510-A Functional Characteristics

Micrel, Inc. MIC3223 January 2010 8 M9999-011510-A Functional Characteristics (continued)

Micrel, Inc. MIC3223 January 2010 9 M9999-011510-A Functional Diagram

Micrel, Inc. MIC3223 January 2010 10 M9999-011510-A Functional Description A constant current output converter is the preferred method for driving LEDs. Small variations in current have a minimal effect on the light output, whereas small variations in voltage have a significant impact on light output. The MIC3223 LED driver is specifically designed to operate as a constant current LED Driver. The MIC3223 is designed to operate as a boost converter, where the output voltage is greater than the input voltage. This configuration allows for the design of driving multiple LEDs in series to help maintain color and brightness. The MIC3223 can also be configured as a SEPIC converter, where the output voltage can be either above or below the input voltage. The MIC3223 has an input voltage range, from 4.5V and 20V, to address a diverse range of applications. In addition, the LED current can be programmed to a wide range of values through the use of an external resistor. This provides design flexibility in adjusting the current for a particular application need. The MIC3223 features a low impedance gate driver capable of switching large MOSFETs. This low impedance provides higher operating efficiency. The MIC3223 can control the brightness of the LEDs via its PWM dimming capability. Applying a PWM signal (up to 20kHz) to the DIM_IN pi n allows for control of the brightness of the LEDs. The MIC3223 boost converter employs peak current mode control. Peak current mode control offers advantages over voltage mode control in the following manner. Current mode contro l can achieve a superior line transient performance compared to voltage mode control and is easier to compensate than voltage mode control, thus allowing for a less complex control loop stability design. Page 9 of this datasheet shows the functional block diagram. Boost Converter operation The boost converter is a peak current mode pulse width modulation (PWM) converter and operates as follows. A flip-flop (FF) is set on the leading edge of the clock cycle. When the FF is set, a gate driver drives the power FET on. Current flows from V IN through the inductor (L) and through the power switch and also through the current sense resistor to PGND. The voltage across the current sense resistor is added to a slope compensation ramp (needed for stability). The sum of the current sense voltage and the slope compensation voltages (called V CS) is fed into the positive terminal of the PWM comparator. The other input to the PWM comparator is the error amp output (called V EA). The error amp’s negative input is the feedback voltage (V FB). VFB is the voltage across R ADJ (R5). In this way the output LED current is regulated. If V FB drops, V EA increases and therefore the power FET remains on longer so that V CS can increase to the level of V EA. The reverse occurs when VFB increases. PWM Dimming This control process just described occurs during each DIM_IN pulse and when ever DIM_IN is high. When DIM_IN is low, the boost converter will no longer switch and the output voltage will drop. For high dimming ratios use an external PWM Dimming switch as shown in the Typical Application. When the dim pulse is on the external switch is on and circuit operates in the closed loop control mode as described. When the DIM_IN is low the boost converter does not switch and the external switch is open and no LED current can flow and the output voltage does not droop. When DIM_IN goes high the external switch is driv en on and LED current flows. The output voltage remains the same (about the same) during each on and off DIM_IN pulse. PWM Dimming can also be used in the Test Circuit in applications that do not require high dimming ratios. In the Test Circuit, the load is not removed from the output voltage between DIM_IN pulses and will therefore drain the output capacitors. The voltage that the output will discharge to is determined by the sum of the V F (forward voltage drops of the LEDs). When V OUT can no longer forward bias the LEDs, then the LED current will stop and the output capacitors will stop discharging. During the next DIM_IN pulse V OUT has to charge back up before the full LED current will flow. For applications that do not require high dimming ratios.

Micrel, Inc. MIC3223 January 2010 11 M9999-011510-A

Application Information

Constant Output Current Converter The MIC3223 is a peak current mode boost converter designed to drive high power LEDs with a constant current output. The MIC3223 operates with an input voltage range from 4.5V to 20V. In the boost configuration, the output can be set from V IN up to 37V. The peak current mode control architecture of the MIC3223 provides the advant ages of superior line transient response as well as an easier to design compensation. The MIC3223 LED driver features a built-in soft start circuitry in order to prevent start-up surges. Other protection features include:

• Current Limit (I LIMIT) – Current sensing for over current and overload protection
• Over Voltage Protection (OVP) – output over voltage protection to prevent operation above a safe upper limit
• Under Voltage Lockout (UVLO) – UVLO designed to prevent operation below a safe lower limit Setting the LED Current The current through the LED string is set via the value chosen for the current sense resistor R ADJ which is R5 in the schematic of the Typical Application. This value can be calculated using Equation 1: Eq. (1) ADJR 0.2VILED = Another important parameter to be aware of in the boost converter design is the ripple current. The amount of ripple current through the LED string is equal to the output ripple voltage divided by the LED AC resistance LED – provided by the LED manufacturer) plus the current sense resistor R ADJ. The amount of allowable ripple through the LED string is dependent upon the application and is left to the designer’s discretion. The equation is shown in Equation 2. Eq. (2) )R(R VΔI ADJLED OUT LED RIPPLE Where SWOUT LED OUT FC DIV RIPPLE × Reference Voltage The voltage feedback loop the MIC3223 uses an internal voltage of 200mV with an accuracy of ±5%. The feedback voltage is the volt age drop across the current sense resistor as shown in the Typical Application. When in regulation the voltage at V FB will equal 200mV. Output Over Voltage Protection (OVP) The MIC3223 provides an OVP circuitry in order to protect the system from an overvoltage fault condition. This OVP threshold can be programmed through the use of external resistors (R3 and R4 in the Typical Application). A reference value of 1.245V is used for the OVP. Equation 3 can be used to calculate the resistor value for R9 to set the OVP point. Normally use 100k for R3. Eq. (3) 1/1.245)(V R3R4 OVP −= VDD An internal linear regulator is used to provide the necessary internal bias voltages. When V IN is 6V or below connect the V DD pin to V IN. Use a 10µF ceramic bypass capacitor. DRVVDD An internal linear regulator is used to provide the necessary internal bias volt ages to the gate driver that drives the external FET. When V IN is above 6V connect DRVVDD to VDD. When V IN is 6V or below connect the DRVVDD pin to VIN. Use a bypass capacitor, 10µF ceramic capacitor. UVLO Internal under voltage lock out (UVLO) prevents the part from being used below a safe V IN voltage. The UVLO is 3.7V. Operation below 4.5V is not recommended. Soft Start Soft start is employed to lessen the inrush currents during turn on. At turn on the following occurs; 1. After about 1.5ms C SS will start to rise in a exponential manner according to; −= × )C(37kΩ t SS SSe10.2V 2. According to the block diagram, V SS is the ref node of the error amp. PWM switching start when VSS begins to rise. 3. When the C SS is fully charged, 0.2V will be at the error amp reference and steady state operation begins. 4. Design for soft-start time using the above equation.

Table 1. Design example parameters

Micrel, Inc. MIC3223 January 2010 15 M9999-011510-A Design Example In this example, we will be designing a boost LED driver operating off a 12V input. This design has been created to drive 6 LEDs at 350mA with a ripple of about 20%. We are designing for 80% efficiency at a switching frequency of 1MHz. Select R ADJ Having chosen the LED drive current to be 350mA in this example, the current can be set by choosing the R ADJ resistor from Equation 1: 0.57Ω0.35A 0.2VRADJ == Use the next lowest standard value 0.56Ω. I LED = 0.36A The power dissipation in this resistor is: 71mWRILEDP ADJ RADJ =×= Use a resistor rated at quarter watt or higher. Operating Duty Cycle The operating duty cycle can be calculated using Equation four provided below: Eq. (4) () DIODEOUT DIODEINOUT VV VVVD + +−= VDIODE is the V f of the output diode D1 in the Typical Application. It is recommended to use a schottky diode because it has a lower Vf than a junction diode. These can be calculated for the nominal (typical) operating conditions, but should also be understood for the minimum and maximum system conditions as listed below. ( ) DIODEOUT(nom) DIODEIN(nom)OUT(nom) VV VVVDnom + +−= ( ) DIODEOUT(max) DIODEIN(min)OUT(max) VV VVVDmax + +−= ( ) DIODEOUT(min) DIODEIN(max)OUT(min) VV VVVDmin + +−= () 44.05.021 5.01221Dnom =+ −−= Therefore Dnom = 44%, Dmax = 72% and Dmin = 15%. Inductor Selection First calculate the RMS input current (nominal, min and max) for the system given the operating conditions listed in the design example table. The minimum value of the RMS input current is necessary to ensure proper operation. Using Equation 5, the following values have been calculated: (RMS) IN(min) OUT(max)OUT(max) )IN_RMS(max 1.54AVeff IVI =× Eq (5) (RMS) IN(nom) OUT(nom)OUT(nom) )IN_RMS(nom 0.74AVeff IVI =× (RMS) IN(max) OUT(min)OUT(min) )IN_RMS(min 0.46AVeff IVI =× IOUT is the same as ILED. Selecting the inductor current (peak-to-peak), I L_PP, to be between 20% to 50% of I IN_RMS(nom), in this case 40%, we obtain: I IN_PP(nom) = 0.4 × IIN_RMS(nom) = 0.4 × 0.74 = 0.30AP-P It can be difficult to find large inductor values with high saturation currents in a surface mount package. Due to this, the percentage of the ripple current may be limited by the available inductor. It is recommended to operate in the continuous conduction mode. The selection of L described here is for continuous conduction mode. Eq. (6) V× DINL= I× FIN_PP SW Using the nominal values, we get: 12V × 0.44L= =1 8 μH 0.3A ×1MHz Select the next higher standard inductor value of 22µH. Going back and calculating the actual ripple current gives: PP SW maxIN(min) IN_PP(max) 0.26A1MHzH22 0.728V FL DV I =× ×=× = μ The average input current is different than the RMS input current because of the ripple current. If the ripple current is low, then the average input current nearly equals the RMS input current. In the case where the average input current is different than the RMS, equation 7 shows the following: Eq. (7) () 12 )(I II IN_PP2 )IN_RMS(max)IN_AVE(max −= A54.112 (0.24))54.1(I )IN_AVE(max ≈−= The Maximum Peak input current I L_PK can found using Equation 8: Eq. (8) I L_PK(max) = IIN_AVE(max) + 0.5 ×IL_PP(max) = 1.67A The saturation current (I SAT) at the highest operating temperature of the inductor must be rated higher than this. The power dissipated in the inductor is:

Micrel, Inc. MIC3223 January 2010 16 M9999-011510-A Eq. (9) P INDUCTOR = IIN_RMS(max) 2 × DCR A Coilcraft # MSS1260-223ML is used in this example. Its DCR is 52mΩ, ISAT =2.7A P INDUCTOR = 1.542 × 52 mΩ = 0.123W Output Capacitor In this LED driver application, the I LED ripple current is a more important factor when compared to that of the output ripple voltage (although the two are directly related). To find the C OUT for a required I LED ripple use the following calculation: For an output ripple ILED(ripple) = 20ma Eq. (10) SWLED_totalADJ)LED(ripple nomLED(nom) OUT F)R(RI DI C ×+× Find the equivalent ac resistance RLED_ac from the datasheet of the LED. This is the inverse slope of the ILED vs. Vf curve i.e.: Eq. (11) ΔLED ΔVR f LED_ac = In this example use RLED_ac = 0.6Ω for each LED. If the LEDs are connected in series, multiply RLED_ac = 0.6Ω by the total number of LEDs. In this example of six LEDs, we obtain the following: R LED_total ≡ Rdynamic = 6 × 0.6Ω = 3.6Ω Eq. (12) F1.9F)R(RI DI C SWLED_totalADJ)LED(ripple nomLED(nom) OUT μ=×+× Use 2.2µF or higher. There is a trade off between the output ripple and the rising edge of the DIM_IN pulse. This is because between PWM dimming pulses, the converter stops pulsing and COUT will start to discharge. The amount that C OUT will discharge depends on the time between PWM Dimming pluses. At the next DIM_IN pulse, C OUT has to be charged up to the full output voltage V OUT before the desired LED current flows. Input Capacitor The input capacitor is shown in the Typical Application. For superior performance, ceramic capacitors should be used because of their low equivalent series resistance (ESR). The input capacitor C IN ripple current is equal to the ripple in the inductor. The ripple voltage across the input capacitor, CIN is the ESR of C IN times the inductor ripple. The input capacitor will also bypass the EMI generated by the converter as well as any voltage spikes generated by the inductance of the input line. For a required V IN(ripple): Eq. (13) F0.751MHz50mV8 (0.3A) FV I C SWIN(ripple) IN_PP IN μ=××=×= This is the minimum value that should be used. To protect the IC from inductive spikes or any overshoot, a larger value of input capacitance may be required. Use 2.2µF or higher as a good safe min. Rectifier Diode Selection A schottky diode is best used here because of the lower forward voltage and the low reverse recovery time. The voltage stress on the diode is the max V OUT and therefore a diode with a higher rating than max VOUT should be used. An 80% de-rating is recommended here as well. Eq. (14) I DIODE(max) = IOUT(max) = 0.36A Since IIN_AVE(max) occurs when D is at a maximum. Eq. (15) P DIODE(max) ≈ VDIODE × IDIODE_(max) A SK35B is used in this example, it’s VDIODE is 0.5V P DIODE(max) ≈ 0.5V × 0.36A = 0.18W MIC3223 Power Losses To find the power losses in the MIC3223: There is about 6mA input from VIN into the VDD pin. The internal power switch has an RDS ON of about 170m Ω at. P MIC3223 = VIN × 6mA + PwrFET Eq. (16) PwrFET = I FET_RMS(max) 2 × Rds_on_@100° + V OUT(max) × IIN_AVE(max) × tsw × Fsw R ds_on_@100° ≈ 160mΩ tsw ≈ 30ns is the internal Power FET ON an OFF transition time. 1.3A12 I IDI L_PP2 )IN_AVE(maxSWRMS(max) PwrFET = 1.3A 2 × 160mΩ + 28V × 1.54A × 30ns × 1MHz = 1.6W P MIC3223 = 8 × 6mA + 1.77W = 1.66W Snubber A snubber is a damping resistor in series with a DC blocking capacitor in parallel with the power switch (same as across the flyback diode because V OUT is an ac ground). When the po wer switch turns off, the drain to source capacitance and parasitic inductance will cause a high frequency ringing at the switch node. A snubber circuit as shown in the application schematic may be required if ringing is present at the switch node. A critically damped circuit at the switch node is where R equals the characteristic impedance of the switch node.

Figure 13. MIC3223 Typical Application without External PWM Dimming Switch

1. All typologies of DC-t o-DC converters have a
2. Even though the RRC is very short (tens of

a 25V diode is needed do not use a 100V etc.

1. The high RRC causes a voltage drop on the ground
2. For good output regulation, it is important to connect

capacitors to avoid the voltage drop caused by RRC.

1. Feedback trace: The high impedance traces of the

Micrel, Inc. MIC3223 January 2010 20 M9999-011510-A Evaluation Board Schematic 37V Max 1A LED Driver

Micrel, Inc. MIC3223 January 2010 21 M9999-011510-A Bill of Materials Item Part Number Manufacturer Description Qty GRM319R61E475KA12D muRata (1) C3216X7R1E475M TDK (2) C1 12063D475KAT2A AVX (3) Ceramic Capacitor, 4.7µF, 25V, Size 1206, X7R 1 C2 GRM188R71C273KA01D muRata Ceramic Capacitor, 0.027µF, 6.3V, Size 0603, X7R 1 GRM188R60J106ME47D muRata C1608X5R0J106K TDK C3, C7 08056D106MAT2A AVX Ceramic Capacitor, 10µF, 6.3V, Size 0603, X7R 2 12105C475KAZ2A AVX C4, C6 GRM32ER71H475KA88L muRata Ceramic Capacitor, 4.7µF, 50V, Size 1210, X7R 2 GRM188R71C473KA01D muRata C5 0603YC473K4T2A AVX Ceramic Capacitor, 0.047µF, 6.3V, Size 0603, X7R 1 C8 GRM188R72A102KA37D muRata Ceramic Capacitor, 1000pF, 100V Size 0603, X7R D1 SK35B MCC (4) Schottky Diode, 3A, 50V (SMB) 1 L1 MSD1260-223ML-LD Coilcraft (6) Inductor, 22µH, 5A 1 R1, R3 CRCW0603100KFKEA Vishay Dale (4) Resistor, 100k, 1%, Size 0603 2 R2 CRCW0603549RFKEA Vishay Dale Resistor, 549 Ω, 1%, Size 0603 1 R4 CRCW06033K24FKEA Vishay Dale Resistor, 3.24k, 1%, Size 0603 1 R5 CRCW1206R560FKEA Vishay Dale Resistor, 0.56Ω, 1%, 1/2W, Size 1206 (for .35A LED current Change for different ILED) 1 R6 RMC 1/4 2 1% R Stackpole Electronics, Inc.(7) Resistor, 2Ω, 1%, 1/2W, Size 1210 1 Si2318DS Vishay Siliconix (4) Q1 AM2340N Analog Power (8) N-Channel 40V MOSFET 1 U1 MIC3223 Micrel, Inc. (9) High Power Boost LED Driver with Integrated FET 1 Notes: 1. Murata: www.murata.com. 2. TDK: www.tdk.com. 3. AVX: www.avx.com. 4. Vishay: www.vishay.com. 5. Internacional Rectifier: www.ift.com. 6. Coilcraft: www.coilcraft.com 7. Stackpole Electronics, Inc.: www. 8. Analog Power: www.analogpowerinc.com

Micrel, Inc. MIC3223 January 2010 22 M9999-011510-A PCB Layout Recommendations Top Layer Bottom Layer

Micrel, Inc. MIC3223 January 2010 23 M9999-011510-A

Package Information

16-Pin ePad TSSOP (TSE)

Micrel, Inc. MIC3223 January 2010 24 M9999-011510-A Recommended Land Pattern MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2009 Micrel, Incorporated.