LTC7801

LTC7800 LINER | Alldatasheet

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Technical content

7800fFor more information www.linear.com/LTC7800 TYPICAL APPLICATION FEATURES DESCRIPTION Low IQ, 60V, High Frequency Synchronous Step-Down Controller The LT C®7800 is a high performance step-down switching regulator DC/DC controller that drives an all N-channel synchronous power MOSFET stage. A constant fre - quency current mode architecture allows a phase-lockable frequency of up to 2.25MHz. The 50μA no-load quiescent current extends operating run t ime in battery-powered systems. OPTI-LOOP® compensa- tion allows the transient response to be optimized over a wide range of output capacitance and ESR values. The LTC7800 features a precision 0.8V reference and power good output indicator . A wide 4V to 60V input supply range encompasses a wide range of intermediate bus voltages and battery chemistries. The output voltage of the LTC7800 can be programmed between 0.8V to 24V. The TRACK/SS pin ramps the output voltages during start-up. Current foldback limits MOSFET heat dissipation during short-circuit conditions. The PLLIN/MODE pin se- lects among Burst Mode operation, pulse-skipping mode, or continuous conduction mode at light loads.

APPLICATIONS

n Wide VIN Range: 4V to 60V (65V Abs Max) n Low Operating IQ: 50μA n Wide Output Voltage Range: 0.8V ≤ VOUT ≤ 24V n RSENSE or DCR Current Sensing n Phase-Lockable Frequency (320kHz to 2.25MHz) n Programmable Fixed Frequency (320kHz to 2.25MHz) n Selectable Continuous, Pulse-Skipping or Low Ripple Burst Mode® Operation at Light Load n Selectable Current Limit n Very Low Dropout Operation: 98% Duty Cycle n Adjustable Output Voltage Soft-Start or T racking n Power Good Output Voltage Monitor n Output Overvoltage Protection n Low Shutdown IQ: < 14μA n Internal LDO Powers Gate Drive from VIN or EXTVCC n No Current Foldback During Start-Up n Small 20-Pin 3mm × 4mm QFN Package n Automotive Always-On Systems n Battery Powered Digital Devices n Distributed DC Power Systems L, L T , L TC, L TM, OPTI-LOOP , Burst Mode, Linear Technology and the Linear logo are registered trademarks of Analog Devices, Inc. All other trademarks are the property of their respective owners. Patents, including 5481178, 5705919, 6611131, 6498466, 6580258, 7230497. 0.33µH 4m/uni03A9 357k 115k 2.2µF 0.1µF 33µF 100k 100k 2.49k 820pF 80pF 0.1µF VIN PGOOD EXTVCC INTVCC PGND FREQ ITH TRACK/SS SGND TG BOOST SW BG SENSE+ SENSE– VFB L TC7800 C 2.2µF C 56µF INTVCC V 28V IN 3.3V V OUT 10A 4V TO C OUT High Efficiency 3.3V 2.1MHz Step-Down Regulator Efficiency and Power Loss vs Output Current V IN = 12V V OUT = 3.3V OUTPUT CURRENT (A) 0.0001 0.001 0.01 0.1 100 0.0001 0.001 0.01 0.1 EFFICIENCY (%) POWER LOSS (W)

3874 TA01b

7800f For more information www.linear.com/LTC7800 PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS 3V to 71V 65V 3V to 6V Note 8) 3V to 8V Ma ximum Current Sourced into Pin from S 0μA 3V to 6V 3V to 6V Operating Junction Temperature Range (Notes 2, 3) LTC78 125°C LTC78 C to 150°C Maximum Junction Temperature (Notes 2, 3) LTC78 C LTC78 0°C C to 150°C (Note 1) 20 19 18 17 7 8 TOP VIEW SGND UDC PACKAGE 20-LEAD (3mm × 4mm) PLASTIC QFN 9 10 16PLLIN/MODE SGND SGND RUN SENSE– SENSE+ PGND EXTV CC INTVCC BG BOOST SW FREQ TRACK/SS I LIM VIN VFB ITH PGOOD TG TJMAX = 150°C, θJA = 52°C/W EXPOSED PAD (PIN 21) IS SGND, MUST BE SOLDERED TO PCB LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC7800EUDC#PBF LTC7800EUDC#TRPBF LHBS 20-Lead (3mm × 4mm) Plastic QFN –40°C to 125°C LTC7800IUDC#PBF LTC7800IUDC#TRPBF LHBS 20-Lead (3mm × 4mm) Plastic QFN –40°C to 125°C LTC7800HUDC#PBF LTC7800HUDC#TRPBF LHBS 20-Lead (3mm × 4mm) Plastic QFN –40°C to 150°C Consult L TC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container . Consult L TC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear .com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear .com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. ORDER INFORMATIONhttp://www.linear .com/product/LTC7800#orderinfo

7800fFor more information www.linear.com/LTC7800 The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2), VIN = 12V, VRUN = 5V, EXTVCC = 0V unless otherwise noted.

ELECTRICAL CHARACTERISTICS

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS VIN Input Supply Operating Voltage Range 4 60 V VFB Regulated Feedback Voltage (Note 4); ITH Voltage = 1.2V –40°C to 85°C LTC7800E, LTC7800I LTC7800H l l 0.792 0.788 0.786 0.800 0.800 0.800 0.808 0.812 0.812 V V V I FB Feedback Current (Note 4) ±5 ±50 nA VREFLNREG Reference Voltage Line Regulation (Note 4); V IN = 4.5V to 60V 0.002 0.02 %/V VLOADREG Output Voltage Load Regulation (Note 4) Measured in Servo Loop; TH Voltage = 1.2V to 0.7V l 0.01 0.1 % (Note 4) Measured in Servo Loop; TH Voltage = 1.2V to 2V l –0.01 –0.1 % gm T ransconductance Amplifier gm (Note 4); ITH = 1.2V; Sink/Source 5µA 2 mmho IQ Input DC Supply Current (Note 5) Pulse Skip or Forced Continuous Mode V FB = 0.83V (No Load) 2 mA Sleep Mode VFB = 0.83V (No Load) 50 75 µA Shutdown RUN = 0V 14 25 µA UVLO Undervoltage Lockout INTVCC Ramping Up INTVCC Ramping Down l l 3.6 3.92 3.80 4.2 4.0 V V V OVL Feedback Overvoltage Protection Measured at V FB Relative to Regulated VFB 7 10 13 % ISENSE+ SENSE+ Pin Current ±1 µA ISENSE– SENSE– Pins Current VSENSE– < INTVCC – 0.5V VSENSE– > INTVCC + 0.5V 700 µA µA DFMAX Maximum Duty Factor In Dropout; VFREQ = 0V 97 98 % ITRACK/SS Soft-Start Charge Current VTRACK/SS = 0V 7 10 14 µA VRUN On RUN Pin On Threshold VRUN Rising l 1.15 1.21 1.27 V VRUN Hyst RUN Pin Hysteresis 50 mV VSENSE(MAX) Maximum Current Sense Threshold V FB = 0.7V, VSENSE– = 3.3V, ILIM = 0V VFB = 0.7V, VSENSE– = 3.3V, ILIM = INTVCC VFB = 0.7V, VSENSE– = 3.3V, ILIM = FLOAT l l l mV mV mV Gate Driver TG Pull-Up On-Resistance Pull-Down On-Resistance 2.5 1.5 Ω Ω BG Pull-Up On-Resistance Pull-Down On-Resistance 2.4 1.1 Ω Ω TG tr TG tf TG T ransition Time: Rise Time Fall Time (Note 6) C LOAD = 3300pF CLOAD = 3300pF ns ns BG t r BG tf BG T ransition Time: Rise Time Fall Time (Note 6) C LOAD = 3300pF CLOAD = 3300pF ns ns TG/BG t 1D Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time C LOAD = 3300pF 20 ns BG/TG t1D Bottom Gate Off to Top Gate On Delay Top Switch-On Delay Time C LOAD = 3300pF 20 ns

7800f For more information www.linear.com/LTC7800 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2), VIN = 12V, VRUN = 5V, EXTVCC = 0V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS tON(MIN) Minimum TG On-Time (Note 7) 45 ns INTVCC Linear Regulator VINTVCCVIN Internal VCC Voltage 6V < VIN < 60V, VEXTVCC = 0V 4.85 5.1 5.35 V VLDOVIN INTVCC Load Regulation ICC = 0mA to 50mA, VEXTVCC = 0V 0.7 1.1 % VINTVCCEXT Internal VCC Voltage 6V < VEXTVCC < 13V 4.85 5.1 5.35 V VLDOEXT INTVCC Load Regulation ICC = 0mA to 50mA, VEXTVCC = 8.5V 0.6 1.1 % VEXTVCC EXTVCC Switchover Voltage ICC = 0mA to 50mA, EXTVCC Ramping Positive 4.5 4.7 4.9 V VLDOHYS EXTVCC Hysteresis 250 mV Oscillator and Phase-Locked Loop f25kΩ Programmable Frequency RFREQ = 25k; PLLIN/MODE = DC Voltage 0.27 0.32 0.36 MHz f65kΩ Programmable Frequency RFREQ = 65k; PLLIN/MODE = DC Voltage

1.18 MHz

f100kΩ Programmable Frequency RFREQ =100k; PLLIN/MODE = DC Voltage l 1.75 2.1 2.4 MHz fLOW Low Fixed Frequency VFREQ = 0V; PLLIN/MODE = DC Voltage 0.79 0.94 1.08 MHz fHIGH High Fixed Frequency VFREQ = INTVCC; PLLIN/MODE = DC Voltage 1.2 1.44 1.7 MHz fSYNC Synchronizable Frequency PLLIN/MODE = External Clock l 0.32 2.25 MHz PLLIN VIH PLLIN/MODE Input High Level PLLIN/MODE = External Clock l 2.5 V PLLIN VIL PLLIN/MODE Input Low Level PLLIN/MODE = External Clock l 0.5 V PGOOD1 Output V PGL PGOOD Voltage Low IPGOOD = 2mA 0.2 0.4 V IPGOOD PGOOD Leakage Current VPGOOD = 5V ±1 µA VPG PGOOD T rip Level VFB with Respect to Set Regulated Voltage VFB Ramping Negative –13 –10 –7 % Hysteresis 2.5 % VFB Ramping Positive 7 10 13 % Hysteresis 2.5 % tPG Delay for Reporting a Fault 25 µs Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC7800 is tested under pulsed load conditions such that T J ≈ TA. The LTC7800E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC7800I is guaranteed over the –40°C to 125°C operating junction temperature range, the LTC7800H is guaranteed over the –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes; operating lifetime is derated for junction temperatures greater than 125°C. Note that the maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. Note 3: The junction temperature (T J, in °C) is calculated from the ambient temperature (TA, in °C) and power dissipation (PD, in Watts) according to the formula: TJ = TA + (PD • θJA), where θJA is 52°C/ W.

7800fFor more information www.linear.com/LTC7800 TYPICAL PERFORMANCE CHARACTERISTICS FCM LOSS BURST EFFICIENCY V IN = 12V V OUT = 3.3V PULSE-SKIPPING LOSS BURST LOSS FCM EFFICIENCY PULSE–SKIPPING EFFICIENCY 0.0001 0.001 0.01 0.1 100 0.0001 0.001 0.01 0.1 POWER LOSS (W) OUTPUT CURRENT (A) FIGURE 11 CIRCUIT EFFICIENCY (%)

7800 G01

V OUT = 5V V OUT = 3.3V Burst Mode OPERATION V IN = 12V 0.0001 0.001 0.01 0.1 100 OUTPUT CURRENT (A)

7800 G02

FIGURES 11, 12 CIRCUITS EFFICIENCY (%) I LOAD = 5A V OUT = 5V V OUT = 3.3V 100

7800 G03INPUT VOL TAGE (V)

EFFICIENCY (%) FIGURES 11, 12 CIRCUITS Efficiency and Power Loss vs Output Current Efficiency vs Output Current Efficiency vs Input Voltage Note 4: The LTC7800 is tested in a feedback loop that servos VITH to a specified voltage and measures the resultant VFB. The specification at 85°C is not tested in production and is assured by design, characterization and correlation to production testing at other temperatures (125°C for the LTC7800E/LTC7800I and 150°C for the LTC7800H.) Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 6: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels Note 7: The minimum on-time condition is specified for an inductor peak-to-peak ripple current ≥ 40% of I MAX (See Minimum On-Time Considerations in the Applications Information section). Note 8: Do not apply a voltage or current source to these pins. They must be connected to capacitive loads only, otherwise permanent damage may occur . Load Step Burst Mode Operation Load Step Forced Continuous Mode Load Step Pulse-Skipping Mode LOAD STEP = 500mA TO 5A VIN = 12V VOUT = 3.3V FIGURE 11 CIRCUIT 20µs/DIV

7800 G04

LOAD STEP = 500mA TO 5A V IN = 12V VOUT = 3.3V FIGURE 11 CIRCUIT 20µs/DIV

7800 G05

LOAD STEP = 500mA TO 5A V IN = 12V VOUT = 3.3V FIGURE 11 CIRCUIT 50µs/DIV

7800 G06

7800f For more information www.linear.com/LTC7800 TYPICAL PERFORMANCE CHARACTERISTICS 100 150 200 250 300 INPUT VOL TAGE (V) SUPPL Y CURRENT (µA)

7800 G10

300µA LOAD FIGURE 13 CIRCUIT TEMPERATURE (°C) –75 4.0 EXTVCC AND INTVCC VOLTAGE (V) 4.2 4.6 4.8 5.0 6.0 5.4 –25 25 50 75 100

7800 G11

4.4 5.6 5.8 5.2 –50 0 125 150 INTVCC EXTVCC RISING EXTVCC FALLING VITH (V) CURRENT SENSE THESHOLD (mV) 0.6 1.0

7800 G13

–20 –40 Burst Mode OPERATION PULSE-SKIPPING MODE 5% DUTY CYCLE FORCED CONTINUOUS MODE ILIM = FLOAT ILIM = INTVCC ILIM = GND VSENSE COMMON MODE VOLTAGE (V) SENSE– CURRENT (µA)

7800 G14

–100 DUTY CYCLE (%) MAXIMUM CURRENT SENSE VOL TAGE (mV)

7800 G15

ILIM = FLOAT ILIM = INTVCC ILIM = GND Total Input Supply Current vs Input Voltage EXTVCC Switchover and INTVCC Voltages vs Temperature INTVCC Line Regulation Maximum Current Sense Voltage vs I TH Voltage SENSE– Pin Input Bias Current Maximum Current Sense Threshold vs Duty Cycle VIN = 12V VOUT = 3.3V ILOAD = 50mA 1µs/DIV

7800 G07

Inductor Current at Light Load Soft Start-Up Shutdown Current vs Input Voltage FIGURES 11, 12 CIRCUITS 2ms/DIV

7800 G08

V OUT = 5V 1V/DIV V OUT = 3.3V 1V/DIV INPUT VOL TAGE (V) 3.0 INTVCC VOL TAGE (V) 4.0 4.5 5.5 20 30 35 40 45 50 55 60 6515 10 0 525

7800 G12

3.5 5.0 ILOAD = 10mA INPUT VOL TAGE (V) SHUTDOWN CURRENT (µA) 20 30 35 40 45 50 55 60 6515 10 5 25

7800 G09

7800fFor more information www.linear.com/LTC7800 TYPICAL PERFORMANCE CHARACTERISTICS FEEDBACK VOL TAGE (mV) MAXIMUM CURRENT SENSE VOL TAGE (mV) 800

7800 G16

ILIM = FLOAT ILIM = INTVCC ILIM = GND TEMPERATURE (°C) –75 QUIESCENT CURRENT (µA) 50 25 0

7800 G17

–50 –25 75 100 125 150 VIN = 12V Foldback Current Limit Quiescent Current vs Temperature INTVCC vs Load Current TEMPERATURE (°C) –75 REGULATED FEEDBACK VOLTAGE (mV) 806 0 25 50

7800 G21

–50 –25 75 794 792 808 804 802 798 100 125 150 TRACK/SS Pull-Up Current vs Temperature Shutdown (RUN) Threshold vs Temperature Regulated Feedback Voltage vs Temperature LOAD CURRENT (mA) INTVCC VOLTAGE (V) 5.25

7800 G18

4.50 20 60 4.00 5.50 5.00 4.75 4.25 80 100 EXTVCC = 5V EXTVCC = 8.5V EXTVCC = 0V VIN = 12V TEMPERATURE (°C) –75 TRACK/SS CURRENT (µA) 11.5 50 25

7800 G19

10.0 –25 0–50 75 12.0 11.0 10.5 9.5 9.0 8.5 8.0 125 100150 TEMPERATURE (°C) –75 RUN PIN VOLTAGE (V) 50 0 25

7800 G20

1.20 –25 –50 75 1.30 1.25 1.15 1.10 125 100150 RUN RISING RUN FALLING FREQUENCY (kHz) DFMAX (%)

7800 G23

96.0 96.5 97.0 97.5 98.0 98.5 99.0 99.5 100.0 TEMPERATURE (°C) FREQUENCY (MHz)

7800 G24

FREQ = 100k FREQ = INTVCC FREQ = GND –75 –50 –25 100 125 150 0.85 1.05 1.25 1.45 1.65 1.85 2.05 2.25 SENSE– Pin Input Bias Current vs Temperature Maximum Duty Factor vs Frequency Oscillator Frequency vs Temperature TEMPERATURE (°C) –75 SENSE– CURRENT (µA) 50 25 0

7800 G22

–25 –50 75 800 700 600 500 200 100 –100 125 100150 VOUT > INTVCC + 0.5V VOUT < INTVCC – 0.5V

7800f For more information www.linear.com/LTC7800 PIN FUNCTIONS PLLIN/MODE (Pin 1): External Synchronization Input to Phase Detector and Forced Continuous Mode Input. When an external clock is applied to this pin, the phase-locked loop will force the rising TG signal to be synchronized with the rising edge of the external clock, and the regulator operates in forced continuous mode. When not synchro- nizing to an external clock, this input determines how the LTC7800 operates at light loads. Pulling this pin to ground selects Burst Mode operation. An internal 100k resistor to ground also invokes Burst Mode operation when the pin is floated. Tying this pin to INTV CC forces continuous inductor current operation. Tying this pin to a voltage greater than 1.2V and less than INTV CC –1.3V selects pulse-skipping operation. This can be done by connecting a 100k resistor from this pin to INTV CC. SGND (Pins 2, 3, Exposed Pad Pin 21): Small-signal ground, must be routed separately from high current grounds to the common (–) terminals of the C IN capacitor . Pins 2, 3, Exposed Pad Pin 21, must both be electrically connected to small signal ground for proper operation. The exposed pad must be soldered to PCB ground for rated thermal performance. RUN (Pin 4): Digital Run Control Input. Forcing this pin below 1.16V shuts down the controller . Forcing this pin below 0.7V shuts down the entire LTC7800, reducing quiescent current to approximately 14µA. SENSE – (Pin 5): The (–) Input to the Differential Current Comparator . When greater than INTVCC – 0.5V, the SENSE– pin supplies power to the current comparator . SENSE + (Pin 6): The (+) input to the differential current comparator is normally connected to DCR sensing net - work or current sensing resistor . The ITH pin voltage and controlled offsets between the SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip threshold. VFB (Pin 7): Receives the remotely sensed feedback volt- age from an external resistive divider across the output. ITH (Pin 8): Error Amplifier Outputs and Switching Regula- tor Compensation Point. The current comparator trip point increases with this control voltage. INPUT VOL TAGE (V) OSCILLATOR FREQUENCY (kHz)

7800 G25

FREQ = GND 935 936 937 938 939 941 942 943 944 945 TEMPERATURE (°C) –75 INTVCC VOLTAGE (V) 3.7 3.8 3.9 4.1 –25 75 100

7800 G26

3.6 4.2 4.0 –50 0 25 50 125 150 FALLING RISING TEMPERATURE (°C) SHUTDOWN CURRENT (µA) 25 50 75 100 125 150–25–75 –50 0

7800 G27

VIN = 12V Oscillator Frequency vs Input Voltage Undervoltage Lockout Threshold vs Temperature Shutdown Current vs Temperature TYPICAL PERFORMANCE CHARACTERISTICS

7800fFor more information www.linear.com/LTC7800 PIN FUNCTIONS PGOOD (Pin 9): Open-Drain Logic Output. PGOOD is pulled to ground when the voltage on the V FB pin is not within 10% of its set point. TG (Pin 10): High Current Gate Drives for Top N-channel MOSFET . This is the output of floating driver with a volt- age swing equal to INTVCC superimposed on the switch node voltage SW . SW (Pin 11): Switch Node Connection to Inductor . BOOST (Pin 12): Bootstrapped Supply to the Topside Floating Driver . A capacitor is connected between the BOOST and SW pin and a Schottky diode is tied between the BOOST and INTV CC pins. Voltage swing at the BOOST pin is from INTVCC to (VIN + INTVCC). BG (Pin 13): High Current Gate Drive for Bottom (Syn - chronous) N-channel MOSFET . Voltage swing at this pin is from ground to INTV CC. INTVCC (Pin 14): Output of the Internal Linear Low Dropout Regulator . The driver and control circuits are powered from this voltage source. Must be decoupled to PGND with a minimum of 2.2µF ceramic or other low ESR capacitor . Do not use the INTV CC pin for any other purpose. EXTVCC (Pin 15): External Power Input to an Internal LDO Connected to INTVCC. This LDO supplies INTV CC power , bypassing the internal LDO powered from V IN whenever EXTVCC is higher than 4.7V. See EXTVCC Connection in the Applications Information section. Do not float or exceed 14V on this pin. PGND (Pin 16): Driver Power Ground. Connects to the source of bottom (synchronous) N-channel MOSFET and the (–) terminal of C IN. VIN (Pin 17): Main Supply Pin. A bypass capacitor should be tied between this pin and the SGND pins. ILIM (Pin 18): Current Comparator Sense Voltage Range Inputs. Tying this pin to SGND, FLOAT or INTVCC sets the maximum current sense threshold to one of three different levels for the comparator . TRACK/SS (Pin 19): External T racking and Soft-Start Input. The LTC7800 regulates the VFB voltage to the smaller of 0.8V or the voltage on the TRACK/SS pin. An internal 10μA pull-up current source is connected to this pin. A capacitor to ground at this pin sets the ramp time to final regulated output voltage. Alternatively, a resistor divider on another voltage supply connected to this pin allows the LTC7800 output to track another supply during start-up. FREQ (Pin 20): The frequency control pin for the internal VCO. Connecting the pin to GND forces the VCO to a fixed low frequency of 0.94MHz. Connecting the pin to INTV CC forces the VCO to a fixed high frequency of 1.44MHz. Other frequencies between 320kHz and 2.25MHz can be programmed by using a resistor between FREQ and GND. An internal 20µA pull-up current develops the voltage to be used by the VCO to control the frequency.

7800f For more information www.linear.com/LTC7800 FUNCTIONAL DIAGRAM SW TOP BOOST TG CB CIND DB PGND BOT BG INTVCC INTVCC VIN COUT VOUT 7800 FD RSENSE DROP OUT DET BOT TOP ONS R Q Q SHDN SLEEP0.425V ICMP 2.7V 0.65V IR 2mV SLOPE COMP SENSE+ SENSE– PGOOD VFB 0.88V 0.72V L FREQ +– + – SWITCH LOGIC VFB RA CC RCCC2 RB 0.80V TRACK/SS 0.88V 7µA 11V RUN ITH TRACK/SS CSS 10µA SHDN CURRENT LIMIT FOLDBACK SHDN RST 2(VFB) PLLIN/MODE 20µA VCO LDO EN INTVCC 5.1V SYNC DET 100k CLK2 CLK1 ILIM VIN EXTVCC LDO PFD EN 4.7V 5.1V SGND EA OV

7800fFor more information www.linear.com/LTC7800 OPERATION Main Control Loop The LTC7800 uses a constant frequency, current mode step-down architecture. During normal operation, the external top MOSFET is turned on when the clock for that channel sets the RS latch, and is turned off when the main current comparator , ICMP , resets the RS latch. The peak inductor current at which ICMP trips and resets the latch is controlled by the voltage on the ITH pin, which is the output of the error amplifier , EA. The error amplifier compares the output voltage feedback signal at the V FB pin (which is generated with an external resistor divider connected across the output voltage, V OUT, to ground) to the internal 0.800V reference voltage. When the load current increases, it causes a slight decrease in V FB rela- tive to the reference, which causes the EA to increase the ITH voltage until the average inductor current matches the new load current. After the top MOSFET is turned off each cycle, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current comparator IR, or the beginning of the next clock cycle. INTV CC/EXTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTV CC pin. When the EXTVCC pin is tied to a voltage less than 4.7V, the VIN LDO (low dropout linear regulator) supplies 5.1V from VIN to INTVCC. If EXTVCC is taken above 4.7V, the VIN LDO is turned off and an EXTVCC LDO is turned on. Once enabled, the EXTVCC LDO supplies 5.1V from EXTVCC to INTVCC. Using the EXTVCC pin allows the INTV CC power to be derived from a high efficiency external source such as one of the LTC7800 switching regulator outputs. The top MOSFET driver is biased from the floating bootstrap capacitor , C B, which normally recharges during each cycle through an external diode when the top MOSFET turns off. If the input voltage, V IN, decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector detects this and forces the top MOSFET off for a short time every tenth cycle to allow C B to recharge resulting in about 98% duty cycle at 1MHz operation. Shutdown and Start-Up (RUN, TRACK/SS Pins) The LTC7800 can be shut down using the RUN pin. Pulling this pin below 1.16V shuts down the main control loop. Pulling the RUN pin below 0.7V disables the controller and most internal circuits, including the INTV CC LDOs. In this state, the LTC7800 draws only 14μA of quiescent current. Releasing the RUN pin allows a small internal current to pull up the pin to enable the controller . The RUN pin has a 7μA pull-up which is designed to be large enough so that the RUN pin can be safely floated (to always enable the controller) without worry of condensation or other small board leakage pulling the pin down. This is ideal for always-on applications where the controller is enabled continuously and never shut down. The RUN pin may be externally pulled up or driven directly by logic. When driving the RUN pin with a low impedance source, do not exceed the absolute maximum rating of 8V. The RUN pin has an internal 11V voltage clamp that allows the RUN pin to be connected through a resistor to a higher voltage (for example, V IN), so long as the maximum current into the RUN pin does not exceed 100μA. The RUN pin can also be implemented as a UVLO by connecting it to the output of an external resistor divider network off V IN (see Applications Information section). The start-up of the controller’s output voltage V OUT is controlled by the voltage on the TRACK/SS pin. When the voltage on the TRACK/SS pin is less than the 0.8V internal reference, the LTC7800 regulates the V FB voltage to the TRACK/SS pin voltage instead of the 0.8V reference. This allows the TRACK/SS pin to be used to program a soft-start by connecting an external capacitor from the TRACK/SS pin to SGND. An internal 10μA pull-up current charges this capacitor creating a voltage ramp on the TRACK/ SS pin. As the TRACK/SS voltage rises linearly from 0V to 0.8V (and beyond up to 5V), the output voltage V OUT rises smoothly from zero to its final value. Alternatively the TRACK/SS pin can be used to cause the start-up of V OUT to track that of another supply. Typically, this requires connecting to the TRACK/SS pin an external resistor divider from the other supply to ground (see Applications Information section).

7800f For more information www.linear.com/LTC7800 Light Load Current Operation (Burst Mode Operation, Pulse-Skipping or Forced Continuous Mode) (PLLIN/MODE Pin) The LTC7800 can be enabled to enter high efficiency Burst Mode operation, constant frequency pulse-skipping mode, or forced continuous conduction mode at low load cur - rents. To select Burst Mode operation, tie the PLLIN/MODE pin to SGND. To select for ced continuous operation, tie the PLLIN/MODE pin to INTVCC. To select pulse-skipping mode, tie the PLLIN/MODE pin to a DC voltage greater than 1.2V and less than INTV CC – 1.3V. When the controller is enabled for Burst Mode opera- tion, the minimum peak current in the inductor is set to approximately 25% of the maximum sense voltage even though the voltage on the ITH pin indicates a lower value. If the average inductor current is higher than the load cur- rent, the error amplifier , EA, will decrease the voltage on the ITH pin. When the ITH voltage drops below 0.425V, the internal sleep signal goes high (enabling sleep mode) and both external MOSFET s are turned off. The ITH pin is then disconnected from the output of the EA and parked at 0.450V. In sleep mode, much of the internal circuitry is turned off, reducing the quiescent current that the LTC7800 draws to only 50μA. In sleep mode, the load current is supplied by the output capacitor . As the output voltage decreases, the EA’s output begins to rise. When the output voltage drops enough, the ITH pin is reconnected to the output of the EA, the sleep signal goes low, and the controller resumes normal operation by turning on the top external MOSFET on the next cycle of the internal oscillator . When the controller is enabled for Burst Mode operation, the inductor current is not allowed to reverse. The reverse current comparator , IR, turns off the bottom external MOSFET just before the inductor current reaches zero, preventing it from reversing and going negative. Thus, the controller operates in discontinuous operation. In forced continuous operation or clocked by an external clock source to use the phase-locked loop (see Frequency Selection and Phase-Locked Loop section), the inductor OPERATION current is allowed to reverse at light loads or under large transient conditions. The peak inductor current is deter - mined by the voltage on the ITH pin, just as in normal operation. In this mode, the efficiency at light loads is lower than in Burst Mode operation. However, continuous operation has the advantage of lower output voltage ripple and less inter ference to audio circuitry. In forced continu- ous mode, the output ripple is independent of load current. When the PLLIN/MODE pin is connected for pulse-skipping mode, the LTC7800 operates in PWM pulse-skipping mode at light loads. In this mode, constant frequency operation is maintained down to approximately 1% of designed maximum output current. At very light loads, the current comparator , ICMP , may remain tripped for several cycles and force the external top MOSFET to stay off for the same number of cycles (i.e., skipping pulses). The inductor current is not allowed to reverse (discontinuous operation). This mode, like forced continuous operation, exhibits low output ripple as well as low audio noise and reduced RF interference as compared to Burst Mode operation. It provides higher low current efficiency than forced continuous mode, but not nearly as high as Burst Mode operation. Frequency Selection and Phase-Locked Loop (FREQ and PLLIN/MODE Pins) The selection of switching frequency is a trade-off between efficiency and component size. Low frequency opera - tion increases efficiency by reducing MOSFET switching losses, but requires larger inductance and/or capacitance to maintain low output ripple voltage. The switching frequency of the LTC7800 can be selected using the FREQ pin. If the PLLIN/MODE pin is not being driven by an external clock source, the FREQ pin can be tied to SGND, tied to INTV CC or programmed through an external resistor . Tying FREQ to SGND selects 0.94MHz while tying FREQ to INTV CC selects 1.44MHz. Placing a resistor between FREQ and SGND allows the frequency to be programmed between 320kHz and 2.25MHz, as shown in Figure 8.

7800fFor more information www.linear.com/LTC7800 A phase-locked loop (PLL) is available on the LTC7800 to synchronize the internal oscillator to an external clock source that is connected to the PLLIN/MODE pin. The LTC7800’s phase detector adjusts the voltage (through an internal lowpass filter) of the VCO input to align the turn-on of the controller’s external top MOSFET to the rising edge of the synchronizing signal. The VCO input voltage is prebiased to the operating fre - quency set by the FREQ pin before the external clock is applied. If prebiased near the external clock frequency , the PLL loop only needs to make slight changes to the VCO input in order to synchronize the rising edge of the external clock’s to the rising edge of TG. The ability to prebias the loop filter allows the PLL to lock-in rapidly without deviating far from the desired frequency. The typical capture range of the phase-locked loop is from approximately 300kHz to 2.3MHz, with a guarantee to be between 320kHz and 2.25MHz. In other words, the LTC7800’s PLL is guaranteed to lock to an external clock source whose frequency is between 320kHz and 2.25MHz. The typical input clock thresholds on the PLLIN/MODE pin are 1.6V (rising) and 1.1V (falling). The LTC7800 is guaranteed to synchronize to an external clock that swings up to at least 2.5V and down to 0.5V or less. Output Overvoltage Protection An overvoltage comparator guards against transient over- shoots as well as other more serious conditions that may overvoltage the output. When the V FB pin rises by more than 10% above its regulation point of 0.800V, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. Power Good Pin The PGOOD pin is connected to an open drain of an internal N-channel MOSFET . The MOSFET turns on and pulls the PGOOD pin low when the V FB pin voltage is not within ±10% of the 0.8V reference voltage. The PGOOD pin is also pulled low when the RUN pin is low (shut down). When the V FB pin voltage is within the ±10% requirement, the MOSFET is turned off and the pin is allowed to be pulled up by an external resistor to a source no greater than 6V. Foldback Current When the output voltage falls to less than 70% of its nominal level, foldback current limiting is activated, pro- gressively lowering the peak current limit in proportion to the severity of the overcurrent or short-cir cuit condition. Foldback current limiting is disabled during the soft-start interval (as long as the V FB voltage is keeping up with the TRACK/SS voltage). OPERATION

input and output capacitors are selected. floated, the maximum current limit threshold is 75mV. margin for tolerances and transients). smaller current to the higher current. coupling into sensitive small-signal nodes. Figure 1. Sense Lines Placement with Inductor or Sense Resistor

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Figure 2. Current Sensing Methods

7800fFor more information www.linear.com/LTC7800 Low Value Resistor Current Sensing A typical sensing circuit using a discrete resistor is shown in Figure 2a. R SENSE is chosen based on the required output current. The current comparator has a maximum threshold VSENSE(MAX) determined by the I LIM setting. The current comparator threshold voltage sets the peak of the induc- tor current, yielding a maximum average output current, I MAX, equal to the peak value less half the peak-to-peak ripple current, ΔIL. To calculate the sense resistor value, use the equation: RSENSE = VSENSE(MAX) IMAX + ΔIL To ensure that the application will deliver full load current over the full operating temperature range, choose the minimum value for the Maximum Current Sense Threshold SENSE(MAX)) in the Electrical Characteristics table (30mV, 50mV or 75mV, depending on the state of the ILIM pin). When using the controller in very low dropout conditions, the maximum output current level will be reduced due to the internal compensation required to meet stability cri - terion for buck regulators operating at greater than 50% duty factor . A curve is provided in the Typical Performance Characteristics section to estimate this reduction in peak inductor current depending upon the operating duty factor . Inductor DCR Sensing For applications requiring the highest possible efficiency at high load currents, the LTC7800 is capable of sensing the voltage drop across the inductor DCR, as shown in Figure 2b. The DCR of the inductor represents the small amount of DC resistance of the copper wire, which can be l ess than 1mΩ for today’s low value, high current inductors. In a high current application requiring such an inductor , power loss through a sense resistor would cost several points of efficiency compared to inductor DCR sensing. If the external (R1||R2) • C1 time constant is chosen to be exactly equal to the L/DCR time constant, the voltage drop across the external capacitor is equal to the drop across the inductor DCR multiplied by R2/(R1 + R2). R2 scales the voltage across the sense terminals for applications where the DCR is greater than the target sense resistor value. To properly dimension the external filter components, the DCR of the inductor must be known. It can be measured using a good RLC meter , but the DCR tolerance is not always the same and varies with temperature; consult the manufacturers’ data sheets for detailed information. Using the inductor ripple current value from the Inductor Value Calculation section, the target sense resistor value is: RSENSE(EQUIV) = VSENSE(MAX) IMAX + ΔIL To ensure that the application will deliver full load current over the full operating temperature range, choose the minimum value for the Maximum Current Sense Threshold SENSE(MAX)) in the Electrical Characteristics table (30mV, 50mV or 75mV, depending on the state of the ILIM pin). Next, determine the DCR of the inductor . When provided, use the manufacturer’s maximum value, usually given at 20°C. Increase this value to account for the temperature coefficient of copper resistance, which is approximately 0.4%/°C. A conservative value for T L(MAX) is 100°C. To scale the maximum inductor DCR to the desired resistor value (R D), use the divider ratio: RD = RSENSE(EQUIV) DCRMAX atTL(MAX) C1 is usually selected to be in the range of 0.1μF to 0.47μF. This forces R1 || R2 to around 2k, reducing error that might have been caused by the SENSE + pin’s ±1μA current. APPLICATIONS INFORMATION

7800f For more information www.linear.com/LTC7800 The equivalent resistance R1 || R2 is scaled to the tem - perature inductance and maximum DCR: R1|| R2= L DCR at 20 °C( ) • C1 The resistor values are: R1= R1|| R2 RD ; R2 = R1• RD 1– RD The maximum power loss in R1 is related to duty cycle, and will occur in continuous mode at the maximum input voltage: PLOSS R1= VIN(MAX) – VOUT( ) • VOUT Ensure that R1 has a power rating higher than this value. If high efficiency is necessary at light loads, consider this power loss when deciding whether to use DCR sensing or sense resistors. Light load power loss can be modestly higher with a DCR network than with a sense resistor , due to the extra switching losses incurred through R1. However , DCR sensing eliminates a sense resistor , reduces conduc- tion losses and provides higher efficiency at heavy loads. Peak efficiency is about the same with either method. Inductor Value Calculation The operating frequency and inductor selection are inter- related n that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET switching and gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current, ΔI L, decreases with higher induc- tance or higher frequency and increases with higher VIN: ΔIL = 1 f( ) L( ) VOUT 1– VOUT VIN Accepting larger values of ΔIL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is ΔI L = 0.3(I MAX). The maximum ΔIL occurs at the maximum input voltage. The inductor value also has secondary effects. The tran- sition to Burst Mode operation begins when the average inductor current required results in a peak current below 25% of the current limit determined by R SENSE. Lower inductor values (higher ΔI L) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite or molypermalloy cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance value selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred for high switching frequencies, so design goals can con- centrate on copper loss and preventing saturation. Ferrite core material saturates hard, which means that induc - tance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! APPLICATIONS INFORMATION

7800fFor more information www.linear.com/LTC7800 Power MOSFET and Schottky Diode (Optional) Selection T wo external power MOSFETs must be selected for the LTC7800 controller: one N-channel MOSFET for the top (main) switch, and one N-channel MOSFET for the bottom (synchronous) switch. The peak-to-peak drive levels are set by the INTV CC voltage. This voltage is typically 5.1V during start-up (see EXTVCC Pin Connection). Consequently, logic-level threshold MOSFETs must be used in most applications. Pay close attention to the BV DSS specification for the MOSFETs as well. Selection criteria for the power MOSFETs include the on- resistance, R DS(ON), Miller capacitance, C MILLER, input voltage and maximum output current. Miller capacitance, C MILLER, can be approximated from the gate charge curve usually provided on the MOSFET manufacturers’ datasheet. C MILLER is equal to the increase in gate charge along the horizontal axis while the curve is approximately flat divided by the specified change in V DS. This result is then multiplied by the ratio of the application applied VDS to the gate charge curve specified V DS. When the IC is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty Cycle = VOUT VIN Synchronous Switch Duty Cycle = VIN − VOUT VIN The MOSFET power dissipations at maximum output current are given by: PMAIN = VOUT VIN IMAX( ) 2 1+ δ( ) RDS(ON) + VIN( )

2 IMAX

   RDR( ) CMILLER( ) • VINTVCC – VPLATEAU + 1 VPLATEAU  f( ) PSYNC = VIN – VOUT VIN IMAX( ) 2 1+ δ( ) RDS(ON) APPLICATIONS INFORMATION where δ is the temperature dependency of R DS(ON) and RDR (approximately 2Ω) is the effective driver resistance at the MOSFET’s Miller threshold voltage. V THMIN is the typical MOSFET minimum threshold voltage. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For V IN < 20V the high current efficiency generally improves with larger MOSFETs, while for V IN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. The term (1+ δ) is generally given for a MOSFET in the form of a normalized R DS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. A Schottky diode can be inserted in parallel with the bot- tom MOSFET to conduct during the dead-time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the dead-time and requiring a reverse recovery period that could cost as much as 3% in efficiency at high V IN. A 1A to 3A Schottky is generally a good compromise for both regions of operation due to the relatively small average current. Larger diodes result in additional transition losses due to their larger junction capacitance. C IN and COUT Selection The selection of CIN is usually based off the worst-case RMS input current. The highest (VOUT)(IOUT) product needs to be used in the formula shown in Equation 1 to determine the maximum RMS capacitor current requirement. In continuous mode, the source current of the top MOSFET is a square wave of duty cycle (V OUT)/(VIN). To prevent large voltage transients, a low ESR capacitor sized for the

current ratings are often based on only 2000 hours of life. (C1) and the VIN pin provides further isolation. capacitance and ΔIL is the ripple current in the inductor . since ΔIL increases with input voltage. RUN pin below 1.16V shuts down the main control loop. the LTC7800 draws only 14μA of quiescent current. RUN pin does not exceed 100μA. off VIN, as shown in Figure 4.

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Figure 3. Setting Output Voltage

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Figure 4. Using the RUN Pin as a UVLO

to track another supply during start-up. from the TRACK/SS pin to ground, as shown in Figure 5. providing a linear ramping voltage at the TRACK/SS pin. OUT to rise smoothly from 0V to its final regulated value. pin of the slave supply (V OUT), as shown in Figure 7.

• RTRACKA + RTRACKB RA + RB For coincident tracking (VOUT = VX during start-up): RA = RTRACKA RB = RTRACKB

Figure 5. Using the TRACK/SS Pin to Program Soft-Start

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Figure 6. T wo Different Modes of Output Voltage T racking

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Figure 7. Using the TRACK/SS Pin for T racking

7800f For more information www.linear.com/LTC7800 INTVCC Regulators The LTC7800 features two separate internal P-channel low dropout linear regulators (LDO) that supply power at the INTV CC pin from either the V IN supply pin or the EXTVCC pin depending on the connection of the EXTV CC pin. INTVCC powers the gate drivers and much of the LTC7800’s internal circuitry. The VIN LDO and the EXTVCC LDO regulate INTVCC to 5.1V. Each of these can supply a peak current of at least 50mA and must be bypassed to ground with a minimum of 2.2μF ceramic capacitor . No matter what type of bulk capacitor is used, an additional 1μF ceramic capacitor placed directly adjacent to the INTV CC and PGND pins is highly recommended. Good bypassing is needed to supply the high transient currents required by the MOSFET gate drivers. High input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC7800 to be exceeded. The INTV CC current, which is dominated by the gate charge current, may be supplied by either the V IN LDO or the EXTVCC LDO. When the voltage on the EXTVCC pin is less than 4.7V, the VIN LDO is enabled. Power dissipation for the IC in this case is highest and is equal to VIN • IINTVCC. The gate charge current is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equa- tions given in Note 3 of the Electrical Characteristics. For example, the LTC7800 INTVCC current is limited to less than 26mA from a 40V supply when not using the EXTVCC supply at a 70°C ambient temperature: To prevent the maximum junction temperature from be - ing exceeded, the input supply current must be checked while operating in forced continuous mode (PLLIN/MODE = INT VCC) at maximum VIN. When the voltage applied to EXTVCC rises above 4.7V, the VIN LDO is turned off and the EXTVCC LDO is enabled. The EXTVCC LDO remains on as long as the voltage applied to EXTVCC remains above 4.5V. The EXTV CC LDO attempts to regulate the INTVCC voltage to 5.1V, so while EXTV CC is less than 5.1V, the LDO is in dropout and the INTV CC voltage is approximately equal to EXTVCC. When EXTVCC is greater than 5.1V, up to an absolute maximum of 14V, INTV CC is regulated to 5.1V. Using the EXTV CC LDO allows the MOSFET driver and control power to be derived from the LTC7800’s switch- ing output (4.7V ≤ VOUT ≤ 14V) during normal operation and from the VIN LDO when the output is out of regulation (e.g., start-up, short-circuit). If more current is required through the EXTV CC LDO than is specified, an external Schottky diode can be added between the EXTV CC and INTVCC pins. In this case, do not apply more than 6V to the EXTVCC pin and make sure that EXTVCC ≤ VIN. Significant efficiency and thermal gains can be realized by powering INTV CC from the output, since the V IN cur- rent resulting from the driver and control currents will be scaled by a factor of (Duty Cycle)/(Switcher Efficiency). For 5V to 14V regulator outputs, this means connecting the EXTVCC pin directly to VOUT. Tying the EXTVCC pin to an 8.5V supply reduces the junction temperature in the previous example from 125°C to: However , for 3.3V and other low voltage outputs, additional circuitry is required to derive INTV CC power from the output. The following list summarizes the three possible connec- tions for EXTVCC: 1. EXT VCC Grounded. This will cause INTVCC to be powered from the internal 5.1V regulator resulting in an efficiency penalty of up to 10% at high input voltages. EXT VCC Connected Directly to VOUT. This is the normal connection for a 5V to 14V regulator and provides the highest efficiency. EXT VCC Connected to an External Supply. If an external supply is available in the 5V to 14V range, it may be used to power EXTV CC providing it is compatible with the MOSFET gate drive requirements. Ensure that EXTVCC < VIN. APPLICATIONS INFORMATION

7800fFor more information www.linear.com/LTC7800 APPLICATIONS INFORMATION Topside MOSFET Driver Supply (CB, DB) An external bootstrap capacitor , C B, connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET . Capacitor CB in the Functional Diagram is charged though external diode DB from INTVCC when the SW pin is low. When the topside MOSFET is to be turned on, the driver places the C B voltage across the gate-source of the MOSFET . This enhances the top MOSFET switch and turns it on. The switch node voltage, SW , rises to V IN and the BOOST pin follows. With the topside MOSFET on, the boost voltage is above the input supply: V BOOST = VIN + VINTVCC. The value of the boost capacitor , CB, needs to be 100 times that of the total input capacitance of the top - side MOSFET(s). The reverse breakdown of the external Schottky diode must be greater than V IN(MAX). Fault Conditions: Current Limit and Current Foldback The LTC7800 includes current foldback to help limit load current when the output is shorted to ground. If the output voltage falls below 70% of its nominal output level, then the maximum sense voltage is progressively lowered from 100% to 45% of its maximum selected value. Under short-circuit conditions with very low duty cycles, the LTC7800 will begin cycle skipping in order to limit the short-circuit current. In this situation the bottom MOSFET will be dissipating most of the power but less than in normal operation. The short-circuit ripple current is determined by the minimum on-time, t ON(MIN), of the LTC7800 (≈45ns), the input voltage and inductor value: ΔIL(SC) = tON(MIN) VIN L The resulting average short-circuit current is: ISC = 45% • ILIM(MAX) – 1 2 ΔIL(SC) Fault Conditions: Overvoltage Protection (Crowbar) The overvoltage crowbar is designed to blow a system input fuse when the output voltage of the regulator rises much higher than nominal levels. The crowbar causes huge currents to flow, that blow the fuse to protect against a shorted top MOSFET if the short occurs while the control- ler is operating. A comparator monitors the output for overvoltage condi- tions. The comparator detects faults greater than 10% above the nominal output voltage. When this condition is sensed, the top MOSFET is turned off and the bottom MOSFET is turned on until the over voltage condition is cleared. The bottom MOSFET remains on continuously for as long as the over voltage condition persists; if VOUT returns to a safe level, normal operation automatically resumes. A shorted top MOSFET will result in a high current condition which will open the system fuse. The switching regulator will regulate properly with a leaky top MOSFET by altering the duty cycle to accommodate the leakage. Frequency Synchronization and Selection The LTC7800 has an internal phase-locked loop (PLL) comprised of a phase frequency detector , a lowpass filter , and a voltage-controlled oscillator (VCO). This allows the turn-on of the top MOSFET to be locked to the rising edge of an external clock signal applied to the PLLIN/MODE pin. The phase detector is an edge sensitive digital type that provides zero degrees phase shift between the external and internal oscillators. This type of phase detector does not exhibit false lock to harmonics of the external clock. If the external clock frequency is greater than the inter - nal oscillator’s frequency, f OSC, then current is sourced continuously from the phase detector output, pulling up the VCO input. When the external clock frequency is less than f OSC, current is sunk continuously, pulling down the VCO input. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. The voltage at the VCO input is adjusted until the phase and frequency of the internal and external oscillators are identical. At the stable operating point, the phase detector output is high impedance and the internal filter capacitor , CLP , holds the voltage at the VCO input.

least 2.5V and down to 0.5V or less. large range of frequencies as the PLL locks. that the LTC7800 is capable of turning on the top MOSFET . but the ripple voltage and current will increase. minimum on-time gradually increases up to about 70ns. respondingly larger current and voltage ripple. the output power divided by the input power times 100%.

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Figure 8. Relationship Between Oscillator Frequency and 320kHz and 2.25MHz, as shown in Figure 8.

7800fFor more information www.linear.com/LTC7800 It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a per cent- age of input power . Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC7800 circuits: 1) IC V IN current, 2) IN- TVCC regulator current, 3) I2R losses, 4) topside MOSFET transition losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics table, which excludes MOSFET driver and control currents. VIN current typically results in a small (<0.1%) loss. 2. INT VCC current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge, dQ, moves from INTV CC to ground. The resulting dQ/dt is a current out of INTV CC that is typically much larger than the control circuit current. In continuous mode, I GATECHG = f(QT + QB), where QT and QB are the gate charges of the topside and bottom side MOSFETs. Supplying INTVCC from an output-derived source power through EXTVCC will scale the VIN current required for the driver and control circuits by a factor of (Duty Cycle)/ (Efficiency). For example, in a 20V to 5V application, 10mA of INTV CC current results in approximately 2.5mA of VIN current. This reduces the midcurrent loss from 10% or more (if the driver was powered directly from V IN) to only a few percent. 3. I2R losses are predicted from the DC resistances of the fuse (if used), MOSFET , inductor , current sense resis- tor and input and output capacitor ESR. In continuous mode the average output current flows through L and APPLICATIONS INFORMATION RSENSE, but is chopped between the topside MOSFET and the synchronous MOSFET . If the two MOSFETs have approximately the same R DS(ON), then the resistance of one MOSFET can simply be summed with the resis- tances of L, RSENSE and ESR to obtain I2R losses. For example, if each RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE = 10mΩ and R ESR = 40mΩ (sum of both input and output capacitance losses), then the total resistance is 130mΩ. This results in losses ranging from 3% to 13% as the output current increases from 1A to 5A for a 5V output, or a 4% to 20% loss for a 3.3V output. Efficiency varies as the inverse square of V OUT for the same external components and output power level. The combined effects of increasingly lower output voltages and higher currents required by high performance digital systems is not doubling but quadrupling the importance of loss terms in the switching regulator system! T ransition losses apply only to the topside MOSFET(s), and become significant only when operating at high input voltages (typically 15V or greater). T ransition losses can be estimated from: T ransition Loss = (1.7) • VIN2 • IO(MAX) • CRSS • f Other hidden losses such as copper trace and internal battery resistances can account for an additional 5% to 10% efficiency degradation in portable systems. It is very important to include these system level losses during the design phase. The internal battery and fuse resistance losses can be minimized by making sure that C IN has adequate charge storage and very low ESR at the switching frequency. A 25W supply will typically require a minimum of 20μF to 40μF of capacitance having a maximum of 20mΩ to 50mΩ of ESR. Other losses including body diode conduction losses during dead-time and inductor core losses generally account for less than 2% total additional loss. Checking T ransient Response The regulator loop response can be checked by looking at the load current transient response. Switching regulators

7800f For more information www.linear.com/LTC7800 APPLICATIONS INFORMATION take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to ΔILOAD (ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or discharge COUT generating the feedback error signal that forces the regulator to adapt to the current change and return V OUT to its steady-state value. During this recov - ery time VOUT can be monitored for excessive overshoot or ringing, which would indicate a stability problem. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The availability of the ITH pin not only allows optimization of control loop behavior , but it also provides a DC coupled and AC filtered closed-loop response test point. The DC step, rise time and settling at this test point truly reflects the closed-loop response. Assuming a predominantly second order system, phase margin and/ or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also be estimated by examining the rise time at the pin. The ITH external components shown in Figure 9 circuit will provide an adequate starting point for most applications. The ITH series RC-CC filter sets the dominant pole-zero loop compensation. The values can be modified slightly to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because the various types and values determine the loop gain and phase. An output current pulse of 20% to 80% of full-load current having a rise time of 1μs to 10μs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly across the output ca - pacitor and driving the gate with an appropriate signal generator is a practical way to produce a realistic load step condition. The initial output voltage step resulting from the step change in output current may not be within the bandwidth of the feedback loop, so this signal cannot be used to determine phase margin. This is why it is better to look at the ITH pin signal which is in the feedback loop and is the filtered and compensated control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop will be increased by de - creasing CC. If RC is increased by the same factor that CC is decreased, the zero frequency will be kept the same, thereby keeping the phase shift the same in the most critical frequency range of the feedback loop. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply per formance. A second, more severe transient is caused by switching in loads with large (>1μF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with C OUT, causing a rapid drop in VOUT. No regulator can alter its delivery of current quickly enough to prevent this sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of C LOAD to COUT is greater than 1:50, the switch rise time should be controlled so that the load rise time is limited to approximately 25 • C LOAD. Thus a 10μF capacitor would require a 250μs rise time, limiting the charging current to about 200mA. Design Example As a design example, assume V IN = 12V (nominal), VIN = 22V (max), V OUT = 3.3V, I MAX = 5A, V SENSE(MAX) = 75mV and f = 1MHz. The inductance value is chosen first based on a 30% ripple current assumption. The highest value of ripple current occurs at the maximum input voltage. Tie the FREQ pin with a 54.9k resistor to GND, generating approximately 1MHz operation. The inductor ripple current can be calculated from the following equation: ΔIL = VOUT f( ) L( ) 1– VOUT VIN(NOM) A 1.5μH inductor will produce 32% ripple current. The peak inductor current will be the maximum DC value plus one half the ripple current, or 5.8A. Increasing the ripple current will also help ensure that the minimum on-time of 45ns is not violated. The minimum on-time occurs at maximum V IN: tON(MIN) = VOUT VIN(MAX) f( ) = 3.3V 22V 1MHz( ) = 150ns

7800fFor more information www.linear.com/LTC7800 APPLICATIONS INFORMATION The equivalent RSENSE resistor value can be calculated by using the minimum value for the maximum current sense threshold (64mV): RSENSE ≤ 64mV 5.8A ≈ 0.01Ω Choosing 1% resistors: RA = 25k and R B = 78.7k yields an output voltage of 3.32V. The power dissipation on the topside MOSFET can be easily estimated. Choosing an Infineon BSZ097N04LSG MOSFET results in: R DS(ON) = 11.4mΩ, CMILLER = 16pF. At maximum input voltage with T(estimated) = 50°C: PMAIN = 3.3V 22V 5A( ) 11.4mΩ ( ) + 22V( ) 2 5A 5V – 1.5V + 1 1.5V  1MHz( ) = 94mW PSYNC = 22V – 3.3V( ) 22V 5A( ) = 273mW A short-circuit to ground will result in a folded back cur- rent of: ISC = 34mV 0.01Ω – 1 45ns 22V( ) 1.5µH  = 3.07A with a typical value of RDS(ON) and δ = (0.005/°C)(25°C) = 0.125. The resulting power dissipated in the bottom MOSFET is: PSYNC,SC = 3.07A( ) = 121 mW CIN is chosen for an RMS current rating of at least 3A at temperature. COUT is chosen with an ESR of 0.02Ω for low output ripple. The output ripple in continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR(∆IL) = 0.02Ω(1.60A) = 32mVP-P PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the IC. Check the following in your layout: Are the signal and power grounds kept separate? The combined IC signal ground pin and the ground return of CINTVCC must return to the combined C OUT (–) ter- minals. The path formed by the top N-channel MOSFET , Schottky diode and the CIN capacitor should have short leads and PC trace lengths. The output capacitor (–) terminals should be connected as close as possible to the (–) terminals of the input capacitor by placing the capacitors next to each other and away from the Schottky loop described above. Does the LTC7800 VFB pin’s resistive divider connect to the (+) terminal of COUT? The resistive divider must be connected between the (+) terminal of COUT and signal ground. The feedback resistor connections should not be along the high current input feeds from the input capacitor(s). Are the SENSE– and SENSE+ leads routed together with minimum PC trace spacing? The filter capacitor between SENSE+ and SENSE – should be as close as possible to the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor . Is the INTV CC decoupling capacitor connected close to the IC, between the INTV CC and the power ground pins? This capacitor carries the MOSFET drivers’ cur- rent peaks. An additional 1μF ceramic capacitor placed immediately next to the INTV CC and PGND pins can help improve noise performance substantially. 5. Keep the SW , TG, and BOOST nodes away from sensi- tive small-signal nodes. All of these nodes have ver y large and fast moving signals and therefore should be kept on the output side of the LTC7800 and occupy minimum PC trace area.

7800f For more information www.linear.com/LTC7800 APPLICATIONS INFORMATION 6. Use a modified star ground technique: a low impedance, large copper area central grounding point on the same side of the PC board as the input and output capacitors with tie-ins for the bottom of the INTV CC decoupling capacitor , the bottom of the voltage feedback resistive divider and the SGND pin of the IC. PC Board Layout Debugging It is helpful to use a DC-50MHz current probe to monitor the current in the inductor while testing the circuit. Monitor the output switching node (SW pin) to synchronize the oscilloscope to the internal oscillator and probe the actual output voltage as well. Check for proper performance over the operating voltage and current range expected in the application. The frequency of operation should be main- tained over the input voltage range down to dropout and until the output load drops below the low current opera- tion threshold—typically 25% of the maximum designed current level in Burst Mode operation. Th e duty cycle percentage should be maintained from cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic rate can sug- gest noise pickup at the current or voltage sensing inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Reduce V IN from its nominal level to verify operation of the regulator in dropout. Check the operation of the un- dervoltage lockout circuit by further lowering V IN while monitoring the outputs to verify operation. Investigate whether any problems exist only at higher out- put currents or only at higher input voltages. If problems coincide with high input voltages and low output currents, look for capacitive coupling between the BOOST , SW , TG, and possibly BG connections and the sensitive voltage and current pins. The capacitor placed across the current sensing pins needs to be placed immediately adjacent to the pins of the IC. This capacitor helps to minimize the effects of differential noise injection due to high frequency capacitive coupling. If problems are encountered with high current output loading at lower input voltages, look for inductive coupling between C IN, Schottky and the top MOSFET components to the sensitive current and voltage sensing traces. In addition, investigate common ground path voltage pickup between these components and the SGND pin of the IC. An embarrassing problem, which can be missed in an otherwise properly working switching regulator , results when the current sensing leads are hooked up backwards. The output voltage under this improper hookup will still be maintained but the advantages of current mode control will not be realized. Compensation of the voltage loop will be much more sensitive to component selection. This behavior can be investigated by temporarily shorting out the current sensing resistor—don’t worry, the regulator will still maintain control of the output voltage.

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CURRENT. KEEP LINES TO A MINIMUM LENGTH.

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Figure 9. Recommended Printed Circuit Layout Diagram Figure 10. Branch Current Waveforms

Figure 11. High Efficiency 3.3V 2.1MHz Step-Down Regulator THE OVERALL SYSTEM AND PRINTED CIRCUIT BOARD DESIGN.

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Figure 12. High Efficiency 5V 2.1MHz Step-Down Regulator

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THE OVERALL SYSTEM AND PRINTED CIRCUIT BOARD DESIGN.

Figure 13. High Efficiency 3.3V 320kHz Step-Down Regulator

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Figure 14. High Efficiency 5V 450kHz Step-Down Regulator Using GaN FETs

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THE OVERALL SYSTEM AND PRINTED CIRCUIT BOARD DESIGN.

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Figure 15. High Efficiency 12V 320kHz Step-Down Regulator Using GaN FETs

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THE OVERALL SYSTEM AND PRINTED CIRCUIT BOARD DESIGN.

7800fFor more information www.linear.com/LTC7800 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However , no responsibility is assumed for its use. Linear Technology Corporation makes no representa- tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. PACKAGE DESCRIPTION Please refer to http://www.linear .com/product/LTC7800#packaging/ for the most recent package drawings. 3.00 ±0.10 1.50 REF 4.00 ±0.10 NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE PIN 1 TOP MARK (NOTE 6) 0.40 ±0.10 19 20 BOTTOM VIEW—EXPOSED PAD

2.50 REF

0.75 ±0.05 R = 0.115 TYP PIN 1 NOTCH R = 0.20 OR 0.25 × 45° CHAMFER 0.25 ±0.05

0.50 BSC

0.200 REF

0.00 – 0.05 (UDC20) QFN 1106 REV Ø RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 0.70 ±0.05 0.25 ±0.05 3.10 ±0.05 4.50 ±0.05

1.50 REF

2.10 ±0.05 3.50 ±0.05 PACKAGE OUTLINE R = 0.05 TYP 1.65 ±0.10 2.65 ±0.10 1.65 ±0.05 2.65 ±0.05 20-Lead Plastic QFN (3mm × 4mm) (Reference LTC DWG # 05-08-1742 Rev Ø)

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THE OVERALL SYSTEM AND PRINTED CIRCUIT BOARD DESIGN. Figure 16. High Efficiency 12V 2.1MHz Step-Down Regulator