LT8361 (Rev. B)
Document overview
- Manufacturer or author: Analog Devices, Inc.
- PDF pages: 28
Technical content
Rev. BFor more information www.analog.com Document Feedback TYPICAL APPLICATION FEATURES DESCRIPTION Low IQ Boost/SEPIC/Inverting Converter with 2A, 100V Switch The LT®8361 is a current mode DC/DC converter with a 100V, 2A switch operating from a 2.8V to 60V input. With a unique single feedback pin architecture it is capable of boost, SEPIC or inverting configurations. Burst Mode operation consumes as low as 9µA quiescent current to maintain high efficiency at very low output currents, while keeping typical output ripple below 15mV. An external compensation pin allows optimization of loop bandwidth over a wide range of input and output volt - ages and programmable switching frequencies between 300kHz and 2MHz. A SYNC/MODE pin allows synchroni- zation to an external clock. It can also be used to select between burst or pulse-skip modes of operation with or without Spread Spectrum Frequency Modulation for low EMI. For increased efficiency, a BIAS pin can accept a second input to supply the INTV CC regulator . Additional features include frequency foldback and programmable soft-start to control inductor current during startup. The LT8361 is available in a thermally enhanced 16-lead MSOP package with four pins removed for high voltage pin spacings. 400kHz, 24V Output SEPIC Converter
APPLICATIONS
n Wide Input Voltage Range: 2.8V to 60V n Ultralow Quiescent Current and Low Ripple Burst Mode® Operation: IQ = 9µA n 2A, 100V Power Switch n Positive or Negative Output Voltage Programming with a Single Feedback Pin n Programmable Frequency (300kHz to 2MHz) n Synchronizable to an External Clock n Spread Spectrum Frequency Modulation for Low EMI n BIAS Pin for Higher Efficiency n Programmable Undervoltage Lockout (UVLO) n Thermally Enhanced 16-lead MSOP packages n AEC-Q100 Qualified for Automotive Applications n Industrial and Automotive n Telecom n Medical Diagnostic Equipment n Portable Electronics All registered trademarks and trademarks are the property of their respective owners. Efficiency and Power Loss 1µF 4.7µF 6.8nF 0.22µF 71.5k
8361 TA01a
16.2k 121k 10µF 22µH 22µH 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 4V TO 48V 24V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 744877220 C5: MURATA GRM32ER71H106KA12L 200mA AT VIN = 5V 450mA AT VIN = 12V 550mA AT VIN = 24V 600mA AT VIN = 48V EFFICIENCY POWER LOSS V IN = 12V V IN = 48V LOAD CURRENT (A) 0.1 0.2 0.3 0.4 0.5 0.6 0.7 100 0.25 0.50 0.75 1.00 1.25 1.50 1.75 2.00 2.25 2.50 EFFICIENCY (%) POWER LOSS (W)
8361 TA01b
Rev. B For more information www.analog.com Operating Junction Temperature (Note 3) ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8361EMSE#PBF LT8361EMSE#TRPBF 8361 16-Lead Plastic MSOP with 4 Pins Removed –40°C to 125°C LT8361IMSE#PBF LT8361IMSE#TRPBF 8361 16-Lead Plastic MSOP with 4 Pins Removed –40°C to 125°C LT8361HMSE#PBF LT8361HMSE#TRPBF 8361 16-Lead Plastic MSOP with 4 Pins Removed –40°C to 150°C AUTOMOTIVE PRODUCTS** LT8361EMSE#WPBF LT8361EMSE#WTRPBF 8361 16-Lead Plastic MSOP with 4 Pins Removed –40°C to 125°C LT8361IMSE#WPBF LT8361IMSE#WTRPBF 8361 16-Lead Plastic MSOP with 4 Pins Removed –40°C to 125°C LT8361HMSE#WPBF LT8361HMSE#WTRPBF 8361 16-Lead Plastic MSOP with 4 Pins Removed –40°C to 150°C Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container . Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. **Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for these models. EN/UVLO VIN INTVCC NC BIAS VC SW1 SW2 SYNC/MODE SS RT FBX TOP VIEW PGND, GND MSE PACKAGE VARIATION: MSE16 (12) 16-LEAD PLASTIC MSOP θJA = 45°C/W , θJC = 10°C/W EXPOSED PAD (PIN 17) IS PGND AND GND, MUST BE SOLDERED TO PCB PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS (Note 1)
Rev. BFor more information www.analog.com PARAMETER CONDITIONS MIN TYP MAX UNITS VIN Operating Voltage Range l 2.8 60 V VIN Quiescent Current at Shutdown VEN/UVLO = 0.2V l μA μA VEN/UVLO = 1.5V l μA μA VIN Quiescent Current Sleep Mode (Not Switching) SYNC = 0V l μA μA Active Mode (Not Switching) SYNC = 0V or INTVCC, BIAS = 0V l 1200 1200 1600 1850 µA µA SYNC = 0V or INTVCC, BIAS = 5V l µA µA BIAS Threshold Rising, BIAS Can Supply INTVCC Falling, BIAS Cannot Supply INTVCC 4.4 4.65 4.25 V V VIN Falling Threshold to Supply INTVCC BIAS = 12V BIAS – 2V V BIAS Falling Threshold to Supply INTVCC VIN = 12V VIN V FBX Regulation FBX Regulation Voltage FBX > 0V FBX < 0V l l 1.568 –0.822 1.6 –0.80 1.636 –0.780 V V FBX Line Regulation FBX > 0V, 2.8V < VIN < 60V FBX < 0V, 2.8V < VIN < 60V 0.005 0.005 0.015 0.015 %/V %/V FBX Pin Current FBX = 1.6V, –0.8V l –10 10 nA Oscillator Switching Frequency (fOSC) RT = 165k RT = 45.3k RT = 20k l l l 265 0.90 1.85 300 327 1.08 2.15 kHz MHz MHz SSFM Maximum Frequency Deviation (∆f/fOSC) • 100, RT = 20k 14 20 28 % Minimum On-Time Burst Mode, VIN = 24V (Note 6) Pulse-Skip Mode, VIN = 24V (Note 6) ns ns Minimum Off-Time l 55 75 ns SYNC/Mode, Mode Thresholds (Note 5) High (Rising), VIN = 24V Low (Falling), VIN = 24V l l 0.14 1.3 0.2 1.7 V V SYNC/Mode, Clock Thresholds (Note 5) Rising, VIN = 24V Falling, VIN = 24V l l 0.4 1.3 0.8 1.7 V V fSYNC/fOSC Allowed Ratio RT = 20k 0.95 1 1.25 kHz/kHz SYNC Pin Current SYNC = 2V SYNC = 0V, Current Out of Pin µA µA Switch Maximum Switch Current Limit Threshold l 2 2.5 3.4 A Switch Overcurrent Threshold Discharges SS Pin 3.75 A Switch RDS(ON) ISW = 0.5A 375 mΩ Switch Leakage Current VSW = 100V 0.1 1 µA ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, EN/UVLO = 12V unless otherwise noted.
Rev. B For more information www.analog.com ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, EN/UVLO = 12V unless otherwise noted. Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: INTVCC cannot be externally driven. No additional components or loading is allowed on this pin. Note 3: The LT8361E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT8361I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8361H is guaranteed over the full –40°C to 150°C operating junction temperature range. Note 4: The IC includes overtemperature protection that is intended to protect the device during overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature will reduce lifetime. Note 5: For SYNC/MODE inputs required to select modes of operation see the Pin Functions and Applications Information sections. Note 6: The IC is tested in a Boost converter configuration with the output voltage programmed for 24V. PARAMETER CONDITIONS MIN TYP MAX UNITS EN/UVLO Logic EN/UVLO Pin Threshold (Rising) Start Switching l 1.576 1.68 1.90 V EN/UVLO Pin Threshold (Falling) Stop Switching l 1.545 1.6 1.645 V EN/UVLO Pin Current VEN/UVLO = 1.6V l –50 50 nA Soft-Start Soft-Start Charge Current SS = 0.5V 2 µA Soft-Start Pull-Down Resistance Fault Condition, SS = 0.1V 220 Ω Error Amplifier Error Amplifier T ransconductance FBX = 1.6V FBX = –0.8V µA/V µA/V Error Amplifier Voltage Gain FBX = 1.6V FBX = –0.8V 185 145 V/V V/V Error Amplifier Max Source Current VC = 1.1V, Current Out of Pin 7 µA Error Amplifier Max Sink Current VC = 1.1V 7 µA
Rev. BFor more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS Switching Frequency vs Temperature Switching Frequency vs VIN Normalized Switching Frequency vs FBX Voltage FBX Positive Regulation Voltage vs Temperature FBX Negative Regulation Voltage vs Temperature EN/UVLO Pin Thresholds vs Temperature JUNCTION TEMPERATURE (°C) –50 –25 100 125 150 175 1.568 1.576 1.584 1.592 1.600 1.608 1.616 1.624 1.632 FBX VOL TAGE (V)
8361 G01
V IN = 12V JUNCTION TEMPERATURE (°C) –50 –25 100 125 150 175 –0.820 –0.815 –0.810 –0.805 –0.800 –0.795 –0.790 –0.785 –0.780 FBX VOL TAGE (V)
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V IN = 12V EN/UVLO RISING (TURN–ON) EN/UVLO FALLING (TURN–OFF) JUNCTION TEMPERATURE (°C) –50 –25 100 125 150 175 1.54 1.56 1.58 1.60 1.62 1.64 1.66 1.68 1.70 1.72 1.74 EN/UVLO PIN VOL TAGE (V)
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JUNCTION TEMPERATURE (°C) –50 –25 100 125 150 175 1.90 1.92 1.94 1.96 1.98 2.00 2.02 2.04 2.06 2.08 2.10 SWITCHING FREQUENCY (MHz)
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V IN = 12V V IN (V) 1.90 1.92 1.94 1.96 1.98 2.00 2.02 2.04 2.06 2.08 2.10 SWITCHING FREQUENCY (MHz)
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VOL TAGE (V) –0.8 –0.4 0.0 0.4 0.8 1.2 1.6 100 125 NORMALIZED SWITCHING FREQUENCY (%)
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V IN = 12V Switch Current Limit vs Duty Cycle Switch Minimum On-Time vs Temperature Switch Minimum Off-Time vs Temperature DUTY CYCLE (%) 100 2.0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 SWITCH CURRENT LIMIT (A)
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V IN = 12V V IN = 12V –50 –25 JUNCTION TEMPERATURE (°C) 100 125 150 175 100 MINIMUM ON TIME (ns)
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V IN = 12V –50 –25 100 125 150 175 100 MINIMUM OFF TIME (ns)
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JUNCTION TEMPERATURE (°C)
Rev. B For more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS Switching Waveforms (in CCM) Switching Waveforms (in DCM/Light Burst Mode) Switching Waveforms (in Deep Burst Mode) Burst Frequency vs Load Current VOUT T ransient Response: Load Current T ransients from 390mA to 790mA to 390mA VOUT T ransient Response: Load Current T ransients from 150mA to 790mA to 150mA VIN Pin Current (Sleep Mode, Not Switching) vs Temperature VIN Pin Current (Active Mode, Not Switching, Bias = 0V) vs Temperature VIN Pin Current (Active Mode, Not Switching, Bias = 5V) vs Temperature V IN = 12 V V BIAS = 0V V SYNC_MODE = 0V JUNCTION TEMPERATURE (°C) –75 –50 –25 100 125 150 175 V IN PIN CURRENT (µA)
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V IN = 12V V BIAS = 0V V SYNC_MODE = FLOAT JUNCTION TEMPERATURE (°C) –50 –25 100 125 150 175 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 V IN PIN CURRENT (mA)
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V IN = 12V V BIAS = 5V V SYNC_MODE = FLOAT JUNCTION TEMPERATURE (°C) –50 –25 100 125 150 175 V IN PIN CURRENT (µA)
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I L1 + IL2 500mA/DIV V SW 20V/DIV
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1µs/DIV IL1 + IL2 200mA/DIV VSW 20V/DIV
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V IN = 24V V OUT = 24V 100µs/DIV V OUT 500mV/DIV I OUT 500mA/DIV
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V IN = 24V V OUT = 24V 100µs/DIV V OUT 1V/DIV I OUT 500mA/DIV
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V IN = 12V V OUT = 48V LOAD CURRENT (mA) 100 0.5 1.0 1.5 2.0 2.5 SWITCHING FREQUENCY (MHz)
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Rev. BFor more information www.analog.com PIN FUNCTIONS EN/UVLO: Shutdown and Undervoltage Detect Pin. The LT8361 is shut down when this pin is low and active when this pin is high. Below an accurate 1.6V threshold, the part enters undervoltage lockout and stops switching. This allows an undervoltage lockout (UVLO) threshold to be programmed for system input voltage by resistively dividing down system input voltage to the EN/UVLO pin. An 80mV pin hysteresis ensures part switching resumes when the pin exceeds 1.68V. EN/UVLO pin voltage below 0.2V reduces V IN current below 1µA. If shutdown and UVLO features are not required, the pin can be tied directly to system input. VIN: Input Supply. This pin must be locally bypassed. Be sure to place the positive terminal of the input capacitor as close as possible to the VIN pin, and the negative terminal as close as possible to the exposed pad PGND copper (near EN/UVLO). INTVCC: Regulated 3.8V Supply for Internal Loads. The INTVCC pin must be bypassed with a 1µF low ESR ceramic capacitor to GND. No additional components or loading is allowed on this pin. INTVCC draws power from the BIAS pin if 4.4V ≤ BIAS ≤ VIN, otherwise INTVCC is powered by the VIN pin. NC: No Internal Connection. Leave this pin open. BIAS: Second Input Supply for Powering INTV CC. Removes the majority of INTVCC current from the VIN pin to improve efficiency when 4.4V ≤ BIAS ≤ VIN. If unused, tie the pin to GND. VC: Error Amplifier Output Pin. Tie external compensation network to this pin. FBX: Voltage Regulation Feedback Pin for Positive or Negative Outputs. Connect this pin to a resistor divider between the output and the exposed pad GND copper (near FBX). FBX reduces the switching frequency during start-up and fault conditions when FBX is close to 0V. RT: A resistor from this pin to the exposed pad GND cop- per (near FBX) programs switching frequency. SS: Soft-Start Pin. Connect a capacitor from this pin to GND copper (near FBX) to control the ramp rate of induc- tor current during converter start-up. SS pin charging current is 2μA. An internal 220Ω MOSFET discharges this pin during shutdown or fault conditions. SYNC/MODE: This pin allows five selectable modes for optimization of performance. SYNC/MODE Pin Input Capable Mode(s) of Operation (1) GND or <0.14V Burst (2) External Clock Pulse-skip/Sync (3) 100k Resistor to GND Burst/SSFM (4) Float (pin open) Pulse-skip (5) INTVCC or >1.7V Pulse-skip/SSFM where the selectable modes of operation are, Burst = low IQ, low output ripple operation at light loads Pulse-skip = skipped pulse(s) at light load (aligned to clock) Sync = switching frequency synchronized to external clock SSFM = Spread Spectrum Frequency Modulation for low EMI SW1, SW2 (SW) : Output of the Internal Power Switch. Minimize the metal trace area connected to these pins to reduce EMI. PGND,GND: Power Ground and Signal Ground for the IC. The package has an exposed pad underneath the IC which is the best path for heat out of the package. The pin should be soldered to a continuous copper ground plane under the device to reduce die temperature and increase the power capability of the LT8361. Connect power ground components to the exposed pad copper exiting near the EN/UVLO and SW pins. Connect signal ground components to the exposed pad copper exiting near the VC and FBX pins.
Rev. B For more information www.analog.com BLOCK DIAGRAM 8361 BD OPT VBIAS (+) VBIAS – 2V(–) 4.4V(+) 4.0V(–) 3.8V REGULATOR SW1 SW OSCILLATOR ERROR AMP SELECT FREQUENCY FOLDBACK INTVCC UVLO SWITCH LOGIC BURST DETECT SLOPE DRIVER INTVCC TJ > 170°C 1.68V(+) 1.6V(–) INTERNAL REFERENCE UVLO CIN SW2 BIAS OPT VIN COUT CVCC DL VOUT UVLO RT SYNC/MODE OVER- CURRENT OVERCURRENT PGND/GND ERROR AMP ERROR AMP SLOPE 1.6VFBX VOUT –0.8V MAX ILIMIT 1.5× MAX ILIMIT RSENSE PWM COMPARATOR ISS 2/uni03BCA SS CSS RC VC CC EN/UVLO VIN UVLO
Rev. BFor more information www.analog.com OPERATION The LT8361 uses a fixed frequency, current mode con - trol scheme to provide excellent line and load regulation. Operation can be best understood by referring to the Block Diagram. An oscillator (with frequency programmed by a resistor at the RT pin) turns on the internal power switch at the beginning of each clock cycle. Current in the induc- tor then increases until the current comparator trips and turns off the power switch. The peak inductor current at which the switch turns off is controlled by the voltage on the VC pin. The error amplifier servos the VC pin by com- paring the voltage on the FBX pin with an internal refer - ence voltage (1.60V or –0.80V, depending on the chosen topology). When the load current increases it causes a reduction in the FBX pin voltage relative to the internal ref- erence. This causes the error amplifier to increase the VC pin voltage until the new load current is satisfied. In this manner , the error amplifier sets the correct peak switch current level to keep the output in regulation. The LT8361 is capable of generating either a positive or negative output voltage with a single FBX pin. It can be configured as a boost or SEPIC converter to generate a positive output voltage, or as an inverting converter to generate a negative output voltage. When configured as a Boost converter , as shown in the Block Diagram, the FBX pin is pulled up to the internal bias voltage of 1.60V by a voltage divider ( R1 and R2) connected from V OUT to GND. Amplifier A2 becomes inactive and amplifier A1 performs (inverting) amplification from FBX to VC. When the LT8361 is in an inverting configuration, the FBX pin is pulled down to –0.80V by a voltage divider from VOUT to GND. Amplifier A1 becomes inactive and amplifier A2 performs (non-inverting) amplification from FBX to VC. If the EN/UVLO pin voltage is below 1.6V, the LT8361 enters undervoltage lockout (UVLO), and stops switch - ing. When the EN/UVLO pin voltage is above 1.68V (typi- cal), the LT8361 resumes switching. If the EN/UVLO pin voltage is below 0.2V, the LT8361 draws less than 1µA from VIN. For the SYNC/MODE pin tied to ground or <0.14V, the LT8361 will enter low output ripple Burst Mode opera - tion for ultra low quiescent current during light loads to maintain high efficiency. For a 100k resistor from SYNC/ MODE pin to GND, the LT8361 uses Burst Mode opera - tion for improved efficiency at light loads but seamlessly transitions to Spread-Spectrum Modulation of switch - ing frequency for low EMI at heavy loads. For the SYNC/ MODE pin floating (left open), the LT8361 uses pulse- skipping mode, at the expense of hundreds of microamps, to maintain output voltage regulation at light loads by skipping switch pulses. For the SYNC/MODE pin tied to INTVCC or >1.7V, the LT8361 uses pulse-skipping mode and performs Spread-Spectrum Modulation of switching frequency. For the SYNC/MODE pin driven by an external clock, the converter switching frequency is synchronized to that clock and pulse-skipping mode is also enabled. See the Pin Functions section for SYNC/MODE pin. The LT8361 includes a BIAS pin to improve efficiency across all loads. The LT8361 intelligently chooses between the VIN and BIAS pins to supply the INTVCC for best effi- ciency. The INTV CC supply current can be drawn from the BIAS pin instead of the VIN pin for 4.4V ≤ BIAS ≤ VIN. Protection features ensure the immediate disable of switching and reset of the SS pin for any of the following faults: internal reference UVLO, INTVCC UVLO, switch cur- rent > 1.5× maximum limit, EN/UVLO < 1.6V or junction temperature > 170°C.
Figure 1. Burst Frequency vs Load Current minimizing the input quiescent current and output ripple. capacitance will decrease the output ripple proportionally. Figure 2. Burst Mode Operation defined by the resistor at the RT pin as shown in Figure 1. operation. A logic input can also control the EN/UVLO pin. minimize their effect on efficiency at light loads.
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rents required by the internal power MOSFET gate driver . (to stop switching and reset soft-start) is typically 2.5V. with a local ceramic capacitor . Table 1. SW Frequency vs RT Value (this can be ground or a logic low output). clock, but instead will pulse skip to maintain regulation. tained at lower output load than in Burst Mode operation. ies the clock with a triangle frequency modulation of 20%. the switching frequency programmed using the RT pin.
Rev. B For more information www.analog.com DUTY CYCLE CONSIDERATION The LT8361 minimum on-time, minimum off-time and switching frequency (fOSC) define the allowable minimum and maximum duty cycles of the converter (see Minimum On-Time, Minimum Off-Time, and Switching Frequency in the Electrical Characteristics table). Minimum Allowable Duty Cycle = Minimum On-Time(MAX) • fOSC(MAX) Maximum Allowable Duty Cycle = 1 – Minimum Off-Time(MAX) • fOSC(MAX) The required switch duty cycle range for a Boost converter operating in continuous conduction mode (CCM) can be calculated as: DMIN = 1 – VIN(MAX) VOUT + VD DMAX = 1 – VIN(MIN) VOUT + VD where VD is the diode forward voltage drop. If the above duty cycle calculations for a given application violate the minimum and/or maximum allowed duty cycles for the LT8361, operation in discontinuous conduction mode (DCM) might provide a solution. For the same VIN and VOUT levels, operation in DCM does not demand as low a duty cycle as in CCM. DCM also allows higher duty cycle operation than CCM. The additional advantage of DCM is the removal of the limitations to inductor value and duty cycle required to avoid sub-harmonic oscilla - tions and the right half plane zero (RHPZ). While DCM provides these benefits, the trade-off is higher inductor peak current, lower available output power and reduced efficiency. SETTING THE OUTPUT VOL TAGE The output voltage is programmed with a resistor divider from the output to the FBX pin. Choose the resistor values for a positive output voltage according to: R1 = R2 • VOUT 1.60V – 1⎛ Choose the resistor values for a negative output voltage according to: R1 = R2 • |VOUT| 0.80V – 1⎛ The locations of R1 and R2 are shown in the Block Diagram. 1% resistors are recommended to maintain output voltage accuracy. Higher-value FBX divider resistors result in the lowest input quiescent current and highest light-load efficiency. FBX divider resistors R1 and R2 are usually in the range from 25k to 1M. SOFT-START The LT8361 contains several features to limit peak switch currents and output voltage (V OUT) overshoot during start-up or recovery from a fault condition. The primary purpose of these features is to prevent damage to external components or the load. High peak switch currents during start-up may occur in switching regulators. Since V OUT is far from its final value, the feedback loop is saturated and the regulator tries to charge the output capacitor as quickly as possible, resulting in large peak currents. A large surge current may cause inductor saturation or power switch failure. The LT8361 addresses this mechanism with a programma- ble soft-start function. As shown in the Block Diagram, the soft-start function controls the ramp of the power switch current by controlling the ramp of V C through Q1. This allows the output capacitor to be charged gradually toward its final value while limiting the start-up peak currents. Figure 3 shows the output voltage and supply current for the first page Typical Applications. It can be seen that both the output voltage and supply current come up gradually. FAUL T PROTECTION An inductor overcurrent fault ( > 3.75A) and/or INTV CC undervoltage (INTV CC < 2.5V) and/or thermal lockout (TJ > 170°C) will immediately prevent switching, will reset the SS pin and will pull down VC. Once all faults are removed, the LT8361 will soft-start VC and hence inductor peak current. APPLICATIONS INFORMATION
Figure 3. Soft-Start Waveforms the Typical Performance Characteristics section). the part will stop switching and go into thermal lockout. regulate the output which simplifies loop compensation. network is usually connected from the V C pin to GND. including load current, input voltage and temperature. Application Note 76 is a good reference. The LT8361 can be configured for different topologies. verter , followed by the SEPIC and inverting converters.
Rev. B For more information www.analog.com APPLICATIONS INFORMATION output condition, the inductor current is limited only by the input supply capability. For applications requiring a step-up converter that is short-circuit protected, please refer to the Applications Information section covering SEPIC converters. The conversion ratio as a function of duty cycle is: VOUT VIN = 1 1 − D in continuous conduction mode (CCM). For a boost converter operating in CCM, the duty cycle of the main switch can be calculated based on the output voltage (VOUT) and the input voltage (VIN). The maximum duty cycle (D MAX) occurs when the converter has the minimum input voltage: DMAX = VOUT − VIN(MIN) VOUT Discontinuous conduction mode (DCM) provides higher conversion ratios at a given frequency at the cost of reduced efficiencies, higher switching currents, and lower available output power . Boost Converter: Maximum Output Current Capability and Inductor Selection For the boost topology, the maximum average inductor current is: IL(MAX)(AVG)= IO(MAX) • 1 1 − DMAX
- 1 η where η (< 1.0) is the converter efficiency. Due to the current limit of its internal power switch, the LT8361 should be used in a boost converter whose maxi- mum output current (IO(MAX)) is: IO(MAX) ≤ VIN(MIN) VOUT
- 2A − 0.5 • ΔISW( ) • η Minimum possible inductor value and switching frequency should also be considered since they will increase inductor ripple current ∆ISW. The inductor ripple current ∆ISW has a direct effect on the choice of the inductor value and the converter’s maximum output current capability. Choosing smaller values of ∆ISW increases output current capability, but requires large inductances and reduces the current loop gain (the converter will approach voltage mode). Accepting larger values of ∆ISW provides fast transient response and allows the use of low inductances, but results in higher input current ripple and greater core losses, and reduces output current capability. It is recommended to choose a ∆ISW of approximately 0.75A. Given an operating input voltage range, and having cho- sen the operating frequency and ripple current in the inductor , the inductor value of the boost converter can be determined using the following equation: L = VIN(MIN) ΔISW • fOSC
- DMAX The peak inductor current is the switch current limit (max- imum 3.4A), and the RMS inductor current is approxi - mately equal to IL(MAX)(AVG). Choose an inductor that can handle at least 3.4A without sat- urating, and ensure that the inductor has a low DCR (copper- wire resistance) to minimize I2R power losses. Note that in some applications, the current handling requirements of the inductor can be lower , such as in the SEPIC topology where each inductor only carries one-half of the total switch cur- rent. For better efficiency, use similar valued inductors with a larger volume. Many different sizes and shapes are available from various manufacturers (see Table 2). Choose a core material that has low losses at the programmed switching frequency, such as a ferrite core. The final value chosen for the inductor should not allow peak inductor currents to exceed 2A in steady state at maximum load. Due to toler- ances, be sure to account for minimum possible inductance value, switching frequency and converter efficiency. For inductor current operation in CCM and duty cycles above 50%, the LT8361's internal slope compensation prevents sub-harmonic oscillations provided the inductor value exceeds a minimum value given by: L > VIN
- 2•D–1( ) 1–D( ) Lower L values are allowed if the inductor current oper - ates in DCM or duty cycle operation is below 50%.
Table 2. Inductor Manufacturers be provided with a low performance electrolytic capacitor . cerns the maximum input voltage rating of the LT8361. easily avoided (see Application Note 88). Figure 4. The Output Ripple Waveform of a Boost Converter
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high frequency switching noise on the converter output. capacitor at audio frequencies, generating audible noise. ceramic capacitors are also available. Figure 5. Suggested Boost Converter Layout Table 3. Ceramic Capacitor Manufacturers A Schottky diode is recommended for use with the LT8361. reverse voltage ratings for the target applications. Table 4. Recommended Schottky Diodes
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ground plane for better thermal performance. in continuous conduction mode (CCM). Figure 6. LT8361 Configured in a SEPIC Topology
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are specified in the Electrical Characteristics table. from the input source when the circuit is in shutdown.
- DMAX For most SEPIC applications, the equal inductor values will fall in the range of 2.2µH to 100µH. By making L1 = L2, and winding them on the same core, the value of inductance in the preceding equation is replaced by 2L, due to mutual inductance: L = VIN(MIN) ΔISW • fOSC
- DMAX This maintains the same ripple current and energy storage in the inductors. The peak inductor currents are: IL1(PEAK) = IL1(MAX) + 0.5 • ∆IL1 IL2(PEAK) = IL2(MAX) + 0.5 • ∆IL2 The maximum RMS inductor currents are approximately equal to the maximum average inductor currents. Based on the preceding equations, the user should choose the inductors having sufficient saturation and RMS cur - rent ratings. Similar to Boost converters, the SEPIC converter also needs slope compensation to prevent subharmonic oscillations while operating in CCM. The equation presented in the Boost Converter section defines the minimum inductance value to avoid sub-harmonic oscillations when coupled inductors are used. For uncoupled inductors, the minimum inductance requirement is doubled. SEPIC Converter: Output Diode Selection To maximize efficiency, a fast switching diode with a low forward drop and low reverse leakage is desirable. The average forward current in normal operation is equal to the output current.
8361 F07
Figure 7. The Switch Current Waveform of the SEPIC Converter loop gain (the converter will approach voltage mode).
margin (a 10V safety margin is usually sufficient). exceed the diode maximum junction temperature rating. SEPIC converter are similar to those of the boost converter . in continuous conduction mode (CCM). negative output voltage (VOUT) and the input voltage (VIN). are specified in the Electrical Characteristics table. Figure 8. A Simplified Inverting Converter
8361 F10
Rev. B For more information www.analog.com APPLICATIONS INFORMATION Inverting Converter: Output Capacitor Selection The inverting converter requires much smaller output capacitors than those of the boost, flyback and SEPIC converters for similar output ripples. This is due to the fact that, in the inverting converter , the inductor L2 is in series with the output, and the ripple current flowing through the output capacitors are continuous. The out - put ripple voltage is produced by the ripple current of L2 flowing through the ESR and bulk capacitance of the output capacitor: ΔVOUT(P–P) = ΔIL2 • ESRCOUT + 1 8 •fOSC •COUT After specifying the maximum output ripple, the user can select the output capacitors according to the preceding equation. The ESR can be minimized by using high quality X5R or X7R dielectric ceramic capacitors. In many applications, ceramic capacitors are sufficient to limit the output volt- age ripple. The RMS ripple current rating of the output capacitor needs to be greater than: IRMS(COUT) > 0.3 • ∆IL2 Inverting Converter: Selecting the DC Coupling Capacitor The DC voltage rating of the DC coupling capacitor (CDC, as shown in Figure 8) should be larger than the maximum input voltage minus the output voltage (negative voltage): VCDC > VIN(MAX) + VOUT CDC has nearly a rectangular current waveform. During the switch off-time, the current through CDC is IIN, while approximately –IO flows during the on-time. The RMS rating of the coupling capacitor is determined by the fol- lowing equation: IRMS(CDC) >IO(MAX) • DMAX 1 − DMAX A low ESR and ESL, X5R or X7R ceramic capacitor works well for CDC. 400kHz, 4V to 48V Input, 24V SEPIC Converter Efficiency 1µF 4.7µF 6.8nF 0.22µF 71.5k 16.2k 121k 10µF 22µH 22µH 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 4V TO 48V 24V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 744877220 C5: MURATA GRM32ER71H106KA12L 200mA AT V IN = 5V 450mA AT V IN = 12V 550mA AT V IN = 24V 600mA AT V IN = 48V V OUT
8361 TA02a
V IN = 5V V IN = 12V V IN = 24V V IN = 48V LOAD CURRENT (A) 0.001 0.01 0.1 100 EFFICIENCY (%)
8361 TA02b
Rev. BFor more information www.analog.com TYPICAL APPLICATIONS 450kHz, 3V to 60V Input, 12V SEPIC Converter 1µF 4.7µF 4.7nF 10nF 154k 28k 107k 10µF 22µH 22µH 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 3V TO 60V 12V D1: DIODES INC. B1100LB L1: WURTH ELEKTRONIK 744873220 C5: MURATA GRM32ER71H106KA12L 150mA AT V IN = 3V 390mA AT V IN = 5V 700mA AT V IN = 12V 900mA AT V IN = 24V 970mA AT V IN = 36V 1.1A AT V IN = 48V V OUT
8361 TA03a
V IN = 5V V IN = 12V V IN = 24V V IN = 48V LOAD CURRENT (A) 0.2 0.4 0.6 0.8 1.0 1.2 100 EFFICIENCY (%)
8361 TA03b
450kHz, 10V to 48V Input, 48V SEPIC Converter 1µF 4.7µF 3.3nF 10nF 34.8k 24k 107k 10µF 47µH 47µH 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 10V TO 48V 48V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 744873470 C5: MURATA GRM32ER71H106KA12L 250mA AT V IN = 10V 290mA AT V IN = 12V 460mA AT V IN = 24V 550mA AT V IN = 36V 610mA AT V IN = 48V V OUT
8361 TA04a
V IN = 12V V IN = 24V V IN = 48V LOAD CURRENT (A) 0.15 0.30 0.45 0.60 0.75 100 EFFICIENCY (%)
8361 TA04b
Rev. B For more information www.analog.com TYPICAL APPLICATIONS 2MHz, 10V to 54V Input, 65V Boost Converter 1µF 4.7µF 2.2nF 10nF 25.5k 22.1k 20k 1.5µF 6.1µH EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 10V TO 54V 65V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 74477006 C5: NIPPON CHEMI-CON KTS101B155M43N0T00 120mA AT V IN = 12V 310mA AT V IN = 24V 470mA AT V IN = 36V 560mA AT V IN = 48V 630mA AT V IN = 54V
8361 TA05a
V IN = 12V V IN = 24V V IN = 48V LOAD CURRENT (A) 0.1 0.2 0.3 0.4 0.5 0.6 100 EFFICIENCY (%)
8361 TA05b
450kHz, 5V to 60V Input, 80V Boost Converter 1µF 4.7µF 3.3nF 10nF 20.5k 30k 107k 3.3µF 68µH EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 5V TO 60V 80V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 744770168 C5: NIPPON CHEMI-CON KTS101B335M55N0T00 220mA AT V IN = 12V 450mA AT V IN = 24V 500mA AT V IN = 48V
8361 TA06a
V IN = 5V V IN = 12V V IN = 24V LOAD CURRENT (A) 0.10 0.20 0.30 0.40 0.50 100 EFFICIENCY (%)
8361 TA06b
Rev. BFor more information www.analog.com TYPICAL APPLICATIONS 1.2MHz, 8V to 16V Input, 25V to 80V Output Boost Converter 1µF 4.7µF 2.2nF 10nF 30.4k 22.1k 37.4k 1.5µF 6.1µH 60k EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 8V TO 16V 25V to 80V V C ONTROL 0V to 3.3V V OUT 80V to 25V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 74477006 C5: NIPPON CHEMI-CON KTS101B155M43N0T00 25V/480mA AT V IN = 12V 48V/230mA AT V IN = 12V 60V/180mA AT V IN = 12V 80V/100mA AT V IN = 12V
8361 TA07a
V IN = 12V , V OUT = 80V V IN = 12V , V OUT = 25V LOAD CURRENT (A) 0.10 0.20 0.30 0.40 0.50 100 EFFICIENCY (%)
8361 TA07b
450kHz, 4V to 60V Input, –24V Inverting Converter 1µF 4.7µF 2.2nF 10nF 34.8k 36.5k 107k 10µF 33µH 33µH 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 4V TO 60V –24V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 744873330 C5: MURATA GRM32ER71H106KA12L 150mA AT V IN = 5V 500mA AT V IN = 12V 750mA AT V IN = 24V 870mA AT V IN = 36V 920mA AT V IN = 48V
8361 TA08a
V IN = 5V V IN = 12V V IN = 24V V IN = 48V LOAD CURRENT (A) 0.2 0.4 0.6 0.8 0.9 100 EFFICIENCY (%)
8361 TA08b
Rev. B For more information www.analog.com TYPICAL APPLICATIONS 450kHz, 3V to 60V Input, –12V Inverting Converter 1µF 4.7µF 1.5nF 10nF 71.5k 59k 107k 10µF 22µH 22µH 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 3V TO 60V –12V D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 744873220 C5: MURATA GRM32ER71H106KA12L 150mA AT V IN = 3V 380mA AT V IN = 5V 680mA AT V IN = 12V 860mA AT V IN = 24V 920mA AT V IN = 36V 1A AT V IN = 48V
8361 TA09a
V IN = 5V V IN = 12V V IN = 24V V IN = 48V LOAD CURRENT (A) 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 100 EFFICIENCY (%)
8361 TA09b
450kHz, 10V to 48V Input, –48V Inverting Converter 1µF 4.7µF 2.2nF 10nF 16.9k 48.7k 107k 10µF 47µH 47µH 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 10V TO 48V –48V D1: DIODES INC. B1100LB L1: WURTH ELEKTRONIK 744873470 C5: MURATA GRM32ER71H106KA12L 250mA AT V IN = 10V 290mA AT V IN = 12V 470mA AT V IN = 24V 550mA AT V IN = 36V 610mA AT V IN = 48V
8361 TA10a
V IN = 12V V IN = 24V V IN = 48V LOAD CURRENT (A) 0.1 0.3 0.4 0.5 0.7 100 EFFICIENCY (%)
8361 TA10b
Rev. BFor more information www.analog.com TYPICAL APPLICATIONS 0.1µF 10µF FB2 0.1µF 4.7µF C10 1µF C12 6.8nF C13 0.22µF 71.5k 16.2k 121k C14 10µF L1A 22µH L1B 22µH C15 1µF 68µF 50V 0.1µF 0.1µF 100k V OUT 24V V IN 5V TO 48V INPUT EMI FIL TER OUTPUT EMI FIL TER 50V 50V 35V 0402 35V 0402 50V 1210 EN/UVLO R T V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE D1: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 744877220 L2: WURTH ELEKTRONIK 74479876147 C2: PANASONIC EEHZC1H680P C14: MURATA GRM32ER71H106KA12L FB2: WURTH ELEKTRONIK 742792040 1206 100V 1206 35V 0402 35V 0402 470nH 1210 V OUT
8361 TA11a
(CISPR25 Class 5 Peak) Conducted EMI Performance (CISPR25 Class 5 Average) Radiated EMI Performance (CISPR25 Class 5 Peak) Radiated EMI Performance (CISPR25 Class 5 Average) Low IQ, Low EMI, 400kHz, 24V Output SEPIC Converter with SSFM 12V INPUT TO 24V OUTPUT AT 500mA, f SW = 400kHz, SSFM ON CLASS 5 PEAK LIMIT MEASURED EMISSIONS AMBIENT NOISE FREQUENCY (MHz) 0.1 –20 –10 PEAK CONDUCTED EMI (dBµV)
8361 TA11b
12V INPUT TO 24V OUTPUT AT 500mA, f SW = 400kHz, SSFM ON CLASS 5 AVERAGE LIMIT MEASURED EMISSIONS AMBIENT NOISE FREQUENCY (MHz) 0.1 –40 –30 –20 –10 AVERAGE CONDUCTED EMI (dBµV)
8361 TA11c
12V INPUT TO 24V OUTPUT AT 500mA, f SW = 400kHz, SSFM ON CLASS 5 PEAK LIMIT MEASURED EMISSIONS AMBIENT NOISE FREQUENCY (MHz) 100 200 300 400 500 600 700 800 900 1000 –20 –10 PEAK RADIATED EMI (dBµV/m)
8361 TA11d
12V INPUT TO 24V OUTPUT AT 500mA, f SW = 400kHz, SSFM ON CLASS 5 AVERAGE LIMIT MEASURED EMISSIONS AMBIENT NOISE FREQUENCY (MHz) 100 200 300 400 500 600 700 800 900 1000 –20 –10 AVERAGE RADIATED EMI (dBµV/m)
8361 TA11e
Rev. B For more information www.analog.com PACKAGE DESCRIPTION MSOP (MSE16(12)) 0213 REV D 0.53 ±0.152 (.021 ±.006) SEATING PLANE 0.18 (.007) 1.10 (.043) MAX 0.17 –/uni00A00.27 (.007 – .011) TYP 0.86 (.034) REF 0.50 (.0197) BSC 1.0 (.039) BSC 1.0 (.039) BSC 16 14 121110 1 3 5 6 7 8 1 8 NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.254 (.010) 0° – 6° TYP DETAIL “A” DETAIL “A” GAUGE PLANE 5.10 (.201) MIN 3.20 – 3.45 (.126 – .136) 0.889 ±0.127 (.035 ±.005) RECOMMENDED SOLDER PAD LAYOUT 0.305 ±0.038 (.0120 ±.0015) TYP 0.50 (.0197) BSC BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ±0.102 (.112 ±.004) 2.845 ±0.102 (.112 ±.004) 4.039 ±0.102 (.159 ±.004) (NOTE 3) 1.651 ±0.102 (.065 ±.004) 1.651 ±0.102 (.065 ±.004) 0.1016 ±0.0508 (.004 ±.002) 3.00 ±0.102 (.118 ±.004) (NOTE 4) 0.280 ±0.076 (.011 ±.003) REF 4.90 ±0.152 (.193 ±.006) DETAIL “B” DETAIL “B” CORNER TAIL IS PART OF THE LEADFRAME FEATURE. FOR REFERENCE ONL Y NO MEASUREMENT PURPOSE
0.12 REF
0.35 REF Variation: MSE16 (12) 16-Lead Plastic MSOP with 4 Pins Removed Exposed Die Pad (Reference LTC DWG # 05-08-1871 Rev D)
Rev. BFor more information www.analog.com Information furnished by Analog Devices is believed to be accurate and reliable. However , no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
REVISION HISTORY
REV DATE DESCRIPTION PAGE NUMBER A 09/21 Added automotive versions. Edited to Electrical Characteristics table and Block Diagram. Updated Figure 5. Suggested Boost Converter Layout. Edited equations. Edited component labels. 1, 2 3, 8 21 to 26 B 02/25 Removed Powered by Linear Logo. Updated INTVCC in Pin Functions section.
Rev. B For more information www.analog.com ANALOG DEVICES, INC. 2018-2025 www.analog.com RELATED PARTS TYPICAL APPLICATION PART NUMBER DESCRIPTION COMMENTS LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch VIN = 6V to 100V, Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8330 60V, 1A, Low IQ Boost/SEPIC/Inverting 2MHz Converter VIN = 3V to 40V, VOUT(MAX) = 60V, IQ = 6µA (Burst Mode Operation), 6-Lead TSOT-23, 3mm × 2mm DFN packages LT8331 Low IQ Boost/SEPIC/Flyback/Inverting Converter with 140V/0.5A Switch VIN = 4.5V to 100V, VOUT(MAX)=140V, IQ = 6µA (Burst Mode Operation), MSOP-16(12)E LT8362 60V, 2A, Low IQ Boost/SEPIC/Inverting Converter VIN = 2.8V to 60V, VOUT(MAX) = 60V, IQ = 9µA (Burst Mode Operation), MSOP-16(12)E 3mm × 3mm DFN-8 packages LT8364 60V, 4A, Low IQ Boost/SEPIC/Inverting Converter VIN = 2.8V to 60V, VOUT(MAX) = 60V, IQ = 9µA (Burst Mode Operation), MSOP-16(12)E 4mm × 3mm DFN-12 packages LT8494 70V, 2A Boost/SEPIC 1.5MHz High Efficiency Step-Up DC/DC Converter VIN = 1V to 60V (2.5V to 32V Start-Up), VOUT(MAX) = 70V, IQ = 3µA (Burst Mode Operation), ISD = <1µA, 20-Lead TSSOP LT8570/LT8570-1 65V, 500mA/250mA Boost/Inverting DC/DC Converter VIN(MIN) = 2.55V, VIN(MAX) = 40V, VOUT(MAX) = ±60V, IQ = 1.2mA, ISD = <1mA, 3mm × 3mm DFN-8, MSOP-8E LT8580 1A (ISW), 65V, 1.5MHz, High Efficiency Step-Up DC/DC Converter VIN: 2.55V to 40V, VOUT(MAX) = 65V, IQ = 1.2mA, ISD = <1µA, 3mm × 3mm DFN-8, MSOP-8E Efficiency 450kHz, 4.5V to 12V Input, –150V Output, Automotive LiDAR APD Bias Power Supply 1µF 4.7µF 4.7nF 10nF 5.36k
8361 TA12a
56.2k 107k 1.5µF 2.2µH 2.2µF 20/uni03A9 20/uni03A9 1.5µF 1µF EN/UVLO RT V IN SW FBX BIAS INTV CC SS GND V C L T8361 SYNC/MODE V IN V OUT 4.5V TO 12V –150V D1, D2, D3, D4: DIODES INC. DFLS2100 L1: WURTH ELEKTRONIK 7447779002 C5: NIPPON CHEMI-CON KTS101B155M32N0T00 V OUT 15mA V IN = 5V V IN = 8V V IN = 12V LOAD CURRENT (mA) 100 EFFICIENCY (%)