LT8304

LT8304: 100VIN Micropower No-Opto Isolated Flyback Converter with 150V/2A Switch

Document overview

• Manufacturer or author: Linear Technology Corporation
• PDF pages: 28

Technical content

Rev. BFor more information www.analog.comDocument Feedback TYPICAL APPLICATION FEATURES DESCRIPTION 100VIN Micropower No-Opto Isolated Flyback Converter with 150V/2A Switch The LT®8304/LT8304-1 are monolithic micropower iso - lated flyback converters. By sampling the isolated output voltage directly from the primary-side flyback waveform, the parts require no third winding or opto-isolator for regulation. The output voltage is programmed with two external resistors and a third optional temperature com- pensation resistor. Boundary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 2A, 150V DMOS power switch is integrated along with all the high voltage circuitry and control logic into a thermally enhanced 8-lead SO package. The LT8304/LT8304-1 operate from an input voltage range of 3V to 100V and deliver up to 24W of isolated output power. The high level of integration and the use of boundary and low ripple Burst Mode operation result in a simple to use, low component count , and high efficiency applica - tion solution for isolated power delivery. The LT8304-1 is specially optimized for high step-up output applications. 4V to 80VIN/5VOUT Isolated Flyback Converter

APPLICATIONS

n 3V to 100V Input Voltage Range n 2A, 150V Internal DMOS Power Switch n Low Quiescent Current: n 116µA in Sleep Mode n 390µA in Active Mode n Quasi-Resonant Boundary Mode Operation at Heavy Load n Low Ripple Burst Mode® Operation at Light Load n Minimum Load < 0.5% (Typ) of Full Output n No T ransformer Third Winding or Opto-Isolator Required for Output Voltage Regulation n Accurate EN/UVLO Threshold and Hysteresis n Internal Compensation and Soft-Start n Temperature Compensation for Output Diode n Output Short-Circuit Protection n Thermally Enhanced 8-Lead SO Package n Isolated Automotive, Industrial, Medical, Telecom Power Supplies n Isolated Auxiliary/Housekeeping Power Supplies All registered trademarks and trademarks are the property of their respective owners. Efficiency vs Load Current VIN L T8304 SW 40µH VIN 4V TO 80V 6:1 1.1µH RFB RREF EN/UVLO 220pF 10µF 1µF 100µF 20mA TO 2.4A (VIN = 24V) 20mA TO 3.6A (VIN = 48V) 20mA TO 4.2A (VIN = 72V) VOUT– 100/uni03A9 309k 100k 10k

8304 TA01a

LOAD CURRENT (A) 0.6 1.2 1.8 2.4 3.0 3.6 4.2 100 EFFICIENCY (%)

8304 TA01b

VIN = 24V VIN = 48V VIN = 72V

Rev. B For more information www.analog.com PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS Operating Junction Temperature TJ Range (Notes 3, 4) (Note 1) TOP VIEW TC RREF RFB SW EN/UVLO INTVCC VIN GND S8E PACKAGE 8-LEAD PLASTIC SO GND θJA = 33°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8304ES8E#PBF LT8304ES8E#TRPBF 8304 8-Lead Plastic SO –40°C to 125°C LT8304IS8E#PBF LT8304IS8E#TRPBF 8304 8-Lead Plastic SO –40°C to 125°C LT8304HS8E#PBF LT8304HS8E#TRPBF 8304 8-Lead Plastic SO –40°C to 150°C LT8304ES8E-1#PBF LT8304ES8E-1#TRPBF 83041 8-Lead Plastic SO –40°C to 125°C LT8304IS8E-1#PBF LT8304IS8E-1#TRPBF 83041 8-Lead Plastic SO –40°C to 125°C LT8304HS8E-1#PBF LT8304HS8E-1#TRPBF 83041 8-Lead Plastic SO –40°C to 150°C Consult L TC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container For more information on lead free part marking, go to: www.adi.com/leadfree/ For more information on tape and reel specifications, go to: www.adi.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. https://www.analog.com/en/products/lt8304.

Rev. BFor more information www.analog.com ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN, CINTVCC = 1µF to GND, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT VIN VIN Voltage Range l 3 100 V IQ VIN Quiescent Current VEN/UVLO = 0.2V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 1.8 116 390 3 µA µA µA µA EN/UVLO Shutdown Threshold For Lowest Off IQ l 0.2 0.5 V EN/UVLO Enable Threshold Falling l 1.178 1.214 1.250 V EN/UVLO Enable Hysteresis 14 mV IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.2V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.3 –0.1 2.5 0.1 2.7 0.1 µA µA µA VINTVCC INTVCC Regulation Voltage IINTVCC = 0mA to 10mA 2.8 3 3.1 V IINTVCC INTVCC Current Limit VINTVCC = 2.8V 16 mA INTVCC UVLO Threshold Falling 2.38 2.47 2.56 V INTVCC UVLO Hysteresis 105 mV (RFB – VIN) Voltage IRFB = 75µA to 125µA –60 60 mV RREF Regulation Voltage l 0.98 1.00 1.02 V RREF Regulation Voltage Line Regulation 3V ≤ VIN ≤ 100V 0.02 0.1 % VTC TC Pin Voltage 1.00 V ITC TC Pin Current VTC = 1.2V (LT8304) VTC = 1.2V (LT8304-1) VTC = 0.8V –200 µA µA µA fMAX Maximum Switching Frequency l 315 350 385 kHz fMIN Minimum Switching Frequency 8 11 14 kHz tON(MIN) Minimum Switch-On Time (LT8304) (LT8304-1) 160 950 ns ns ISW(MAX) Maximum Switch Current Limit 2.0 2.4 2.8 A ISW(MIN) Minimum Switch Current Limit 0.43 0.48 0.53 A RDS(ON) Switch On-Resistance ISW = 0.8A 0.5 Ω ILKG Switch Leakage Current VSW = 150V 0.1 0.5 µA tSS Soft-Start Timer 11 ms Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 150V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 150V as shown in Figure 5. Note 3: The LT8304E/LT8304E-1 are guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT8304I/LT8304I-1 are guaranteed over the full –40°C to 125°C operating junction temperature range. LT8304H/LT304H-1 are guaranteed over the full –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125°C. Note 4: The LT8304/LT8304-1 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.

Rev. B For more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Operation Waveforms VIN Shutdown Current VIN Quiescent Current, Sleep Mode VIN Quiescent Current, Active Mode Output Load and Line Regulation Output Temperature Variation Switching Frequency vs Load Current TA = 25°C, unless otherwise noted. LOAD CURRENT (A) 0.6 1.2 1.8 2.4 3.0 3.6 4.2 4.80 4.85 4.90 4.95 5.00 5.05 5.10 5.15 5.20 OUTPUT VOL TAGE (V)

8304 G01

VIN = 24V VIN = 48V VIN = 72V TEMPERATURE (°) –50 –25 100 125 150 4.7 4.8 4.9 5.0 5.1 5.2 5.3 OUTPUT VOL TAGE (V)

8304 G02

VIN = 48V IOUT = 1A RTC = 100k RTC = OPEN LOAD CURRENT (A) 0.6 1.2 1.8 2.4 3.6 4.2 100 200 300 400 500 FREQUENCY (kHz)

8304 G03

VIN = 24V VIN = 48V VIN = 72V VSW 50V/DIV VOUT 50mV/DIV 2µs/DIV FRONT PAGE APPLICATION VIN = 48V IOUT = 3A

8304 G04

2µs/DIV FRONT PAGE APPLICATION VIN = 48V IOUT = 0.5A

8304 G05

20µs/DIV FRONT PAGE APPLICATION VIN = 48V IOUT = 20mA

8304 G06

T J = 150°C T J = 25°C T J = –50°C V IN (V) 100 I Q (µA) IN

8304 G07

T J = 150°C T J = 25°C T J = –50°C V IN (V) 100 100 110 120 130 140 150 I Q (µA) Sleep Mode

8304 G08

T J = 150°C T J = 25°C T J = –50°C V IN (V) 100 350 370 390 410 430 450 I Q (µA) Active Mode

8304 G09

Rev. BFor more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS INTVCC Voltage vs VIN INTVCC UVLO Threshold (RFB – VIN) Voltage RREF Regulation Voltage RREF Line Regulation TC Pin Voltage EN/UVLO Enable Threshold EN/UVLO Hysteresis Current INTVCC Voltage vs Temperature TA = 25°C, unless otherwise noted. RISING FALLING TEMPERATURE (°C) –50 –25 100 125 150 1.200 1.205 1.210 1.215 1.220 1.225 1.230 1.235 1.240 V EN/UVLO (V) EN/UVLO Enable Threshold

8304 G10

TEMPERATURE (°C) –50 –25 100 125 150 I HYST (µA) EN/UVLO Hysteresis Current

8304 G11

I INTVCC = 0mA I INTVCC = 10mA TEMPERATURE (°C) –50 –25 100 125 150 2.80 2.85 2.90 2.95 3.00 3.05 3.10 V INTVCC (V) CC

8304 G12

I INTVCC = 0mA I INTVCC = 10mA V IN (V) 100 2.80 2.85 2.90 2.95 3.00 3.05 3.10 V INTVCC (V) CC IN

8304 G13

TEMPERATURE (°C) –50 –25 100 125 150 2.2 2.3 2.4 2.5 2.6 2.7 2.8 V INTVCC (V) INTV CC UVLO Threshold

8304 G14

I RFB = 125µA I RFB = 100µA I RFB = 75µA TEMPERATURE (°C) –50 –25 100 125 150 –40 –30 –20 –10 VOL TAGE (mV) FB - V IN ) Voltage

8304 G15

TEMPERATURE (°C) –50 –25 100 125 150 0.990 0.992 0.994 0.996 0.998 1.000 1.002 1.004 1.006 1.008 1.010 V RREF (V) R REF Regulation Voltage

8304 G16

V IN (V) 100 0.990 0.992 0.994 0.996 0.998 1.000 1.002 1.004 1.006 1.008 1.010 V RREF (V) R REF Line Regulation

8304 G17

TEMPERATURE (°C) –50 –25 100 125 150 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 V TC (V) TC Pin Voltage

8304 G18

Rev. B For more information www.analog.com Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time RDS(ON) Switch Current Limit Maximum Switching Frequency TYPICAL PERFORMANCE CHARACTERISTICSTA = 25°C, unless otherwise noted. TEMPERATURE (°C) –50 –25 100 125 150 0.2 0.4 0.6 0.8 1.0 1.2 RESISTANCE (Ω) DS(ON)

8304 G19

TEMPERATURE (°C) –50 –25 100 125 150 0.5 1.0 1.5 2.0 2.5 3.0 ISW (A)

8304 G20

TEMPERATURE (°C) –50 –25 100 125 150 100 200 300 400 500 FREQUENCY (kHz)

8304 G21

TEMPERATURE (°C) –50 –25 100 125 150 FREQUENCY (kHz)

8304 G22

TEMPERATURE (°C) –50 –25 100 125 150 100 200 300 400 TIME (ns)

8304 G23

TEMPERATURE (°C) –50 –25 100 125 150 100 200 300 400 500 TIME (ns)

8304 G24

Rev. BFor more information www.analog.com PIN FUNCTIONS EN/UVLO (Pin 1 ): Enable /Undervoltage Lockout . The EN/UVLO pin is used to enable the LT8304. Pull the pin below 0.2V to shut down the LT8304. This pin has an ac- curate 1.214V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resistor divider from V IN to ground . A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. INTVCC (Pin 2): Internal 3V Linear Regulator Output. The INTVCC pin is supplied from VIN and powers the internal control circuitry and gate driver . Do not overdrive the INTVCC pin with any external supply, such as a third winding supply. Locally bypass this pin to ground with a minimum 1µF ceramic capacitor. VIN (Pin 3): Input Supply. The VIN pin supplies current to the internal circuitry and serves as a reference voltage for the feedback circuitry connected to the R FB pin. Locally bypass this pin to ground with a capacitor. GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed pad provides both electrical contact to ground and good thermal contact to the printed circuit board . Solder the exposed pad directly to the ground plane. SW (Pin 5): Drain of the Internal DMOS Power Switch . Minimize trace area at this pin to reduce EMI and voltage spikes. RFB (Pin 6 ): Input Pin for External Feedback Resistor . Connect a resistor from this pin to the transformer primary SW pin. The ratio of the RFB resistor to the RREF resistor, times the internal voltage reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. RREF (Pin 7): Input Pin for External Ground Referred Ref- erence Resistor. The resistor at this pin should be in the range of 10k, but for convenience in selecting a resistor divider ratio, the value may range from 9.09k to 11.0k. TC (Pin 8): Output Voltage Temperature Compensation. The voltage at this pin is proportional to absolute temperature (PTAT) with temperature coefficient equal to 3.35mV/°C, i.e., equal to 1V at room temperature 25 °C. The TC pin voltage can be used to estimate the LT8304 junction tem- perature. Connect a resistor from this pin to the RREF pin to compensate the output diode temperature coefficient.

Rev. B For more information www.analog.com BLOCK DIAGRAM OPERATION DRIVER INTVCC VIN N:1 A2 RSENSE TC 8304 BD RREF RREF REN2 REN1 RTC RFB gm 1.214V OSCILLATOR LDO BOUNDARY DETECTOR START-UP , REFERENCE, CONTROL PTAT VOL TAGE R M1 GND 4, EXPOSED PAD PIN 9 Q S VIN VIN CIN RFB SW L1A L1B COUT DOUT VOUT+ VOUT– 25µA INTVCC

1 EN/UVLO

2.5µA 1:4 – + The LT8304 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation bound- ary. Opto-isolator circuits waste output power , and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unit- to-unit variation and aging over lifetime. Circuits employing extra transformer windings also exhibit deficiencies , as using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT8304 samples the isolated output voltage through the primary-side flyback pulse waveform. In this manner, neither opto-isolator nor extra transformer winding is re- quired for regulation. Since the LT8304 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method im- proves load regulation without the need of external load compensation components.

Rev. BFor more information www.analog.com OPERATION The LT8304 is a simple to use micropower isolated fly - back converter housed in a thermally enhanced 8-lead SO package. The output voltage is programmed with two external resistors. An optional TC resistor provides easy output diode temperature compensation . By integrating the loop compensation and soft-start inside , the part reduces the number of external components . As shown in the Block Diagram , many of the blocks are similar to those found in traditional switching regulators including reference, regulators, oscillator, logic, current amplifier, current comparator, driver, and power switch. The novel sections include a flyback pulse sense circuit, a sample- and-hold error amplifier, and a boundary mode detector, as well as the additional logic for boundary conduction mode, discontinuous conduction mode , and low ripple Burst Mode operation. Quasi-Resonant Boundary Mode Operation The LT8304 features quasi-resonant boundary conduction mode operation at heavy load, where the chip turns on the primary power switch when the secondary current is zero and the SW rings to its valley. Boundary conduction mode is a variable frequency , variable peak-current switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero , the SW pin voltage collapses and rings around VIN. A boundary mode detector senses this event and turns the power switch back on at its valley. Boundary conduction mode returns the secondary current to zero every cycle , so parasitic resistive voltage drops do not cause load regulation errors . Boundary conduc - tion mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit subharmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter , boundary conduction mode in - creases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switching and gate charge losses. To avoid this scenario, the LT8304 has an additional internal oscillator, which clamps the maximum switching frequency to be less than 350kHz (TYP). Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters , the LT8304 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch cur- rent limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8304 starts to fold back the switching frequency while keeping the minimum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sample-and- hold error amplifier. Meanwhile, the part switches between sleep mode and active mode, thereby reducing the effec- tive quiescent current to improve light load efficiency. In this condition, the LT8304 runs in low ripple Burst Mode operation. The typical 11kHz minimum switching frequency determines how often the output voltage is sampled and also the minimum load requirement. High Step-Up VOUT Applications Typically, high step-up output applications have excessive primary inductor current ringing during primary switch turn-on due to the huge reflected capacitance on SW node. Such current ringing can falsely trigger LT8304 current comparator after 160ns typical blanking time and create large signal oscillation , especially at high V IN and light load condition . The LT8304-1, specially optimized for high step-up output applications, is more immune to the current ringing without requiring longer blanking time . For any 1 :N step-up transformer turns ratio larger than or equal to 5, the LT8304-1 is recommended.

Rev. B For more information www.analog.com APPLICATIONS INFORMATION Output Voltage The RFB and RREF resistors as depicted in the Block Diagram are external resistors used to program the output voltage. The LT8304 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier, which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the V IN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: VFLBK = (VOUT + VF + ISEC • ESR) • NPS VF = Output diode forward voltage ISEC = T ransformer secondary current ESR = Total impedance of secondary circuit NPS = T ransformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current , IRFB, by the R FB resistor and the flyback pulse sense circuit (M2 and M3). This current, IRFB, also flows through the RREF resistor to generate a ground-referred voltage. The resulting voltage feeds to the inverting input of the sample- and-hold error amplifier. Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the VFLBK equation can be assumed to be zero. The internal reference voltage, VREF, 1.00V, feeds to the noninverting input of the sample-and-hold error ampli - fier. The relatively high gain in the overall loop causes the voltage at the RREF pin to be nearly equal to the internal reference voltage VREF. The resulting relationship between VFLBK and VREF can be expressed as: VFLBK RFB   •RREF =VREF or VFLBK =VREF • RFB RREF   VREF = Internal reference voltage 1.00V Combination with the previous V FLBK equation yields an equation for VOUT, in terms of the RFB and RREF resistors, transformer turns ratio, and diode forward voltage: VOUT =VREF • RFB RREF   • 1 NPS   – VF Output Temperature Compensation The first term in the VOUT equation does not have tempera- ture dependence, but the output diode forward voltage, VF, has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coefficient pro- duces approximately 200mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs , such as 12V and 24V, the output diode temperature coefficient has a negligible ef - fect on the output voltage regulation . For lower voltage outputs, such as 3.3V and 5V, however, the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. The LT8304 junction temperature usually tracks the output diode junction temperature to the first order. To compensate the negative temperature coefficient of the output diode, a resistor, RTC, connected between the TC and RREF pins generates a proportional-to-absolute-temperature (PTAT) current. The PTAT current is zero at 25°C, flows into the RREF pin at hot temperature, and flows out of the RREF pin at cold temperature . With the R TC resistor in place , the output voltage equation is revised as follows: VOUT = VREF • RFB RREF

• 1 NPS – VF TO( ) – VTC / T( ) • T –TO( ) • RFB RTC
• 1 NPS – VF / T( ) • T–TO( ) TO=Room temperature 25° C VF / T( ) = Output diode forward voltage temperature coefficient VTC / T( ) = 3.35mV/ C

Rev. BFor more information www.analog.com APPLICATIONS INFORMATION To cancel the output diode temperature coefficient , the following two equations should be satisfied: VOUT = VREF • RFB RREF

• 1 NPS – VF TO( ) VTC/ T( ) • RFB RTC
• 1 NPS = – VF/ T( ) Selecting Actual RREF, RFB, RTC Resistor Values The LT8304 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature , the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the R FB and R TC resistor values . Therefore, a simple 2-step sequential process is recommended for selecting resistor values. Rearrangement of the expression for VOUT in the previous sections yields the starting value for RFB: RFB = RREF •NPS • VOUT + VF TO( )( ) VREF VOUT = Output voltage VF (TO) = Output diode forward voltage at 25°C = ~0.3V NPS = T ransformer effective primary-to-secondary turns ratio The equation shows that the RFB resistor value is indepen- dent of the RTC resistor value. Any RTC resistor connected between the TC and RREF pins has no effect on the output voltage setting at 25°C because the TC pin voltage is equal to the RREF regulation voltage at 25°C. The R REF resistor value should be approximately 10k because the LT8304 is trimmed and specified using this value. If the RREF resistor value varies considerably from 10k, additional errors will result. However, a variation in RREF up to 10% is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB/RREF ratios. First, build and power up the application with the starting RREF, RFB values (no RTC resistor yet) and other compo- nents connected, and measure the regulated output volt- age, VOUT(MEAS). The new RFB value can be adjusted to: RFB(NEW) = VOUT VOUT(MEAS)
• RFB Second, with a new RFB resistor value selected, the output diode temperature coefficient in the application can be tested to determine the R TC value. Still without the R TC resistor, the VOUT should be measured over temperature at a desired target output load . It is very important for this evaluation that uniform temperature be applied to both the output diode and the LT8304. If freeze spray or a heat gun is used , there can be a significant mismatch in temperature between the two devices that causes sig- nificant error. Attempting to extrapolate the data from a diode data sheet is another option if there is no method to apply uniform heating or cooling such as an oven. With at least two data points spreading across the operating temperature range, the output diode temperature coef - ficient can be determined by: – δVF /δT( )=VOUT T1( )– VOUT T2( ) T1– T2 Using the measured output diode temperature coefficient, an exact R TC value can be selected with the following equation: RTC = δVTC/δT( ) – δVF /δT( )
• RFB NPS   Once the RREF, RFB, and RTC values are selected, the regula- tion accuracy from board to board for a given application will be very consistent, typically under ±5% when includ- ing device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ±1%). However , if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT.

output current at the expense of a higher switch voltage. the amount of output power at given input voltages. Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output Figure 3. Output Power for 12V Output Figure 4. Output Power for 24V Output

8304 F02

8304 F03

20 N = 3:1

8304 F04

20 N = 3:2

8304 F01

and may cause instability at light load. information should be carefully considered. shows the details of these transformers. Table 1. Predesigned T ransformers – Typical Specifications

Rev. B For more information www.analog.com APPLICATIONS INFORMATION Turns Ratio Note that when choosing an RFB/RREF resistor ratio to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application . In contrast, the use of simple ratios of small integers , e.g., 3:1, 2:1, 1:1, etc., provides more freedom in settling total turns and mutual inductance. Typically, choose the transformer turns ratio to maximize available output power . For low output voltages (3.3V or 5V), a N :1 turns ratio can be used with multiple pri - mary windings relative to the secondary to maximize the transformer’s current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 150V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch . To- gether these conditions place an upper limit on the turns ratio, NPS, for a given application . Choose a turns ratio low enough to ensure NPS < 150V – VIN(MAX) – VLEAKAGE VOUT + VF For larger N:1 step-down turns ratio, choose a transformer with a larger physical size to deliver additional current. In addition, choose a large enough inductance value to en - sure that the switch-off time is long enough to accurately sample the output voltage. Always choose the LT8304 for N:1 step-down transformer turns ratio. For lower output power levels or higher output voltage , choose a 1:1 or 1:N step-up transformer for the absolute smallest transformer size. A 1:N step-up transformer will minimize the magnetizing inductance and size, but will also limit the available output power. A higher 1:N step-up turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch. For any 1:N step-up transformer turns ratio larger than or equal to 5, the LT8304-1 is recommended. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer speci- fies turns ratio accuracy within ±1%. Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8304, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output voltage regulation will be maintained independent of winding re- sistance due to the boundary /discontinuous conduction mode operation of the LT8304. Leakage Inductance and Snubbers T ransformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored en- ergy must be dissipated. It is very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus V IN should be kept below 110V. This leaves at least 40V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound trans - formers or for excessive leakage inductance.

Figure 5. Maximum Voltages for SW Pin Flyback Waveform the boundary mode detector for approximately 250ns. should be limited to less than 250ns. (RC + DZ) snubber circuit in Figure 6 is recommended. SW pin from exceeding its 150V absolute maximum rating. Figure 6. (RC + DZ) Snubber Circuit

8304 F06

converted to heat and will not be delivered to the load . tor in series with a 100Ω resistor is a good starting point. voltage rating higher than the maximum SW pin voltage.

LT8304 in shutdown with quiescent current less than 3µA.

8304 F07

Figure 7. Undervoltage Lockout (UVLO) minimum amount of time and with a minimum frequency. higher minimum load, typically 2% of full load.

input range from 36V to 75V. Step 1: Select the transformer turns ratio. different transformer turns ratio. Table 2. Switch Voltage Stress and Output Current Capability vs Step 2: Determine the primary inductance. 60% larger than the minimum values calculated above . LPRI = 40µH is then chosen in this example. saturation current level across line and load conditions . chosen as the flyback transformer.

Rev. B For more information www.analog.com APPLICATIONS INFORMATION Step 3: Choose the output diode. T wo main criteria for choosing the output diode include forward current rating and reverse-voltage rating . The maximum load requirement is a good first-order guess at the average current requirement for the output diode . Under output short-circuit condition , the output diode needs to conduct much higher current. Therefore, a con- servative metric is 60 % of the maximum switch current limit multiplied by the turns ratio: IDIODE(MAX) = 0.6 • ISW(MAX) • NPS Example: IDIODE(MAX) = 8.6A Next calculate reverse voltage requirement using maxi - mum VIN: VREVERSE = VOUT + VIN(MAX) NPS Example: VREVERSE = 5V + 75V 6 =17.5V The PDS835L (8A, 35V diode) from Diodes Inc. is chosen. Step 4: Choose the output capacitor. The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. Use the following equation to calculate the output capacitance: COUT = LPRI •ISW2 2• VOUT •∆VOUT Example: Design for output voltage ripple less than ±1% of VOUT, i.e., 100mV. COUT = 40µH• 2.4A( )2 2•5V •0.1V = 230µF Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop up to 40 % of quoted capacitance at the maximum voltage rating . So three 100µF, 10V rating ceramic capacitors are chosen. Step 5: Design snubber circuit. The snubber circuit protects the power switch from leak- age inductance voltage spike . A (RC + DZ ) snubber is recommended for this application. A 220pF capacitor in series with a 100Ω resistor is chosen as the RC snubber. The maximum Zener breakdown voltage is set according to the maximum VIN: VZENNER(MAX) ≤ 145V – VIN(MAX) Example: VZENNER(MAX) ≤ 145V – 75V = 70V A 62V Zener with a maximum of 65V will provide optimal protection and minimize power loss. So a 62V, 1.5W Zener from Central Semiconductor (CMZ5944B) is chosen. Choose a diode that is fast and has sufficient reverse voltage breakdown: VREVERSE > VSW(MAX) VSW(MAX) = VIN(MAX) + VZENNER(MAX) Example: VREVERSE > 150V A 150V, 1A diode from Diodes Inc. (DFLS1150) is chosen. Step 6: Select the RREF and RFB resistors. Use the following equation to calculate the starting values for RREF and RFB: RFB = RREF •NPS • VOUT + VF TO( )( ) VREF RREF = 10k Example: 1.00V = 318k For 1% standard values, a 316k resistor is chosen.

Rev. BFor more information www.analog.com APPLICATIONS INFORMATION Step 7: Adjust RFB resistor based on output voltage. Build and power up the application with application com- ponents and measure the regulated output voltage. Adjust RFB resistor based on the measured output voltage: RFB(NEW) = VOUT VOUT(MEASURED)

• RFB Example: RFB = 5V 5.11V •316k = 309k Step 8 : Select R TC resistor based on output voltage temperature variation. Measure output voltage in a controlled temperature envi- ronment like an oven to determine the output temperature coefficient. Measure output voltage at a consistent load current and input voltage, across the operating tempera- ture range. Calculate the temperature coefficient of VF: – δVF /δT( )=VOUT T1( )– VOUT T2( ) T1– T2 RTC =3.35mV/°C – δVF /δT( )
• RFB NPS   Example: – δVF /δT( )=5.149V – 4.977V 100°C– 0°C( ) =1.72mV / °C RTC =3.35mV/°C 1.72mV/°C • 309  =100k Step 9: Select the EN/UVLO resistors. Determine the amount of hysteresis required and calculate R1 resistor value: VIN(HYS) = 2.5µA • R1 Example: Choose 2.5V of hysteresis, R1 = 1M Determine the UVLO thresholds and calculate R2 resistor value: VIN(UVLO+)=1.228V • R1+ R2( ) R2 + 2.5µA •R1 Example: Set VIN UVLO rising threshold to 34.5V: R2 = 40.2k VIN(UVLO+) = 34.3V VIN(UNLO–) = 31.4V Step 10: Ensure minimum load. The theoretical minimum load can be approximately estimated as: ILOAD(MIN) = 40µH• 0.53A( )2 •14kHz 2•5V =15.7mA Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the con - verter delivers more energy than what is consumed at the output. The real minimum load for this application is about 20mA. In this example, a 249Ω resistor is selected as the minimum load.

Rev. B For more information www.analog.com TYPICAL APPLICATIONS 18V to 80VIN/3.3VOUT Isolated Flyback Converter 18V to 80VIN/5VOUT Isolated Flyback Converter VIN L T8304 SW 40µH VIN 18V TO 80V 8:1 0.63µH RFB RREF 220pFZ1 10µF 1µF 330µF VOUT+ 3.3V 25mA TO 3.4A (VIN = 24V) 25mA TO 4.8A (VIN = 48V) 25mA TO 5.6A (VIN = 72V) VOUT– 100/uni03A9 274k 100k 88.7k 10k D1: DIODES DFLS1150 D2: DIODES SBR15U30SP5 T1: SUMIDA 13324-T083 Z1: CENTRAL CMZ5944B

8304 TA02aTC

40µH VIN 18V TO 80V 6:1 1.1µH RFB RREF 220pFZ1 10µF 1µF 100µF VOUT+ 20mA TO 2.4A (VIN = 24V) 20mA TO 3.6A (VIN = 48V) 20mA TO 4.2A (VIN = 72V) VOUT– 100/uni03A9 309k 100k 88.7k 10k D1: DIODES DFLS1150 D2: DIODES PDS835L T1: WURTH 750315125 Z1: CENTRAL CMZ5944B

8304 TA03aTC

LOAD CURRENT (A) 0.8 1.6 2.4 3.2 4.0 4.8 5.6 100 EFFICIENCY (%)

8304 TA02b

VIN = 24V VIN = 48V VIN = 72V LOAD CURRENT (A) 0.6 1.2 1.8 2.4 3.0 3.6 4.2 100 EFFICIENCY (%)

8304 TA03b

VIN = 24V VIN = 48V VIN = 72V Efficiency vs Load Current Efficiency vs Load Current

Rev. BFor more information www.analog.com TYPICAL APPLICATIONS 18V to 80VIN/12VOUT Isolated Flyback Converter 18V to 80VIN/24VOUT Isolated Flyback Converter VIN L T8304 SW 40µH VIN 18V TO 80V 2:1 10µH RFB RREF 220pFZ1 10µF 1µF 47µF VOUT+ 12V 10mA TO 1.0A (VIN = 24V) 10mA TO 1.4A (VIN = 48V) 10mA TO 1.6A (VIN = 72V) VOUT– 100/uni03A9 237k OPEN 88.7k 10k D1: DIODES DFLS1150 D2: DIODES PMEG6030EP T1: WURTH 750315126 Z1: CENTRAL CMZ5944B

8304 TA04aTC

40µH VIN 18V TO 80V 1:1 40µH RFB RREF 220pFZ1 10µF 1µF 10µF VOUT+ 24V 5mA TO 0.5A (VIN = 24V) 5mA TO 0.7A (VIN = 48V) 5mA TO 0.8A (VIN = 72V) VOUT– 100/uni03A9 237k OPEN 88.7k 10k D1: DIODES DFLS1150 D2: DIODES SBR2U150SA T1: SUMIDA 13324-T084 Z1: CENTRAL CMZ5944B

8304 TA05aTC

LOAD CURRENT (A) 0.4 0.8 1.2 1.6 100 EFFICIENCY (%)

8304 TA04b

VIN = 24V VIN = 48V VIN = 72V LOAD CURRENT (A) 0.1 0.2 0.3 0.4 0.5 0.6 0.8 0.7 100 EFFICIENCY (%)

8304 TA05b

VIN = 24V VIN = 48V VIN = 72V Efficiency vs Load Current Efficiency vs Load Current

Rev. B For more information www.analog.com 18V to 80VIN/48VOUT Isolated Flyback Converter VIN L T8304 SW 40µH VIN 18V TO 80V 1:2 160µH RFB RREF 220pFZ1 10µF 1µF 2.2µF VOUT+ 48V 2mA TO 0.24A (VIN = 24V) 2mA TO 0.34A (VIN = 48V) 2mA TO 0.40A (VIN = 72V) VOUT– 100/uni03A9 232k OPEN 88.7k 10k D1: DIODES DFLS1150 D2: DIODES SBR1U400P1 T1: SUMIDA 13324-T085 Z1: CENTRAL CMZ5944B

8304 TA06aTC

LOAD CURRENT (A) 0.1 0.2 0.3 0.4 100 EFFICIENCY (%)

8304 TA06b

VIN = 24V VIN = 48V VIN = 72V Efficiency vs Load Current TYPICAL APPLICATIONS

Rev. BFor more information www.analog.com Efficiency, VOUT = 200V Load Regulation, VOUT = 200V 4V to 36VIN/200VOUT Isolated Flyback Converter VIN L T8304-1 SW 40µH VIN 4V TO 36V 1:5 1mH RFB RREF 10µF 50V 1µF 6.3V 0.33µF 250V VOUT+ 200V 0.3mA TO 12mA (VIN = 4V) 0.5mA TO 35mA (VIN = 12V) 1.5mA TO 75mA (VIN = 36V) VOUT– 392k 10k 10pF T1: SUMIDA 13324-T086 D1: CENTRAL CMMR1U-06 TR

8304 TA07a

V IN = 4V V IN = 12V V IN = 36V LOAD CURRENT (mA) EFFICIENCY (%) Efficiency, V OUT = 200V

8304 TA07b

V IN = 4V V IN = 12V V IN = 36V LOAD CURRENT (mA) 190 195 200 205 210 OUTPUT VOL TAGE (V) Load Regulation, V OUT = 200V

8304 TA07c

Rev. B For more information www.analog.com Efficiency, VOUT = 400V Load Regulation, VOUT = 400V 4V to 18VIN/400VOUT Isolated Flyback Converter VIN L T8304-1 SW 40µH VIN 4V TO 18V 1:10 4mH RFB RREF 10µF 25V 1µF 6.3V 0.15µF 600V VOUT+ 400V 0.4mA TO 6mA (VIN = 4V) 0.4mA TO 20mA (VIN = 12V) 0.4mA TO 30mA (VIN = 18V) VOUT– 392k 10k 10pF T1: SUMIDA 13324-T087 D1: CENTRAL CMMR1U-06 TR

8304 TA08a

V IN = 4V V IN = 12V V IN = 18V LOAD CURRENT (mA) EFFICIENCY (%) OUT

8304 TA08b

V IN = 4V V IN = 12V V IN = 18V LOAD CURRENT (mA) 380 390 400 410 420 OUTPUT VOL TAGE (V) Load Regulation, V OUT = 400V

8304 TA08c

Rev. BFor more information www.analog.com –18V to –80VIN/–12VOUT Negative Buck Converter Efficiency vs Load Current Efficiency vs Load Current –4V to –80VIN/12VOUT Buck-Boost Converter TYPICAL APPLICATIONS VIN SW L T8304 33µH D1 GND RFB RREF EN/UVLO 47µF D1: DIODES SBR2U150SA L1: WURTH 744771133 Z1: CENTRAL CMHZ5243BC2 1µF VIN –4V TO –80V 10µF 10k

8304 TA09a

+12V 5mA TO 0.25A (VIN = –5V) 5mA TO 0.7A (VIN = –24V) 5mA TO 0.8A (VIN = –48V) 5mA TO 0.9A (VIN = –72V)R4 121k INTVCC LOAD CURRENT (A) 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.9 0.8 100 EFFICIENCY (%)

8304 TA09b

VIN = –5V VIN = –24V VIN = –48V VIN = –72V VIN L T8304 33µH VOUT –12V 10mA TO 1A D1 Z1 SW RREF EN/UVLO RFBEN/UVLO D1: DIODES SBR2U150SA L1: WURTH 744771133 Z1: CENTRAL CMHZ5243B 1µFVIN –18V TO –80V 10µF 47µF 10k

8304 TA10a

71.5k 806k INTVCC LOAD CURRENT (A) 0.2 0.4 0.6 0.8 1.0 100 EFFICIENCY (%)

8304 TA10b

VIN = –24V VIN = –48V VIN = –72V

Rev. B For more information www.analog.com PACKAGE DESCRIPTION .016 – .050 (0.406 – 1.270) .010 – .020 0°– 8° TYP .008 – .010 (0.203 – 0.254) S8E 1015 REV C .053 – .069 (1.346 – 1.752) .014 – .019 (0.355 – 0.483) TYP .004 – .010 (0.101 – 0.254) 0.0 – 0.005 (0.0 – 0.130) .080 – .099 (2.032 – 2.530) .118 – .139 (2.997 – 3.550) .050 (1.270) BSC 1 2 3 4 .150 – .157 (3.810 – 3.988) NOTE 3 8 7 .005 (0.13) MAX 6 5 .189 – .197 (4.801 – 5.004) NOTE 3 .228 – .244 (5.791 – 6.197) .160 ±.005 (4.06 ±0.127) .118 (2.99) REF RECOMMENDED SOLDER PAD LAYOUT .045 ±.005 (1.143 ±0.127) .050 (1.27) BSC INCHES (MILLIMETERS) NOTE: 1. DIMENSIONS IN 2. DRAWING NOT TO SCALE 3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010" (0.254mm) 4. STANDARD LEAD STANDOFF IS 4mils TO 10mils (DATE CODE BEFORE 542) 5. LOWER LEAD STANDOFF IS 0mils TO 5mils (DATE CODE AFTER 542) 8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad (Reference LTC DWG # 05-08-1857 Rev C) .089 (2.26) REF .030 ±.005 (0.76 ±0.127) TYP .245 (6.22) MIN 4 5 Please refer to www.analog.com/en/products/lt8304.html#packaging for the most recent package drawings.

Rev. BFor more information www.analog.com Information furnished by Analog Devices is believed to be accurate and reliable . However , no responsibility is assumed by Analog Devices for its use , nor for any infringements of patents or other rights of third parties that may result from its use . Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices .

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER A 02/17 Added LT8304-1 and H-Grade options Changed TC Pin Current conditions Changed TC pin description to °C Added High Step-Up VOUT Applications section Updated Predesigned T ransformers – Typical Specifications table Revised Turns Ratio section Added new application circuits and graphs All 23, 24 B 04/24 Updated Order Information table 2

Rev. B For more information www.analog.com  ANALOG DEVICES, INC. 2017–2024 www.analog.com RELATED PARTS TYPICAL APPLICATION 4V to 100VIN/140VOUT Boost Converter Efficiency vs Load Current PART NUMBER DESCRIPTION COMMENTS LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6A Switch Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E LT8303 100VIN Micropower Isolated Flyback Converter with 150V/450mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 LT3573/LT3574 LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12) LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components LT3757A/LT3759/ LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/80V Boost/Flyback Converters Monolithic with Integrated 5A/3.3A Switch VIN SW L T8304 150µH D1 GND RFB RREF EN/UVLO C3 1µF D1: DIODES DFLS1200 L1: COILCRAFT DS5022P-154MLB Z1, Z2: CENTRAL CMHZ5207B 1µF VIN 4V TO 100V 10µF 3.57k

8304 TA11a

1.5mA TO 25mA (VIN = 5V) 2mA TO 300mA (VIN = 48V) 7mA TO 700mA (VIN = 100V) 499k INTVCC LOAD CURRENT (mA) 100 1000 100 EFFICIENCY (%)

8304 TA11b

VIN = 5V VIN = 48V VIN = 100V