LT8303

LT8303 (Rev. B)

Document overview

• Manufacturer or author: Analog Devices Inc.
• PDF pages: 24

Technical content

Rev. BFor more information www.analog.com TYPICAL APPLICATION FEATURES DESCRIPTION 100VIN Micropower Isolated Flyback Converter with 150V/450mA Switch The LT®8303 is a micropower high voltage isolated flyback converter . By sampling the isolated output voltage directly from the primary-side flyback waveform, the part requires no third winding or opto-isolator for regulation. The output voltage is programmed with a single external resistor . Internal compensation and soft-start further reduce external component count. Boundary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 450mA, 150V DMOS power switch is integrated along with all high voltage circuitry and control logic into a 5-lead ThinSOT™ package. The LT8303 operates from an input voltages range of 5.5V to 100V and can deliver up to 5W of isolated output power . The high level of integration and the use of boundary mode and low ripple Burst Mode operations result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. L, L T , L TC, L TM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, and 7471522. 6V to 80VIN, 5VOUT Isolated Flyback Converter

APPLICATIONS

n 5.5V to 100V Input Voltage Range n 450mA, 150V Internal DMOS Power Switch n Up to 5W of Output Power n Low Quiescent Current: 70µA in Sleep Mode 280µA in Active Mode n Boundary Mode Operation at Heavy Load n Low Ripple Burst Mode® Operation at Light Load n Minimum Load <0.5% (Typ) of Full Output n VOUT Set with a Single External Resistor n No T ransformer Third Winding or Opto-Isolator Required for Regulation n Accurate EN/UVLO Threshold and Hysteresis n Internal Compensation and Soft-Start n 5-Lead TSOT-23 Package AEC-Q100 Qualified for Automotive Applications n Isolated Telecom, Datacom, Automotive, Industrial, and Medical Power Supplies n Isolated Auxiliary/Housekeeping Power Supplies Efficiency vs Load Current V IN = 12V V IN = 24V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 200 300 400 500 600 700 800 900 100 EFFICIENCY (%)

8303 TA01b

6:1 RFB SW 150µH 4.2µH EN/UVLO 4.7µF VIN VIN 6V TO 80V VOUT+ VOUT– GND 316k 2.5mA TO 0.33A (VIN = 12V) 2.5mA TO 0.52A (VIN = 24V) 2.5mA TO 0.73A (VIN = 48V) 2.5mA TO 0.84A (VIN = 72V) 100µF

8303 TA01a

Rev. B For more information www.analog.com PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS Operating Junction Temperature Range (Notes 3, 4) (Note 1) EN/UVLO 1 GND 2 TOP VIEW S5 PACKAGE 5-LEAD PLASTIC TSOT-23 RFB 3

5 VIN

θJA = 215°C/W ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8303ES5#TRMPBF LT8303ES5#TRPBF L TGXH 5-Lead Plastic TSOT-23 –40°C to 125°C LT8303IS5#TRMPBF LT8303IS5#TRPBF L TGXH 5-Lead Plastic TSOT-23 –40°C to 125°C LT8303HS5#TRMPBF LT8303HS5#TRPBF L TGXH 5-Lead Plastic TSOT-23 –40°C to 150°C AUTOMOTIVE PRODUCTS** LT8303RS5#WTRMPBF LT8303RS5#WTRPBF L TGXH 5-Lead Plastic TSOT-23 –40°C to 150°C LT8303HS5#WTRMPBF LT8303HS5#WTRPBF L TGXH 5-Lead Plastic TSOT-23 –40°C to 150°C Consult L TC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container . For more information on lead free part marking, go to: http://www.linear .com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear .com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. **Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for these models. http://www.linear .com/product/LT8303#orderinfo

Rev. BFor more information www.analog.com

ELECTRICAL CHARACTERISTICS

Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 150V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 150V as shown in Figure 5. Note 3: The LT8303E is guaranteed to meet performance specifications from 0°C to 125°C operating junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT VIN Input Voltage Range 5.5 100 V VIN UVLO Threshold Rising Falling 5.3 3.2 5.5 V V IQ VIN Quiescent Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 1.5 200 280 2.5 µA µA µA µA EN/UVLO Shutdown Threshold For Lowest Off IQ l 0.3 0.75 V EN/UVLO Enable Threshold Falling, E-Grade/I-Grade/H-Grade Falling, R-Grade Hysteresis l l 1.186 1.161 1.223 1.223 0.016 1.284 1.284 V V V IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.1 –0.1 2.5 0.1 2.9 0.1 µA µA µA fMAX Maximum Switching Frequency 320 350 380 kHz fMIN Minimum Switching Frequency 5 7 9 kHz tON(MIN) Minimum Switch-On Time 160 ns tOFF(MIN) Minimum Switch-Off Time VIN = VEN/UVLO = 12V 350 ns tOFF(MAX) Maximum Switch-Off Time Backup Timer 200 µs ISW(MAX) Maximum SW Current Limit 450 535 620 mA ISW(MIN) Minimum SW Current Limit 70 105 140 mA SW Over Current Limit To Initiate Soft-Start 1 A RDS(ON) Switch On-Resistance ISW = 100mA 3.2 Ω ILKG Switch Leakage Current VIN = 100V, VSW = 150V 0.1 0.5 µA IRFB RFB Regulation Current l 97.5 100 102.5 µA RFB Regulation Current Line Regulation 5.5V ≤ VIN ≤ 100V 0.001 0.01 %/V design, characterization and correlation with statistical process controls. The LT8303I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8303R/LT8303H are guaranteed over the full –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 4: The LT8303 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.

Rev. B For more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms VIN Shutdown Current VIN Quiescent Current, Sleep Mode VIN Quiescent Current, Active Mode Output Load and Line Regulation Output Temperature Variation Switching Frequency vs Load Current TA = 25°C, unless otherwise noted. FRONT PAGE APPLICATION V IN = 12V V IN = 24V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 200 300 400 500 600 700 800 900 4.7 4.8 4.9 5.0 5.1 5.2 5.3 OUTPUT VOL TAGE (V)

8303 G01

V IN = 48V I OUT = 3mA I OUT = 200mA I OUT = 700mA TEMPERATURE (°C) –50 –25 100 125 150 4.7 4.8 4.9 5.0 5.1 5.2 5.3 OUTPUT VOL TAGE (V)

8303 G02

V IN = 12V V IN = 24V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 200 300 400 500 600 700 800 900 100 150 200 250 300 350 400 FREQUENCY (kHz)

8303 G03

V IN = 48V , I OUT = 700mA 2µs/DIV V SW 50V/DIV V OUT 50mV/DIV

8303 G04

V IN = 48V , I OUT = 200mA 2µs/DIV V SW 50V/DIV V OUT 50mV/DIV

8303 G05

V IN = 48V , I OUT = 3mA 20µs/DIV V SW 50V/DIV V OUT 50mV/DIV

8303 G06

T J = –50°C T J = 25°C T J = 150°C V IN (V) 100 I Q (µA)

8303 G07

T J = 150°C T J = 25°C T J = –50°C V IN (V) 100 100 I Q (µA)

8303 G08

T J = 150°C T J = 25°C T J = –50°C V IN (V) 100 240 260 280 300 320 340 I Q (µA)

8303 G09

Rev. BFor more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS RDS(ON) Switch Current Limit Maximum Switching Frequency Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time EN/UVLO Enable Threshold EN/UVLO Hysteresis Current RFB Regulation Current TA = 25°C, unless otherwise noted. RISING FALLING TEMPERATURE (°C) –50 –25 100 125 150 1.20 1.21 1.22 1.23 1.24 1.25 1.26 1.27 1.28 V EN/UVLO (V)

8303 G10

TEMPERATURE (°C) –50 –25 100 125 150 I HYST (µA)

8303 G11

TEMPERATURE (°C) –50 –25 100 125 150 100 101 102 103 104 105 I RFB (µA)

8303 G12

TEMPERATURE (°C) –50 –25 100 125 150 RESISTANCE (Ω)

8303 G13

1SW = 100mA MAXIMUM CURRENT LIMIT MINIMUM CURRENT LIMIT TEMPERATURE (°C) –50 –25 100 125 150 100 200 300 400 500 600 700 I SW (mA)

8303 G14

TEMPERATURE (°C) –50 –25 100 125 150 100 200 300 400 500 FREQUENCY (kHz)

8303 G15

TEMPERATURE (°C) –50 –25 100 125 150 100 200 300 400 TIME (ns)

8303 G17

TEMPERATURE (°C) –50 –25 100 125 150 100 200 300 400 TIME (ns)

8303 G18

TEMPERATURE (°C) –50 –25 100 125 150 FREQUENCY (kHz)

8303 G16

Rev. B For more information www.analog.com PIN FUNCTIONS EN/UVLO (Pin 1) : Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8303. Pull the pin below 0.3V to shut down the LT8303. This pin has an ac- curate 1.223V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resistor divider from V IN to ground. A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. GND (Pin 2): Ground. Tie this pin directly to local ground plane. RFB (Pin 3) : Input Pin for External Feedback Resistor . Connect a resistor from this pin to the transformer pri - mary SW pin. The ratio of the RFB resistor to the internal trimmed 12.23k resistor , times the internal bandgap reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. SW (Pin 4) : Drain of the 150V Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. VIN (Pin 5) : Input Supply. The V IN pin supplies current to internal circuitry and serves as a reference voltage for the feedback circuitry connected to the R FB pin. Locally bypass this pin to ground with a capacitor . BLOCK DIAGRAM 8303 BD OSCILLATOR 1:4 S R Q1.223V 25µA M2M3 BOUNDARY DETECTOR DRIVER RSENSE gmRREF 12.23kΩ RFB 2.5µA EN/UVLO 3 45 +1.223V REFERENCE REGULATORS VIN 2GND RFB SWVIN VIN NPS:1 DOUT LSECLPRI VOUT+ VOUT– COUT CIN

Rev. BFor more information www.analog.com OPERATION The LT8303 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation bound- ary. Opto-isolator circuits waste output power , and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unit- to-unit variation and aging over lifetime. Circuits employing extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT8303 samples the isolated output voltage through the primary-side flyback pulse waveform. In this manner , neither opto-isolator nor extra transformer winding is re- quired for regulation. Since the LT8303 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method im- proves load regulation without the need of external load compensation components. The LT8303 is a simple to use micropower isolated flyback converter housed in a 5-lead TSOT-23 package. The output voltage is programmed with a single external resistor . By integrating the loop compensation and soft-start inside, the part further reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switching regulators including reference, regulators, oscillator , logic, current amplifier , current comparator , driver , and power switch. The novel sections include a flyback pulse sense circuit, a sample-and-hold error amplifier , and a boundary mode detector , as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Boundary Conduction Mode Operation The LT8303 features boundary conduction mode operation at heavy load, where the chip turns on the primary power switch when the secondary current is zero. Boundary conduction mode is a variable frequency, variable peak- current switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around V IN. A boundary mode detector senses this event and turns the power switch back on. Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduc - tion mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit sub-harmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter , boundary conduction mode in - creases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switching and gate charge losses. To avoid this scenario, the LT8303 has an additional internal oscillator , which clamps the maximum switching frequency to be less than 350kHz (typical). Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8303 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch cur- rent limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8303 starts to fold back the switching frequency while keeping the minimum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sample-and- hold error amplifier . Meanwhile, the part switches between sleep mode and active mode, thereby reducing the effec-

Rev. B For more information www.analog.com OPERATION tive quiescent current to improve light load efficiency. In this condition, the LT8303 operates in low ripple Burst Mode. The typical 7kHz minimum switching frequency Output Voltage The RFB resistor as depicted in the Block Diagram is the only external resistor used to program the output voltage. The LT8303 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier , which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the V IN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: V FLBK = (VOUT + VF + ISEC • ESR) • NPS V F = Output diode forward voltage I SEC = T ransformer secondary current ESR = Total impedance of secondary circuit N PS = T ransformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current IRFB by the flyback pulse sense circuit ( M2 and M3). This cur - rent IRFB also flows through the internal trimmed 12.23k RREF resistor to generate a ground-referred voltage. The resulting voltage feeds to the inverting input of the sample- and-hold error amplifier . Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the VFLBK equation can be assumed to be zero. The bandgap reference voltage VBG, 1.223V, feeds to the non-inverting input of the sample-and-hold error ampli - fier . The relatively high gain in the overall loop causes the voltage across RREF resistor to be nearly equal to the APPLICATIONS INFORMATION determines how often the output voltage is sampled and also the minimum load requirement. bandgap reference voltage VBG. The resulting relationship between VFLBK and VBG can be expressed as: VFLBK RFB  •RREF = VBG or VFLBK = VBG RREF  •RFB = IRFB •RFB V BG = Bandgap reference voltage I RFB = RFB regulation current = 100µA Combination with the previous V FLBK equation yields an equation for VOUT, in terms of the RFB resistor , transformer turns ratio, and diode forward voltage: VOUT = 100µA • RFB NPS  − VF Output Temperature Coefficient The first term in the VOUT equation does not have tempera- ture dependence, but the output diode forward voltage VF has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coefficient pro- duces approximately 200mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible effect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however , the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. For customers requiring tight output voltage regulation across temperature, please refer to other L TC parts with integrated temperature compensation features.

Rev. BFor more information www.analog.com APPLICATIONS INFORMATION Selecting Actual RFB Resistor Value The LT8303 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the R FB resistor value. Therefore, a simple two-step process is required to choose feedback resistor RFB. Rearrangement of the expression for VOUT in the Output Voltage section yields the starting value for RFB: RFB = NPS • VOUT + VF( ) 100µA V OUT = Output voltage V F = Output diode forward voltage = ~0.3V N PS = T ransformer effective primary-to-secondary turns ratio Power up the application with the starting RFB value and other components connected, and measure the regulated output voltage, V OUT(MEAS). The final R FB value can be adjusted to: RFB(FINAL) = VOUT VOUT(MEAS)

• RFB Once the final RFB value is selected, the regulation accuracy from board to board for a given application will be very consistent, typically under ±5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ±1%). However , if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. Output Power A flyback converter has a complicated relationship between the input and output currents compared to a buck or a boost converter . A boost converter has a relatively constant maximum input current regardless of input voltage and a buck converter has a relatively constant maximum output current regardless of input voltage. This is due to the continuous non-switching behavior of the two currents. A flyback converter has both discontinuous input and output currents which make it similar to a non-isolated buck-boost converter . The duty cycle will affect the input and output currents, making it hard to predict output power . In ad- dition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. The graphs in Figures 1 to 4 show the typical maximum output power possible for the output voltages 3.3V, 5V, 12V, and 24V. The maximum output power curve is the calculated output power if the switch voltage is 120V dur- ing the switch-off time. 30V of margin is left for leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 120V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 30V and a maximum input voltage of 80V. A six-to-one winding ratio fits this design example perfectly and outputs equal to 4.35W at 80V but lowers to 2.95W at 30V. The following equations calculate output power: POUT = η • VIN •D •ISW(MAX) • 0.5 η = Efficiency = 85% D = DutyCycle = VOUT + VF( ) •NPS VOUT + VF( ) •NPS + VIN I SW(MAX) = Maximum switch current limit = 450mA

Figure 4. Output Power for 24V Output Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output Figure 3. Output Power for 12V Output

8303 F01

8303 F02

8303 F04

8303 F03

and may cause instability at light load. information should be carefully considered. shows the details of these transformers. Table 1. Predesigned T ransformers – Typical Specifications

Rev. B For more information www.analog.com Typically, choose the transformer turns ratio to maximize available output power . For low output voltages (3.3V or 5V), a larger N :1 turns ratio can be used with multiple primary windings relative to the secondary to maximize the transformer ’s current gain (and output power). However , remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 150V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure: NPS < 150V − VIN(MAX) − VLEAKAGE VOUT + VF For lower output power levels, choose a smaller N:1 turns ratio to alleviate the SW pin voltage stress. Although a 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch, the multiplied parasitic capacitance through turns ratio coupled with the relatively resistive 150V internal power switch may cause the switch turn-on current spike ringing beyond 160ns leading-edge blanking, thereby producing light load instability in certain applica- tions. So any 1 :N turns ratio should be fully evaluated before its use with the LT8303. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer speci- fies turns ratio accuracy within ±1%. APPLICATIONS INFORMATION Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8303, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output voltage regulation will be maintained independent of winding re- sistance due to the boundary/discontinuous conduction mode operation of the LT8303. Leakage Inductance and Snubbers T ransformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored en- ergy must be dissipated. It is very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus V IN should be kept below 120V. This leaves at least 30V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound trans - formers or for excessive leakage inductance. In addition to the voltage spikes, the leakage inductance also causes the SW pin ringing for a while after the power switch turns off. To prevent the voltage ringing falsely trig- ger boundary mode detector , the LT8303 internally blanks the boundary mode detector for approximately 250ns. Any remaining voltage ringing after 250ns may turn the power switch back on again before the secondary current falls to zero. So the leakage inductance spike ringing should be limited to less than 250ns.

A snubber circuit is recommended for most applications. provides better load regulation and EMI performance. voltage rating higher than the maximum SW pin voltage. compromise is to choose the largest voltage breakdown. around 65V and below the 70V maximum. applications when the highest VZENER is chosen. Figure 5. Maximum Voltages for SW Pin Flyback Waveform Figure 6. Snubber Circuits

8303 F05

8303 F06

Table 2. Recommended Zener Diodes Table 3. Recommended Diodes Figure 7. Undervoltage Lockout (UVLO)

8303 F07

converted to heat and will not be delivered to the load. may need to be sized for thermal dissipation. falling threshold is set at 1.223V with 16mV hysteresis. LT8303 in shutdown with quiescent current less than 2.5µA.

output, use a 6V Zener with cathode connected to the output. input range from 30V to 80V. Step 1: Select the T ransformer Turns Ratio. bility at different transformer turns ratio. Table 4. Switch Voltage Stress and Output Current Capability

Rev. B For more information www.analog.com APPLICATIONS INFORMATION Step 2: Determine the Primary Inductance. Primary inductance for the transformer must be set above a minimum value to satisfy the minimum switch-off and switch-on time requirements: LPRI ≥ tOFF(MIN) •NPS • VOUT + VF( ) ISW(MIN) LPRI ≥ tON(MIN) • VIN(MAX) ISW(MIN) t OFF(MIN) = 350ns t ON(MIN) = 160ns I SW(MIN) = 105mA Example: LPRI ≥ 350ns • 2 •(12V + 0.3V) 105mA = 82µH LPRI ≥ 160ns • 80V 105mA = 122µH Most transformers specify primary inductance with a toler- ance of ±20%. With other component tolerance considered, choose a transformer with its primary inductance 40% to 60% larger than the minimum values calculated above. LPRI = 150µH is then chosen in this example. The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 620mA is necessary to work with the LT8303. The PS15-111 from Sumida is chosen as the flyback transformer . Step 3: Choose the Output Diode. T wo main criteria for choosing the output diode include forward current rating and reverse voltage rating. The maximum load requirement is a good first-order guess as the average current requirement for the output diode. A conservative metric is the maximum switch current limit multiplied by the turns ratio, I DIODE(MAX) = ISW(MAX) • NPS Example: I DIODE(MAX) = 1.07A Next calculate reverse voltage requirement using maxi - mum VIN: VREVERSE = VOUT + VIN(MAX) NPS Example: VREVERSE = 12V + 72V 2 = 48V The DFLS2100 (2A, 100V diode) from Diodes Inc. is chosen. Step 4: Choose the Output Capacitor . The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor . Use the equation below to calculate the output capacitance: COUT = LPRI •ISW 2 • VOUT • ∆VOUT Example: Design for output voltage ripple less than 1% of V OUT, i.e., 120mV. COUT = 150µH •(0.535A)2 2 • 12V • 0.12V = 14.9µF Remember ceramic capacitors lose capacitance with ap- plied voltage. The capacitance can drop to 40% of quoted capacitance at the maximum voltage rating. So a 22µF, 25V rating X5R or X7R ceramic capacitor is chosen. Step 5: Design Snubber Circuit. The snubber circuit protects the power switch from leakage inductance voltage spike. A DZ snubber is recommended for this application because of lower leakage inductance and larger voltage margin. The Zener and the diode need to be selected.

Rev. BFor more information www.analog.com The maximum Zener breakdown voltage is set according to the maximum VIN: V ZENER(MAX) ≤ 150V – VIN(MAX) Example: V ZENER(MAX) ≤ 150V – 80V = 70V A 62V Zener with a maximum of 65V will provide optimal protection and minimize power loss. So a 62V, 0.5W Zener from Central Semiconductor (CMHZ5265B) is chosen. Choose a diode that is fast and has sufficient reverse voltage breakdown: V REVERSE > VSW(MAX) V SW(MAX) = VIN(MAX) + VZENER(MAX) Example: V REVERSE > 144V A 200V , 1A diode from Central Semiconductor (CMMRIU-02) is chosen. Step 6: Select the RFB Resistor . Use the following equation to calculate the starting value for RFB: RFB = NPS •(VOUT + VF ) 100µA Example: 100µA = 246k Depending on the tolerance of standard resistor values, the precise resistor value may not exist. For 1% standard values, a 243k resistor in series with a 3.01k resistor should be close enough. As discussed in the Application Information section, the final RFB value should be adjusted on the measured output voltage. APPLICATIONS INFORMATION Step 7: Select the EN/UVLO Resistors. Determine the amount of hysteresis required and calculate R1 resistor value: V IN(HYS) = 2.5µA • R1 Example: Choose 2.5V of hysteresis, R1 = 1M Determine the UVLO thresholds and calculate R2 resistor value: VIN(UVLO+) = 1.239V •(R1+ R2) R2 + 2.5µA •R1 Example: Set VIN UVLO rising threshold to 34.5V, R2 = 49.9k V IN(UVLO+) = 28.6V V IN(UVLO–) = 25.7V Step 8: Ensure minimum load. The theoretical minimum load can be approximately estimated as: ILOAD(MIN) = 150µH •(140mA)2 • 9kHz 2 • 12V = 1.1mA Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the con - verter delivers more energy than what is consumed at the output. The real minimum load for this application is about 1mA. In this example, a 12.1k resistor is selected as the minimum load.

Rev. B For more information www.analog.com TYPICAL APPLICATIONS 30V to 80VIN, 3.3VOUT Isolated Flyback Converter Efficiency vs Load Current Output Load and Line Regulation V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (A) 0.2 0.4 0.6 0.8 1.0 1.2 100 EFFICIENCY (%)

8303 TA02b

V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (A) 0.2 0.4 0.6 0.8 1.0 1.2 3.10 3.15 3.20 3.25 3.30 3.35 3.40 3.45 3.50 OUTPUT VOL TAGE (V)

8303 TA02c

8:1 RFB SW 150µH 2.3µH EN/UVLO 4.7µF 100V 49.9k VIN VIN 30V TO 80V VOUT+ 3.3V 4mA TO 0.9A (VIN = 36V) 4mA TO 1A (VIN = 48V) 4mA TO 1.1A (VIN = 72V) VOUT– GND 287k D1: CENTRAL CMMR1U-02 D2: DIODES SBR3U30P1-7 T1: SUMIDA PS15-108 Z1: CENTRAL CMHZ5265B 330µF 6.3V

8303 TA02a

Rev. BFor more information www.analog.com TYPICAL APPLICATIONS 30V to 80VIN, 5VOUT Isolated Flyback Converter Efficiency vs Load Current Output Load and Line Regulation V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 200 300 400 500 600 700 800 900 100 EFFICIENCY (%)

8303 TA03b

V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 200 300 400 500 600 700 800 900 4.7 4.8 4.9 5.0 5.1 5.2 5.3 OUTPUT VOL TAGE (V)

8303 TA03c

6:1 RFB SW 150µH 4.2µH EN/UVLO 4.7µF 100V 49.9k VIN VIN 30V TO 80V VOUT+ 2.5mA TO 0.65A (VIN = 36V) 2.5mA TO 0.73A (VIN = 48V) 2.5mA TO 0.84A (VIN = 72V) VOUT– GND 316k D1: CENTRAL CMMR1U-02 D2: DIODES SBR3U30P1-7 T1: SUMIDA PS15-109 Z1: CENTRAL CMHZ5265B 100µF 10V

8303 TA03a

Rev. B For more information www.analog.com Efficiency vs Load Current Output Load and Line Regulation TYPICAL APPLICATIONS 30V to 80VIN, 12VOUT Isolated Flyback Converter V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 120 160 200 240 280 320 11.4 11.6 11.8 12.0 12.2 12.4 12.6 OUTPUT VOL TAGE (V)

8303 TA04c

V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 120 160 200 240 280 320 100 EFFICIENCY (%)

8303 TA04b

2:1 RFB SW 150µH 37.5µH EN/UVLO 4.7µF 100V 49.9k VIN VIN 30V TO 80V VOUT+ 12V 1mA TO 250mA (VIN = 36V) 1mA TO 270mA (VIN = 48V) 1mA TO 310mA (VIN = 72V) VOUT– GND 249k D1: CENTRAL CMMR1U-02 D2: DIODES DFLS2100-7 T1: SUMIDA PS15-111 Z1: CENTRAL CMHZ5265B 22µF 25V

8303 TA04a

Rev. BFor more information www.analog.com TYPICAL APPLICATIONS LT8303 1:1 RFB SW 150µH 150µH EN/UVLO 4.7µF 100V 49.9k VIN VIN 30V TO 80V VOUT+ 24V 0.6mA TO 120mA (VIN = 36V) 0.6mA TO 140mA (VIN = 48V) 0.6mA TO 150mA (VIN = 72V) VOUT– GND 249k D1: CENTRAL CMMR1U-02 D2: DIODES DFLS1200-7 T1: SUMIDA PS15-112 Z1: CENTRAL CMHZ5265B 22µF 50V

8303 TA05a

V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 120 140 160 100 EFFICIENCY (%)

8303 TA05b

V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 120 140 160 22.8 23.2 23.6 24.0 24.4 24.8 25.2 OUTPUT VOL TAGE (V)

8303 TA05c

Efficiency vs Load Current Output Load and Line Regulation 30V to 80VIN, 24VOUT Isolated Flyback Converter

Rev. B For more information www.analog.com PACKAGE DESCRIPTION Please refer to http://www.linear .com/product/LT8303#packaging for the most recent package drawings. 1.50 – 1.75 (NOTE 4)2.80 BSC 0.30 – 0.45 TYP

5 PLCS (NOTE 3)

DATUM ‘A’ 0.09 – 0.20 (NOTE 3) S5 TSOT-23 0302 PIN ONE

2.90 BSC

(NOTE 4)

0.95 BSC

1.90 BSC

0.80 – 0.90

1.00 MAX

0.01 – 0.100.20 BSC 0.30 – 0.50 REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193

3.85 MAX

0.62 MAX 0.95 REF RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 1.4 MIN2.62 REF

1.22 REF

(Reference LTC DWG # 05-08-1635)

Rev. BFor more information www.analog.com

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER A 1/17 Added H-grade version 2, 3 B 11/22 AECQ100 Statement #W Models Added R grade 2, 3 Information furnished by Analog Devices is believed to be accurate and reliable. However , no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

Rev. B For more information www.analog.com RELATED PARTS TYPICAL APPLICATION 30V to 80VIN, 48VOUT Isolated Flyback Converter PART NUMBER DESCRIPTION COMMENTS L T8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 L T8304 100VIN Micropower Isolated Flyback Converter with 150V/2A Switch Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E L T8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 L T8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6mA Switch Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E L T8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 L T3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12) L T3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components L T3757/L T3759/ L T3758 40V/100V Flyback/Boost Controller Universal Controllers with Small Package and Powerful Gate Drive L T3957/L T3958 40V/80V Boost/Flyback Converter Monolithic with Integrated 5A/3.3A Switch LTC3803/LTC3803-3/ LTC3803-5 200kHz/300kHz Flyback Controller in SOT-23 VIN and VOUT Limited Only by External Components LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited Only by External Components LT8303 1:2 RFB SW 150µH 600µH EN/UVLO 4.7µF 100V 49.9k VIN VIN 30V TO 80V VOUT+ 48V 0.3mA TO 60mA (VIN = 36V) 0.3mA TO 70mA (VIN = 48V) 0.3mA TO 75mA (VIN = 72V) VOUT– GND 243k D1: CENTRAL CMMR1U-02 D2: DIODES SBR1U400P1-7 T1: SUMIDA PS15-113 Z1: CENTRAL CMHZ5265B 4.7µF 100V

8303 TA06a

V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 100 EFFICIENCY (%)

8303 TA06b

V IN = 36V V IN = 48V V IN = 72V LOAD CURRENT (mA) 45.6 46.4 47.2 48.0 48.8 49.6 50.4 OUTPUT VOL TAGE (V)

8303 TA06c

Efficiency vs Load Current Output Load and Line Regulation  ANALOG DEVICES, INC. 2017–2022 www.analog.com