LT8301HS5#WTRMPBF

LT8301 AD | Alldatasheet

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Rev. BFor more information www.analog.com Document Feedback TYPICAL APPLICATION FEATURES DESCRIPTION 42VIN Micropower No-Opto Isolated Flyback Converter with 65V/1.2A Switch The LT®8301 is a micropower isolated flyback converter . By sampling the isolated output voltage directly from the primary-side flyback waveform, the part requires no third winding or opto-isolator for regulation. The output voltage is programmed with a single external resistor . Internal compensation and soft-start further reduce external com- ponent count. Boundary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 1.2A, 65V DMOS power switch is integrated along with all high voltage circuitry and control logic into a 5-lead ThinSOT™ package. The LT8301 operates from an input voltage range of 2.7V to 42V and can deliver up to 6W of isolated output power . The high level of integration and the use of boundary and low ripple burst modes result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. All registered trademarks and trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, and 7471522. 2.7V to 36VIN/5VOUT Micropower Isolated Flyback Converter

APPLICATIONS

n 2.7V to 42V Input Voltage Range n 1.2A, 65V Internal DMOS Power Switch n Low Quiescent Current: n 100µA in Sleep Mode n 350µA in Active Mode n Boundary Mode Operation at Heavy Load n Low-Ripple Burst Mode® Operation at Light Load n Minimum Load <0.5% (Typ) of Full Output n VOUT Set with a Single External Resistor n No T ransformer Third Winding or Opto-Isolator Required for Regulation n Accurate EN/UVLO Threshold and Hysteresis n Internal Compensation and Soft-Start n Output Short-Circuit Protection n 5-Lead TSOT-23 Package n AEC-Q100 Qualified for Automotive Applications n Isolated Telecom, Automotive, Industrial, Medical Power Supplies n Isolated Auxiliary/Housekeeping Power Supplies Efficiency vs Load Current LT8301 3:1 RFB SW 40µH 4.4µH EN/UVLO 10µF VIN VIN 2.7V TO 36V VOUT+ 6mA TO 0.40A (VIN = 5V) 6mA TO 0.70A (VIN = 12V) 6mA TO 1.00A (VIN = 24V) 6mA TO 1.15A (VIN = 36V) VOUT– GND 154k 100µF

8301 TA01a

LOAD CURRENT (A) EFFICIENCY (%) 0.2 0.4 0.6 0.8

8301 TA01b

1.0 1.2 VIN = 5V VIN = 12V VIN = 24V VIN = 36V

Rev. B For more information www.analog.com PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS Operating Junction Temperature Range (Notes 3, 4) (Note 1) EN/UVLO 1 GND 2 TOP VIEW S5 PACKAGE 5-LEAD PLASTIC TSOT-23 θJA = 150°C/W RFB 3

5 VIN

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8301ES5#TRMPBF LT8301ES5#TRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8301IS5#TRMPBF LT8301IS5#TRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8301JS5#TRMPBF LT8301JS5#TRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 150°C LT8301HS5#TRMPBF LT8301HS5#TRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 150°C LT8301MPS5#TRMPBF LT8301MPS5#TRPBF L TGMF 5-Lead Plastic TSOT-23 –55°C to 150°C AUTOMOTIVE PRODUCTS** LT8301ES5#WTRMPBF LT8301ES5#WTRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8301IS5#WTRMPBF LT8301IS5#WTRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8301JS5#WTRMPBF LT8301JS5#WTRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 150°C LT8301HS5#WTRMPBF LT8301HS5#WTRPBF L TGMF 5-Lead Plastic TSOT-23 –40°C to 150°C Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container . Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. **Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for these models.

Rev. BFor more information www.analog.com

ELECTRICAL CHARACTERISTICS

Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 65V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 65V as shown in Figure 5. Note 3: The LT8301E is guaranteed to meet performance specifications from 0°C to 125°C operating junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VEN/UVLO = VIN unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT VIN Input Voltage Range l 2.7 42 V VIN UVLO Threshold Rising Falling 2.5 2.3 2.65 V V IQ VIN Quiescent Current VEN/UVLO = 0.2V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 0.8 215 100 350 2 µA µA µA µA EN/UVLO Shutdown Threshold For Lowest Off IQ l 0.2 0.55 V EN/UVLO Enable Threshold Falling Hysteresis 1.204 1.228 0.014 1.248 V V IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.2V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.2 –0.1 2.5 0.1 2.8 0.1 µA µA µA fMIN Minimum Switching Frequency 9.4 10 10.6 kHz tON(MIN) Minimum Switch-On Time 170 ns tOFF(MAX) Maximum Switch-Off Time Backup Timer 190 µs ISW(MAX) Maximum SW Current Limit l 1.200 1.375 1.550 A ISW(MIN) Minimum SW Current Limit l 0.22 0.29 0.36 A RDS(ON) Switch On-Resistance ISW = 500mA 0.4 Ω ILKG Switch Leakage Current VIN = 42V, VSW = 65V 0.1 0.5 µA IRFB RFB Regulation Current l 97.5 100 102.5 µA RFB Regulation Current Line Regulation 2.7V ≤ VIN ≤ 42V 0.02 0.1 %/V The LT8301I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8301J and LT8301H are guaranteed over the full –40°C to 150°C operating junction temperature range. The LT8301MP is guaranteed over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125°C. Note 4: The LT8301 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.

Rev. B For more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms VIN Shutdown Current VIN Quiescent Current, Sleep Mode VIN Quiescent Current, Active Mode Output Load and Line Regulation Output Short-Circuit Protection Switching Frequency vs Load Current TA = 25°C, unless otherwise noted. LOAD CURRENT (A) 350 300 250 200 150 100 0.6 1.0

8301 G03

0.2 0.4 0.8 1.2 SWITCHING FREQUENCY (kHz) VIN = 5V VIN = 12V VIN = 24V VIN = 36V FRONT PAGE APPLICATION VIN (V) IQ (µA) 10 20 25 45

8301 G07

TJ = 150°C TJ = 25°C TJ = –55°C VIN (V) IQ (µA)100 110 120

8301 G08

TJ = 150°C TJ = 25°C TJ = –55°C VIN (V) IQ (µA) 320 340 360

8301 G09

TJ = 150°C TJ = 25°C TJ = –55°C LOAD CURRENT (A) OUTPUT VOLTAGE (V) 5.00

8301 G01

4.90 4.80 5.10

5.20 FRONT PAGE APPLICATION

4.95 4.85 5.05 5.15 1.2 VIN = 5V VIN = 12V VIN = 24V VIN = 36V LOAD CURRENT (A) OUTPUT VOLTAGE (V) 0.4 0.8 1.2 1.6

8301 G02

0.2 0.6 1.0 1.4 VIN = 5V VIN = 12V VIN = 24V VIN = 36V FRONT PAGE APPLICATION VOUT 50mV/DIV VSW 20V/DIV 5µs/DIV FRONT PAGE APPLICATION VIN = 12V ILOAD = 600mA

8301 G04

5µs/DIV FRONT PAGE APPLICATION VIN = 12V ILOAD = 200mA

8301 G05

20µs/DIV FRONT PAGE APPLICATION VIN = 12V ILOAD = 6mA

8301 G06

Rev. BFor more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS RDS(ON) Switch Current Limit Maximum Switching Frequency Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time EN/UVLO Enable Threshold EN/UVLO Hysteresis Current RFB Regulation Current TA = 25°C, unless otherwise noted. TEMPERATURE (°C) VEN/UVLO (V) 1.245 1.210 1.215 1.230 1.235 1.240 1.220 1.225 1.205

8301 G10

1507550 125100250–25–50 TEMPERATURE (°C) IHYS (µA)

8301 G11

1507550 125100250–25–50 TEMPERATURE (°C) IRFB (µA) 105 101 102 103 104 100

8301 G12

1507550 125100250–25–50 TEMPERATURE (°C) RESISTANCE (mΩ) 1000 200 400 600 800

8301 G13

1507550 125100250–25–50 TEMPERATURE (°C) –50 ISW (A)0.8 1.2 150

8301 G14

0.4 0 50 100–25 25 75 125 1.6 0.6 1.0 0.2

1.4 MAXIMUM CURRENT LIMIT

TEMPERATURE (°C) –50 FREQUENCY (kHz) 100 200 300 400 0 50 100 150

8301 G15

–25 25 75 125 TEMPERATURE (°C) –50 FREQUENCY (kHz) –25 0 25 50

8301 G16

TEMPERATURE (°C) TIME (ns) 500 100 200 300 400

8301 G17

1507550 125100250–25–50 TEMPERATURE (°C) TIME (ns) 500 100 200 300 400

8301 G18

1507550 125100250–25–50

Rev. B For more information www.analog.com PIN FUNCTIONS EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8301. Pull the pin below 0.2V to shut down the LT8301. This pin has an accurate 1.228V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resis- tor divider from VIN to ground. A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. GND (Pin 2): Ground. Tie this pin directly to local ground plane. RFB (Pin 3): Input Pin for External Feedback Resistor . Connect a resistor from this pin to the transformer primary SW pin. The ratio of the R FB resistor to an internal 10k resistor , times a trimmed 1.0V reference voltage, deter- mines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. SW (Pin 4): Drain of the 65V Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. VIN (Pin 5) : Input Supply. The V IN pin supplies current to internal circuitry and serves as a reference voltage for the feedback circuitry connected to the R FB pin. Locally bypass this pin to ground with a capacitor . BLOCK DIAGRAM 8301 BD OSCILLATOR 1:4 S R Q1.0V 25µA M2M3 BOUNDARY DETECTOR DRIVER RSENSE gmRREF 10kΩ RFB 2.5µA EN/UVLO 3 45 +1.228V REFERENCE REGULATORS VIN 2GND RFB SWVIN VIN NPS:1 DOUT LSECLPRI VOUT+ VOUT– COUT CIN

Rev. BFor more information www.analog.com OPERATION The LT8301 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to commu - nicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra trans- former windings communicate this information across the isolation boundary. Opto-isolator circuits waste output power , and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unit-to-unit variation and aging over life - time. Circuits employing extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer’ s physical size and cost, and dynamic response is often mediocre. The LT8301 samples the isolated output voltage through the primary-side flyback pulse waveform. In this man - ner , neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8301 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method improves load regulation without the need of external load compensation components. The LT8301 is a simple to use micropower isolated fly - back converter housed in a 5-lead TSOT-23 package. The output voltage is programmed with a single external resis- tor . By integrating the loop compensation and soft-start inside, the part further reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switch- ing regulators including reference, regulators, oscillator , logic, current amplifier , current comparator , driver , and power switch. The novel sections include a flyback pulse sense circuit, a sample-and-hold error amplifier , and a boundary mode detector , as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Boundary Conduction Mode Operation The LT8301 features boundary conduction mode opera- tion at heavy load, where the chip turns on the primary power switch when the secondary current is zero. Boundary conduction mode is a variable frequency, vari- able peak-current switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around V IN. A boundary mode detector senses this event and turns the power switch back on. Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduc - tion mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit sub-harmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter , boundary conduction mode increases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switch- ing and gate charge losses. To avoid this scenario, the LT8301 has an additional internal oscillator , which clamps the maximum switching frequency to be less than 430kHz (typ). Once the switching frequency hits the internal fre- quency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8301 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling

Rev. B For more information www.analog.com Output Voltage The RFB resistor as depicted in the Block Diagram is the only external resistor used to program the output voltage. The LT8301 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier , which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the V IN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: V FLBK = (VOUT + VF + ISEC • ESR) • NPS V F = Output diode forward voltage I SEC = T ransformer secondary current ESR = Total impedance of secondary circuit N PS = T ransformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current IRFB by the flyback pulse sense circuit (M2 and M3). This current IRFB also flows through the internal 10k R REF resistor to generate a ground-referred voltage. The resulting volt - age feeds to the inverting input of the sample-and-hold error amplifier . Since the sample-and-hold error ampli - fier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the V FLBK equation can be assumed to be zero. APPLICATIONS INFORMATION An internal trimmed reference voltage,V IREF 1.0V, feeds to the non-inverting input of the sample-and-hold error amplifier . The relatively high gain in the overall loop causes the voltage across RREF resistor to be nearly equal to VIREF. The resulting relationship between V FLBK and VIREF can be expressed as: VFLBK RFB ⎠⎟•RREF = VIREF or VFLBK = VIREF RREF ⎠⎟•RFB =IRFB •RFB V IREF = Internal trimmed reference voltage I RFB = RFB regulation current = 100µA Combination with the previous V FLBK equation yields an equation for V OUT, in terms of the R FB resistor , trans- former turns ratio, and diode forward voltage: VOUT = 100µA • RFB NPS ⎠⎟−VF Output Temperature Coefficient The first term in the V OUT equation does not have tem - perature dependence, but the output diode forward volt- age VF has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coef- ficient produces approximately 200mV to 300mV voltage variation on the output voltage across temperature. OPERATION of the output voltage. The inherent minimum switch cur- rent limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8301 starts to fold back the switching frequency while keeping the mini - mum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sample-and-hold error amplifier . Meanwhile, the part switches between sleep mode and active mode, thereby reducing the effective quiescent current to improve light load efficiency. In this condition, the LT8301 operates in low ripple Burst Mode. The 10kHz (typ) minimum switch- ing frequency determines how often the output voltage is sampled and also the minimum load requirement.

Rev. BFor more information www.analog.com APPLICATIONS INFORMATION For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible effect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however , the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. For customers requiring tight output voltage regulation across temperature, please refer to other ADI parts with integrated temperature compensa- tion features. Selecting Actual RFB Resistor Value The LT8301 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evalua- tion of the RFB resistor value. Therefore, a simple two-step process is required to choose feedback resistor RFB. Rearrangement of the expression for VOUT in the Output Voltage section yields the starting value for RFB: RFB = NPS • VOUT + VF( ) 100µA V OUT = Output voltage V F = Output diode forward voltage = ~0.3V N PS = T ransformer effective primary-to-secondary turns ratio Power up the application with the starting RFB value and other components connected, and measure the regulated output voltage, V OUT(MEAS). The final R FB value can be adjusted to: RFB(FINAL) = VOUT VOUT(MEAS)

• RFB Once the final RFB value is selected, the regulation accu- racy from board to board for a given application will be very consistent, typically under ± 5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ± 1%). However , if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. Output Power A flyback converter has a complicated relationship between the input and output currents compared to a buck or a boost converter . A boost converter has a rela- tively constant maximum input current regardless of input voltage and a buck converter has a relatively constant maximum output current regardless of input voltage. This is due to the continuous non-switching behavior of the two currents. A flyback converter has both discontinu - ous input and output currents which make it similar to a non-isolated buck-boost converter . The duty cycle will affect the input and output currents, making it hard to predict output power . In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. The graphs in Figures 1 to 4 show the typical maximum output power possible for the output voltages 3.3V, 5V, 12V, and 24V. The maximum output power curve is the calculated output power if the switch voltage is 50V dur- ing the switch-off time. 15V of margin is left for leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 50V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 8V and a maximum input volt- age of 32V. A three-to-one winding ratio fits this design example perfectly and outputs equal to 5.42W at 32V but lowers to 2.71W at 8V. The following equations calculate output power: POUT = η•VIN •D•ISW(MAX) •0.5 η=Efficiency = ∼85% D=DutyCycle = VOUT + VF( ) •NPS VOUT + VF( ) •NPS + VIN I SW(MAX) = Maximum switch current limit = 1.2A (min)

Figure 1. Output Power for 3.3V Output

8301 F01

8301 F02

8301 F03

8301 F04

the power switch shorter than approximately 170ns. Figure 2. Output Power for 5V Output Figure 3. Output Power for 12V Output Figure 4. Output Power for 24V Output

following information should be carefully considered. the details of these transformers. Table 1. Predesigned T ransformers—Typical Specifications

Rev. B For more information www.analog.com Typically, choose the transformer turns ratio to maximize available output power . For low output voltages (3.3V or 5V), a larger N:1 turns ratio can be used with multiple pri- mary windings relative to the secondary to maximize the transformer’s current gain (and output power). However , remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 65V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure: NPS < 65V −VIN(MAX) −VLEAKAGE VOUT + VF For lower output power levels, choose a smaller N:1 turns ratio to alleviate the SW pin voltage stress. Although a 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch, the multiplied parasitic capacitance through turns ratio may cause the switch turn-on cur - rent spike ringing beyond 170ns leading-edge blanking, thereby producing light load instability in certain applica- tions. So any 1: N turns ratio should be fully evaluated before its use with the LT8301. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer speci- fies turns ratio accuracy within ±1%. APPLICATIONS INFORMATION Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8301 , the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output volt - age regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction mode operation of the LT8301. Leakage Inductance and Snubbers T ransformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. It is very impor- tant to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus VIN should be kept below 50V. This leaves at least 15V margin for the leakage spike across line and load condi - tions. A larger voltage margin will be required for poorly wound transformers or for excessive leakage inductance. In addition to the voltage spikes, the leakage inductance also causes the SW pin ringing for a while after the power switch turns off. To prevent the voltage ringing falsely trig- gering the boundary mode detector , the LT8301 internally blanks the boundary mode detector for approximately 350ns. Any remaining voltage ringing after 350ns may turn the power switch back on again before the second - ary current falls to zero. So the leakage inductance spike ringing should be limited to less than 350ns.

A snubber circuit is recommended for most applications. Zener) snubber and the RC (resistor-capacitor) snubber . compromise is to choose the largest voltage breakdown. around 21V and below the 33V maximum. highest at maximum load and minimum input voltage. Figure 5. Maximum Voltages for SW Pin Flyback Waveform Figure 6. Snubber Circuits

8301 F05

8301 F06b8300 F06a

Table 2. Recommended Zener Diodes Table 3. Recommended Diodes DFLS1100 1 100 PowerDI-123 Diodes Inc. DFLS1150 1 150 PowerDI-123 Diodes Inc. Figure 7. Undervoltage Lockout (UVLO)

8301 F07

converted to heat and will not be delivered to the load. may need to be sized for thermal dissipation. falling threshold is set at 1.228V with 14mV hysteresis. LT8301 in shutdown with quiescent current less than 2µA.

the average output diode current. Step 1: Select the T ransformer Turns Ratio. bility at different transformer turns ratio. Table 4. Switch Voltage Stress and Output Current Capability requirement, NPS = 3 is chosen in this example.

Rev. B For more information www.analog.com APPLICATIONS INFORMATION Step 2: Determine the Primary Inductance. Primary inductance for the transformer must be set above a minimum value to satisfy the minimum switch-off and switch-on time requirements: LPRI ≥ tOFF(MIN) •NPS • VOUT + VF( ) ISW(MIN) LPRI ≥ tON(MIN) •VIN(MAX) ISW(MIN) t OFF(MIN) = 450ns t ON(MIN) = 170ns I SW(MIN) = 290mA (typ) Example: LPRI ≥450ns•3•(5V +0.3V) 290mA = 25µH LPRI ≥170ns•32V 290mA = 19µH Most transformers specify primary inductance with a tol- erance of ±20%. With other component tolerance consid- ered, choose a transformer with its primary inductance 30% larger than the minimum values calculated above. LPRI = 40µH is then chosen in this example. Once the primary inductance has been determined, the maximum load switching frequency can be calculated as: fSW = 1 tON + tOFF = 1 LPRI •ISW VIN + LPRI •ISW NPS •(VOUT + VF ) ISW = VOUT •IOUT •2 η•VIN •D Example: D= (5V +0.3V)•3 ISW = 5V •0.5A •2 fSW = 199kHz The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 2A is necessary to work with the LT8301. The 750313974 from Würth is chosen as the flyback transformer . Step 3: Choose the Output Diode. T wo main criteria for choosing the output diode include forward current rating and reverse voltage rating. The maximum load requirement is a good first-order guess as the average current requirement for the output diode. A conservative metric is the maximum switch current limit multiplied by the turns ratio, I DIODE(MAX) = ISW(MAX) • NPS Example: I DIODE(MAX) = 4.125A Next calculate reverse voltage requirement using maxi - mum VIN: VREVERSE = VOUT + VIN(MAX) NPS Example: VREVERSE = 5V + 32V 3 = 15.6V The CMS H5-20 ( 5A, 20V diode) from Central Semiconductor is chosen.

Rev. BFor more information www.analog.com Step 4: Choose the Output Capacitor . The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor . Use the equation below to calculate the output capacitance: COUT = LPRI •ISW 2•VOUT •ΔVOUT Example: Design for output voltage ripple less than 1% of V OUT, i.e., 50mV. COUT = 40µH•(0.86A)2 2•5V •0.05V = 60µF Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop to 40% of quoted capacitance at the maximum voltage rating. So a 100µF, 10V rating ceramic capacitor is chosen X5R or X7R type is recommended. Step 5: Design Snubber Circuit. The snubber circuit protects the power switch from leak- age inductance voltage spike. A DZ snubber is recom - mended for this application because of lower leakage inductance and larger voltage margin. The Zener and the diode need to be selected. The maximum Zener breakdown voltage is set according to the maximum VIN: V ZENER(MAX) ≤ 65V – VIN(MAX) Example: V ZENER(MAX) ≤ 65V – 32V = 33V APPLICATIONS INFORMATION A 20V Zener with a maximum of 21V will provide optimal protection and minimize power loss. So a 20V, 0.25W Zener from Central Semiconductor (CMDZ5250B) is chosen. Choose a diode that is fast and has sufficient reverse volt- age breakdown: V REVERSE > VSW(MAX) V SW(MAX) = VIN(MAX) + VZENER(MAX) Example: V REVERSE > 53V A 100V, 0.25A diode from Central Semiconductor (CMHD4448) is chosen. Step 6: Select the RFB Resistor . Use the following equation to calculate the starting value for RFB: RFB = NPS •(VOUT + VF ) 100µA Example: RFB = 3•(5V +0.3V) 100µA = 159k Depending on the tolerance of standard resistor values, the precise resistor value may not exist. For 1% standard values, a 158k resistor should be close enough. As dis - cussed in the Application Information section, the final RFB value should be adjusted on the measured output voltage.

Rev. B For more information www.analog.com APPLICATIONS INFORMATION Step 7: Select the EN/UVLO Resistors. Determine the amount of hysteresis required and calcu - late R1 resistor value: V IN(HYS) = 2.5µA • R1 Example: Choose 2V of hysteresis, R1 = 806k Determine the UVLO thresholds and calculate R2 resistor value: VIN(UVLO+) = 1.242V •(R1+R2) R2 +2.5µA •R1 Example: Set VIN UVLO rising threshold to 7.5V, R2 = 232k V IN(UVLO+) = 7.5V V IN(UVLO–) = 5.5V Step 8: Ensure minimum load. The theoretical minimum load can be approximately esti- mated as: ILOAD(MIN) = 40µH•(360mA)2 •10.6kHz 2•5V = 5.5mA Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the con - verter delivers more energy than what is consumed at the output. The real minimum load for this application is about 6mA. In this example, a 820Ω resistor is selected as the minimum load.

Rev. BFor more information www.analog.com TYPICAL APPLICATIONS 2.7V to 36VIN/15VOUT Micropower Isolated Flyback Converter 8V to 36VIN/3.3VOUT Micropower Isolated Flyback Converter Efficiency vs Load Curent LT8301 1:1 RFB SW 40µH 40µH D1: CENTRAL CMHD4448 D2: CENTRAL CMMR1U-02 T1: SUMIDA 12387-T041 Z1: CENTRAL CMDZ5248B EN/UVLO 10µF VIN VIN 2.7V TO 36V VOUT+ 15V 2mA TO 130mA (VIN = 5V) 2mA TO 230mA (VIN = 12V) 2mA TO 320mA (VIN = 24V) 2mA TO 370mA (VIN = 36V) VOUT– GND 150k 10µF

8301 TA02a

LOAD CURRENT (mA) EFFICIENCY (%) 100 200 300 400

8301 TA02b

VIN = 5V VIN = 12V VIN = 24V VIN = 36V LT8301 4:1 RFB SW 40µH 2.5µH D1: CENTRAL CMHD4448 D2: NXP PMEG2020EH T1: SUMIDA 12387-T036 Z1: CENTRAL CMDZ5250B EN/UVLO 806k 4.7µF VIN VIN 8V TO 36V VOUT+ 3.3V 8.5mA TO 0.95A (VIN = 12V) 8.5mA TO 1.30A (VIN = 24V) 8.5mA TO 1.50A (VIN = 36V) VOUT– GND 137k232k 47µF

8301 TA03

Rev. B For more information www.analog.com TYPICAL APPLICATIONS 8V to 36VIN/24VOUT Micropower Isolated Flyback Converter 8V to 36VIN/48VOUT Micropower Isolated Flyback Converter Efficiency vs Load Curent LT8301 1:2 RFB SW 40µH 160µH D1: CENTRAL CMHD4448 D2: ST STPS1150A T1: WÜRTH 750313975 Z1: CENTRAL CMDZ5248B EN/UVLO 806k 4.7µF VIN VIN 8V TO 36V VOUT+ 24V 1.2mA TO 130mA (VIN = 12V) 1.2mA TO 180mA (VIN = 24V) 1.2mA TO 200mA (VIN = 36V) VOUT– GND 121k232k 4.7µF

8301 TA04a

LOAD CURRENT (mA) EFFICIENCY (%) 50 100 150 200

8301 TA04b

VIN = 12V VIN = 24V VIN = 36V LT8301 1:4 RFB SW 40µH 640µH D1: CENTRAL CMHD4448 D2: DIODES BAV21W-7-F T1: WÜRTH 750313976 Z1: CENTRAL CMDZ5252B EN/UVLO 806k 4.7µF VIN VIN 8V TO 36V VOUT+ 48V 0.6mA TO 70mA (VIN = 12V) 0.6mA TO 90mA (VIN = 24V) 0.6mA TO 100mA (VIN = 36V) VOUT– GND 118k232k 1µF

Rev. BFor more information www.analog.com TYPICAL APPLICATIONS VIN to (VIN + 10V)/(VIN – 10V) Micropower Converter 12V to 24VIN/Four 15VOUT Micropower Isolated Flyback Converter LT8301 D1 D2T1 1:1:1:1:1 RFB SW 30µH 30µH EN/UVLO 806k 4.7µF 232k VIN VIN 12V TO 24V GND 150k D1: CENTRAL CMHD4448 D2-D5: CENTRAL CMMR1U-02 T1: SUMIDA EPH2815-ADBN-A0349 Z1: CENTRAL CMDZ5248B 7.5k VOUT1+ 15V 60mA VOUT1–

8301 TA07

2.2µF 30µH 7.5k VOUT2+ 15V 60mA VOUT2– 2.2µF 30µH 7.5k VOUT3+ 15V 60mA VOUT3– 2.2µF 30µH 7.5k VOUT4+ 15V 60mA VOUT4– 2.2µF LT8301 RFB SWEN/UVLO 10µF VIN 1:1 VIN 2.7V TO 42V VIN + 10V 150mA VIN – 10V VIN 150mA GND 102k 40µH40µH D1, D2: DIODES INC. DFLS160 T1: SUMIDA 12387-T041 Z1: CENTRAL CMDZ12L 4.7µF 4.7µF

8301 TA06

Z2•

Rev. B For more information www.analog.com PACKAGE DESCRIPTION 1.50 – 1.75 (NOTE 4)2.80 BSC 0.30 – 0.45 TYP

5 PLCS (NOTE 3)

DATUM ‘A’ 0.09 – 0.20 (NOTE 3) S5 TSOT-23 0302 REV B PIN ONE

2.90 BSC

(NOTE 4)

0.95 BSC

1.90 BSC

0.80 – 0.90

1.00 MAX

0.01 – 0.100.20 BSC 0.30 – 0.50 REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193

3.85 MAX

0.62 MAX 0.95 REF RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 1.4 MIN2.62 REF

1.22 REF

(Reference LTC DWG # 05-08-1635 Rev B)

Rev. BFor more information www.analog.com Information furnished by Analog Devices is believed to be accurate and reliable. However , no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER A 04/19 Added AEC-Q100 Qualified Front Page Feature Bullet Added Automotive (W) Flow Parts to Order Information Section B 05/20 Added J-Grade Option and Specifications 2, 3

Rev. B For more information www.analog.com www.analog.com  ANALOG DEVICES, INC. 2014-2020 RELATED PARTS TYPICAL APPLICATION PART NUMBER DESCRIPTION COMMENTS L T8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT-23 L T8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6A Switch Low IQ Monolithic No-Opto Flybacks, 8-Lead SO-8E L T8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 L T3511/L T3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) L T3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing L T3798 Off-Line Isolated No Opto-Coupler Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components L T3573/L T3574/L T357540V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch L T3757A/L T3759/ L T3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive L T3957/L T3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch LT C 3803/LTC3803-3/ LTC3803-5 200kHz/300kHz Flyback Controllers in SOT-23 VIN and VOUT Limited by External Components LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited by External Components Efficiency vs Load Current Output Load and Line Regulation 8V to 36VIN/12VOUT Micropower Isolated Flyback Converter LT8301 1:1 RFB SW 40µH 40µH D1: CENTRAL CMHD4448 D2: DIODE INC. DFLS160 T1: WÜRTH 750313972 Z1: CENTRAL CMDZ5250B EN/UVLO 806k 4.7µF VIN VIN 8V TO 36V VOUT+ 12V 2.5mA TO 270mA (VIN = 12V) 2.5mA TO 360mA (VIN = 24V) 2.5mA TO 400mA (VIN = 36V) VOUT– GND 118k232k 10µF

8301 TA08a

LOAD CURRENT (mA) EFFICIENCY (%) 100 200 300 400

8301 TA08b

VIN = 12V VIN = 24V VIN = 36V LOAD CURRENT (mA) OUTPUT VOLTAGE (V) 12.0 12.1 12.2 400

8301 TA08c

11.9 11.8 11.6 100 200 300 11.7 12.4 12.3 VIN = 12V VIN = 24V VIN = 36V