LT3748

LT3748 (Rev. D)

Document overview

• Manufacturer or author: Analog Devices, Inc.
• PDF pages: 34

Technical content

Rev. DFor more information www.analog.com Document Feedback

FEATURES

APPLICATIONS

DESCRIPTION

The LT®3748 is a switching regulator controller specifi - cally designed for the isolated flyback topology and capa- ble of high power . It drives a low side external N-channel power MOSFET from an internally regulated 7V supply. No third winding or opto-isolator is required for regulation as the part senses the isolated output voltage directly from the primary-side flyback waveform. The LT3748 utilizes boundary mode to provide a small magnetic solution without compromising load regula - tion. Operating frequency is set by load current and transformer magnetizing inductance. The gate drive of the LT3748 combined with a suitable external MOSFET allow it to deliver load power up to several tens of watts from input voltages as high as 100V. The LT3748 is available in a high voltage 16-lead MSOP package with four leads removed. 25W, 12V Output, Isolated Telecom Supply n 5V to 100V Input Voltage Range n 1.9A Average Gate Drive Source and Sink Current n Boundary Mode Operation n No T ransformer Third Winding or Opto-Isolator Required for Regulation n Primary-Side Winding Feedback Load Regulation n VOUT Set with T wo External Resistors n INTVCC Pin for Control of Gate Driver Voltage n Programmable Soft Start n Programmable Undervoltage Lockout n Available in MSOP Package n AEC-Q100 Qualified for Automotive Applications n Isolated Telecom Converters n High Power Automotive Supplies n Isolated Industrial Power Supplies n Military and High Temperature Applications Output Load and Line Regulation TYPICAL APPLICATION EN/UVLO TC SS RFB RREF VC GND INTVCC L T3748 3748 TA01a56.2k 2nF VIN 36V TO 72V VOUT+ 12V VOUT– VIN 4:1 412k 15.4k 10µF 3.8µH60.8µH 100µF GATE SENSE 4700pF 4.7µF 6.04k 243k 0.033/uni03A9 10k LOAD CURRENT (A)

11.4 VOUT (V)

11.6 11.8 12.0 12.2 12.4 12.6 0.5 1.0 1.5 2.0

3748 TA01b

VIN = 72V VIN = 48V VIN = 36V All registered trademarks and trademarks are the property of their respective owners. Protected by U.S. patents, including 5438499 and 7471522.

Rev. D For more information www.analog.com ABSOLUTE MAXIMUM RATINGS Operating Junction Temperature Range (Note 2) ORDER INFORMATION ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, unless otherwise noted. LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3748EMS#PBF LT3748EMS#TRPBF 3748 16-Lead Plastic MSOP –40°C to 125°C LT3748IMS#PBF LT3748IMS#TRPBF 3748 16-Lead Plastic MSOP –40°C to 125°C LT3748HMS#PBF LT3748HMS#TRPBF 3748 16-Lead Plastic MSOP –40°C to 150°C LT3748MPMS#PBF LT3748MPMS#TRPBF 3748 16-Lead Plastic MSOP –55°C to 150°C AUTOMOTIVE PRODUCTS** LT3748IMS#WPBF LT3748IMS#WTRPBF 3748 16-Lead Plastic MSOP –40°C to 125°C LT3748HMS#WPBF LT3748HMS#WTRPBF 3748 16-Lead Plastic MSOP –40°C to 150°C Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container . Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. **Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for these models. PARAMETER CONDITIONS MIN TYP MAX UNITS Input Voltage Range l 5 100 V Quiescent Current Not Switching VEN/UVLO = 0.2V 1.3 1.75 mA µA VIN Quiescent Current, INTVCC Overdriven VINTVCC = 10V 300 450 µA INTVCC Voltage Range l 4.5 20 V INTVCC Pin Regulation Voltage 6.8 7 7.2 V INTVCC Dropout (VIN – VINTVCC), IINTVCC = 10mA, VIN = 5V 0.7 V INTVCC Undervoltage Lockout Falling Threshold l 3.45 3.6 3.75 V (Note 1) PIN CONFIGURATION VIN EN/UVLO INTVCC GATE SENSE GND RFB RREF TC VC SS GND TOP VIEW MS PACKAGE 16 (12)-LEAD PLASTIC MSOP TJMAX = 150°C, θJA = 90°C/W

Rev. DFor more information www.analog.com Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3748E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design characterization and correlation with statistical process controls. The LT3748I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT3748H is guaranteed over the full –40°C to 150°C operating junction temperature range. The LT3748MP is guaranteed over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 3: Current flows out of the pin. PARAMETER CONDITIONS MIN TYP MAX UNITS EN/UVLO Pin Threshold EN/UVLO Pin Voltage Rising l 1.19 1.223 1.25 V EN/UVLO Pin Hysteresis Current EN/UVLO = 1V 1.9 2.4 2.9 µA Soft-Start Current VSS = 0.4V (Note 3) 5 µA Soft-Start Threshold 0.65 V Soft-Start Reset Current 3 mA Maximum SENSE Current Limit Threshold VC = 2.2V l 100 100 105 110 mV mV Minimum SENSE Current Limit Threshold VC = 0V 15 mV Maximum to Minimum SENSE Threshold Ratio l 5.2 6.6 8.2 mV/mV SENSE Overcurrent Threshold VC = 2.2V 115 130 145 mV SENSE Input Bias Current VSENSE = 10mV (Note 3) 10 15 20 µA RREF Voltage VC = 1.1V l 1.20 1.195 1.223 1.24 1.245 V V RREF Voltage Line Regulation 5V < VIN < 100V 0.005 0.025 %/V RREF Pin Bias Current (Note 3) l 35 500 nA TC Current into RREF RTC = 20k 27.5 µA Error Amplifier Voltage Gain 115 V/V Error Amplifier T ransconductance ∆I = 10µA 155 µmhos VC Source Current VC = 1.1V, VRREF = 0.5V –45 µA VC Sink Current VC = 1.1V, VRREF = 2V 48 µA Flyback Comparator T rip Current Current into RFB Pin, RREF = 6.04k 10 µA Minimum GATE Off-Time 700 ns Minimum GATE On-Time 250 ns Maximum Discontinuous Off-Time VC = 0V 24 µs Maximum GATE Off-Time VRREF = 0.5V 55 µs Maximum GATE On-Time VSENSE = 0V 55 µs GATE Output Rise Time CL = 3300pF, 10% to 90% 16 ns GATE Output Fall Time CL = 3300pF, 10% to 90% 16 ns GATE Output Low (VOL) 0.05 V GATE Output High (VOH) VINTVCC – 0.05 V ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, unless otherwise noted.

Rev. D For more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS INTVCC Undervoltage Lockout vs TemperatureINTVCC Voltage vs VIN Voltage INTVCC Regulator Dropout vs INTVCC Current Soft-Start Current vs Temperature Output Regulation vs Temperature Quiescent Current vs Temperature INTVCC Voltage vs Temperature TA = 25°C, unless otherwise noted. Quiescent Current vs VIN Voltage INTVCC Dropout vs Temperature TEMPERATURE (°C) –55

14.4 VOUT (V)

14.6 14.8 15.0 15.2 0 50 100 150

3748 G01

15.4 15.6 –25 25 75 125 FIGURE 16 CIRCUIT IOUT = 150mA ON EACH OUTPUT VIN = 12V TEMPERATURE (°C) –55 QUIESCENT CURRENT (mA) 1.3 1.4 1.5 150

3748 G02

1.2 1.1 0.8 0 50 100–25 25 75 125 1.0 0.9 1.7 1.6 VSS = 0V INTVCC = OPEN VIN = 72V VIN = 36V VIN = 12V VIN = 6V VIN (V) QUIESCENT CURRENT (mA) 0.6 0.8 1.0 60 100

3748 G03

0.4 0.2 20 40 80 1.2 1.4

1.6 VSS = 0V

INTVCC = OPEN TEMPERATURE (°C) –55 6.5 INTVCC VOLTAGE (V) 6.6 6.8 6.9 7.0 7.5 7.2 0 50 75

3748 G04

6.7 7.3 7.4 7.1 –25 25 100 125 150 IINTVCC = 0mA IINTVCC = 10mA VIN VOLTAGE (V) 4 6 VINTVCC (V)6.0 6.5 7.0

3748 G05

5.5 5.0 8 20 8010 40 100 4.5 4.0 7.5 IINTVCC = 0mA IINTVCC = 10mA TEMPERATURE (°C) –55 INTVCC UVLO (V) 3.7 3.8 3.9 25 50 75 100 125

3748 G06

3.6 3.5 –25 0 150 3.4 3.3 4.0 RISING THRESHOLD FALLING THRESHOLD INTVCC CURRENT (mA) INTVCC REGULATOR DROPOUT (V) 0.5 1.0 1.5 2.0 2.5 3.0 VIN = 5V 10 20 30 40

3748 G07

150°C 100°C 25°C –50°C TEMPERATURE (°C) –55 INTVCC DROPOUT (V) 0.5 1.0 1.5 2.0 0 50 100 150

3748 G08

2.5 3.0 –25 25 75 125 VIN = 5V IINTVCC = 20mA IINTVCC = 10mA IINTVCC = 5mA TEMPERATURE (°C) –55 SOFT-START CURRENT (µA) 0 50 100 150

3748 G09

–25 25 75 125

Rev. DFor more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS Error Amplifier T ransconductance vs Temperature Error Amplifier Output Current vs RREF Pin Voltage SENSE Pin Threshold vs Temperature EN/UVLO Current vs Temperature EN/UVLO Threshold vs Temperature TC Pin Voltage vs Temperature TA = 25°C, unless otherwise noted. Maximum Discontinuous Off-Time vs Temperature GATE Rise and Fall Time vs Charge GATE Rise and Fall Time vs INTVCC Voltage TEMPERATURE (°C) –55 EN/UVLO CURRENT (µA) 0.5 1.0 1.5 2.0 0 50 100 150

3748 G10

2.5 3.0 –25 25 75 125 VEN/UVLO = 1.1V VEN/UVLO = 0.9V VEN/UVLO = 1.3V TEMPERATURE (°C) –55 EN/UVLO THRESHOLD (V) 1.20 1.30 150

3748 G11

1.10 1.00 0 50 100–25 25 75 125 1.40 1.15 1.25 1.05 1.35 TEMPERATURE (°C) –55 TC VOLTAGE (V) 0.5 0.6 0.7 150

3748 G12

0.4 0.3 0 50 100–25 25 75 125 0.2 0.1 0.9 0.8 TEMPERATURE (°C) –55

100 TRANSCONDUCTANCE (µmhos)

3748 G13

–25 25 100 125 150 VIN = 100V VIN = 6V VREF (V) IVC (µA)0 2.0

3748 G14

–20 –40 –10 –30 –50 –60 0.5 1.0 1.5 2.5 150°C 100°C 25°C –50°C TEMPERATURE (°C) –55 SENSE THRESHOLD (mV) 120 150

3748 G15

0 50 100–25 25 75 125 160 100

140 OVERCURRENT

VC = 2.2V VC = 0.2V TEMPERATURE (°C) –55 MAXIMUM DISCONTINUOUS OFF-TIME (µs) 0 50 75

3748 G16

–25 25 100 125 150 TOTAL GATE CHARGE (nC) GATE RISE AND FALL TIME (ns) AVERAGE GATE SOURCE, SINK CURRENT (A) 20 40 60 80

3748 G17

0.5 1.0 1.5 2.0 RISE TIME FALL TIME Q = C • V VINTVCC = 7V tr, tf 10% TO 90% AVERAGE CURRENT VINTVCC (V) GATE RISE AND FALL TIME (ns)

3748 G18

CGATE = 3.3nF tr, tf 10% TO 90%

Rev. D For more information www.analog.com PIN FUNCTIONS VIN (Pin 1) Input Voltage. This pin supplies current to the internal start-up circuitry and is the reference voltage for the feedback circuitry connected to the RFB pin. This pin must be locally bypassed with a capacitor . EN/UVLO (Pin 3): Enable/Undervoltage Lockout. A resis- tor divider connected to VIN is tied to this pin to program the minimum input voltage at which the LT3748 will oper- ate. At a voltage below ~0.5V, the part draws less than 1µA quiescent current. When below 1.223V but above ~0.5V, the part will draw quiescent current but will not regulate the INTVCC supply or power the gate drive cir - cuitry. Above 1.223V, all internal circuitry will start and the SS pin will source 5μA. When EN/UVLO falls below 1.223V, 2.4μA is sunk from the pin to provide program - mable hysteresis for undervoltage lockout. INTVCC (Pin 5): Gate Driver Bias Voltage. This pin supplies current to the internal gate driver circuitry of the LT3748. The INTVCC pin must be locally bypassed with a capacitor . This pin may also be connected to V IN if a third winding is not used and if V IN ≤ 20V. If a third winding is used, the INTVCC voltage should be lower than the input voltage for proper operation. GATE (Pin 6): N-Channel MOSFET Gate Driver Output. Switches between INTVCC and GND. SENSE (Pin 7): The Current Sense Input for the Control Loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor , RSENSE, in the source of the N-channel MOSFET . The negative terminal of the current sense resistor should be connected to the GND plane close to the IC. GND (Pins 8, 9): Ground. SS (Pin 10): Soft-Start Pin. This pin delays start-up and clamps VC pin voltage. Soft-start timing is set by the size of the external capacitor at the pin. Switching starts when VSS reaches ~0.65V. VC (Pin 11): Compensation Pin for the Internal Error Amplifier . Connect a series RC from this pin to ground to compensate the switching regulator . A 100pF capacitor in parallel helps eliminate noise. TC (Pin 12): Output Voltage Temperature Compensation. Connect a resistor to ground to produce a current pro - portional to absolute temperature to be sourced into the RREF node. ITC = 0.55V/RTC. RREF (Pin 14): Input Pin for the External Ground-Referred Reference Resistor . The resistor at this pin should be 6.04k, but for convenience in selecting a resistor divider ratio, the value may range from 5.76k to 6.34k. The resis- tor should be as close to the LT3748 as possible. RFB (Pin 16): Input Pin for the External Feedback Resistor . This pin is connected to the transformer primary at the external MOSFET power switch. The ratio of this resis - tor to the RREF resistor , times the internal bandgap refer- ence, determines the output voltage (plus the effect of any non-unity transformer turns ratio). The average current through this resistor during the flyback period should be approximately 200μA. The resistor should be as close to the LT3748 as possible.

Rev. DFor more information www.analog.com BLOCK DIAGRAM MASTER LATCH BOUNDARY MODE DETECT VARIABLE DELAY TIMER INTVCC CBIAS CSS RTC VOUT+ VOUT– VIN TC SS VIN SENSE VC NPS:1 5µA 20µA COUT DOUT CIN LPRI LSEC RFB RREF RSENSE RC CC INTERNAL REFERENCE AND REGULATORS CURRENT LIMIT 50µs MAX ON TIMER 50µs MAX OFF TIMER TC CURRENT R QS R S ERROR AMP 100mV +– – 2.4µA GND 8, 9 3748 BD Q2Q1 1.223V 1.223V 1.223V EN/UVLO RFB RREF GATE NMOS gm 6.04k 161

Rev. D For more information www.analog.com OPERATION The LT3748 is a current mode switching regulator con - troller designed specifically for the isolated flyback topol- ogy. The special problem normally encountered in such circuits is that information relating to the output voltage on the isolated secondary side of the transformer must be communicated to the primary side in order to main - tain regulation. Historically, this has been done with opto- isolators or extra transformer windings. Opto-isolator circuits waste output power and the extra components increase the cost and physical size of the power sup - ply. Opto-isolators can also exhibit trouble due to limited dynamic response, nonlinearity, unit-to-unit variation and aging over life. Circuits employing extra transformer windings also exhibit deficiencies. Using an extra wind - ing adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT3748 derives its information about the isolated output voltage by examining the primary-side flyback pulse waveform. In this manner , no opto-isolator nor extra transformer winding is required for regulation. The out - put voltage is easily programmed with two resistors. The LT3748 features a boundary mode control method, (also called critical conduction mode) where the part operates at the boundary between continuous conduction mode and discontinuous conduction mode. Due to the bound- ary control mode operation, the output voltage can be calculated from the transformer primary voltage when the secondary current is almost zero. This method improves load regulation without external resistors and capacitors. The Block Diagram shows an overall view of the system. Many of the blocks are similar to those found in traditional switching regulators, including current comparators, internal reference and regulators, logic, timers and an N-channel MOSFET gate driver . The novel sections include a special sampling error amplifier and a temperature com- pensation circuit. Boundary Mode Operation Boundary mode is a variable frequency, current mode switching scheme. The external N-channel MOSFET turns on and the inductor current increases until it reaches the VC pin-controlled current limit. After the external MOSFET is turned off, the voltage on the drain of the MOSFET rises to the output voltage multiplied by the primary-to-second- ary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the voltage on the drain of the MOSFET falls below VIN. A boundary mode detection comparator detects this event and turns the external MOSFET back on. Boundary mode returns the secondary current to zero every cycle, so the parasitic resistive voltage drops do not cause load regulation errors. Boundary mode also allows the use of a smaller transformer compared to continu - ous conduction mode and does not exhibit subharmonic oscillation. At low output currents the LT3748 delays turning on the external MOSFET and thus operates in discontinuous mode. Unlike traditional flyback converters, the exter - nal MOSFET has to turn on to update the output volt - age information. Below 0.6V on the V C pin, the current comparator level decreases to its minimum value and a variable delay timer waits to reset before turning on the external MOSFET . With the addition of delay before turn- ing the MOSFET back on, the part starts to operate in discontinuous mode. The average output current is able to decrease while still allowing a minimum off-time for the error amplifier sampling circuitry. The typical maximum discontinuous off-time with VC equal to 0V is 24µs.

Rev. DFor more information www.analog.com Pseudo-DC Theory of Operation The R REF and R FB resistors as depicted in the Block Diagram are external resistors used to program the out- put voltage. The LT3748 operates much the same way as traditional current mode switchers with the exception of the unique error amplifier which derives its feedback information from the flyback pulse. Operation is as follows: when the NMOS output switch turns off, its drain voltage rises above VIN. The amplitude of this flyback pulse (i.e., the difference between it and VIN) is given as: V FLBK = (VOUT + VF + ISEC • ESR) • NPS VF = DOUT forward voltage ISEC = T ransformer secondary current ESR = Total impedance of secondary circuit NPS = T ransformer effective primary-to-secondary turns ratio The flyback voltage is converted to a current by R FB and Q2. Nearly all of this current flows through resistor RREF to form a ground-referred voltage. This voltage is fed into the flyback error amplifier . The flyback error amplifier samples this output voltage information when the secondary-side winding current reaches zero. The error amplifier uses a bandgap voltage, 1.223V, as the reference voltage. The relatively high gain in the overall loop will then cause the voltage at the RREF resistor to be nearly equal to the bandgap reference voltage, VBG. The relationship between VFLBK and VBG may then be expressed as: VFLBK RFB ⎟ = VBG RREF or VFLBK = VBG RFB RREF V BG = Internal bandgap reference Combining with the previous VFLBK expression yields an expression for V OUT, in terms of the internal reference, programming resistors, transformer turns ratio and diode forward voltage drop: VOUT = VBG RFB RREF NPS ⎟ −VF −ISEC (ESR) Additionally, it includes the effect of nonzero secondary output impedance (ESR). This term can be assumed to be zero in boundary control mode. Temperature Compensation The first term in the VOUT equation does not have a tem- perature dependence, but the diode forward drop, VF , has a significant negative temperature coefficient. To compen- sate for this, a positive temperature coefficient current source is internally connected to the RREF pin. The current is set by resistor R TC to ground connected between the TC pin and ground. To cancel the temperature coefficient, the following equation is used: δVF δT = − RFB RTC

• 1 NPS
• δVTC δT or, RTC = −RFB NPS
• 1 δVF / δT • δVTC δT ≈ RFB NPS (dVF/dT) = Diode’ s forward voltage temperature coefficient (dVTC/dT) = 1.85mV/°C VTC = 0.55V The resistor value given by this equation should also be verified experimentally and adjusted, if necessary, to achieve optimal regulation over temperature. The revised output voltage is as follows: VOUT = VBG RFB RREF NPS ⎟ −VF VTC RTC ⎠⎟ •RFB NPS –ISEC (ESR) APPLICATIONS INFORMATION

Rev. D For more information www.analog.com APPLICATIONS INFORMATION Selecting Actual RREF , RFB and RTC Resistor Values The preceding equations define how the LT3748 would regulate the output voltage if the system had no time delays and no error sources. However , there are a num- ber of repeatable delays and parasitics in each applica - tion which will affect the output voltage and force a re- evaluation of the R FB and R TC component values. The following approach is the best method for selecting the correct values. The expression for VOUT, developed in the Operation sec- tion, can be rearranged to yield the following expression for RFB: RFB = RREF •NPS VOUT + VF( ) + VTC⎡⎣ ⎤⎦ VBG where: V OUT = Output voltage V F = Output diode forward voltage N PS = Effective primary-to-secondary turns ratio V TC = 0.55V The equation assumes the temperature coefficients of the output diode and VTC are equal and substitutes R FB/NPS for the value of RTC. This is a good first order approxima- tion but will be revisited later . First, the value of R REF should be approximately 6.04k since the LT3748 is trimmed and specified using this value. If the impedance of RREF varies considerably from 6.04k, additional errors will result. However , a variation in RREF of several percent is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB/RREF ratios. With starting values for R FB and R TC, an initial iteration of the application should be built with final selections of all external components (transformer , diode, MOSFET , etc.). The resulting V OUT should be measured and used to re-evaluate the value of RFB due to non-idealities in the sampling system: RFB(NEW) = VOUT(DESIRED) VOUT(MEASURED)

• RFB(OLD) With a new value of R FB selected, the temperature coef- ficient of the output diode in the application can be tested to verify the nominal RTC value. The RTC resistor should be removed from the circuit under test (this will cause VOUT to increase for this step) and V OUT should be measured over temperature at a desired target output load. It is very important for this evaluation that uniform temperature be applied to both the output diode and the LT3748—if freeze spray or a heat gun is used there can be a signifi- cant mismatch in temperature between the two devices that causes significant error . Attempting to extrapolate the data from a diode data sheet or assuming the nominal RTC value may yield a better result if there is no method to apply uniform heat or cooling such as an oven. With at least two data points (although more data points from hot to cold are recommended), the change in V/°C can be determined by: ΔVOUT ΔTEMP = VOUT1 – VOUT2 TEMP1–TEMP2 Using the measured VOUT temperature coefficient, an exact RTC value can be selected using the following equation: RTC = RFB NPS
• 1.85mV/°C ΔVOUT ΔTEMP If the value of R TC has changed significantly, which can happen with the use of some output diodes that have a very low forward drop, the R FB value may need to be changed to restore V OUT to the desired value. As in the previous iteration, after measuring V OUT , a new RFB can once again be selected using: RFB(NEW) = VOUT(DESIRED) VOUT(MEASURED)
• RFB(OLD) Once the values of RFB and RTC are selected, the regula- tion accuracy from board to board for a given application will be very consistent, typically under ±5% when includ- ing device variation of all the components in the system (assuming resistor tolerances and transformer windings matching of 1% or better). However , if the transformer , the output diode or MOSFET switch are changed or the layout is dramatically altered, there may be some change in VOUT .

GATE node high for shorter than approximately 250ns. the current control loop will lose its ability to regulate. current limitations and thermal limitations. ages and the use of any additional snubbing components.

3748 F01

Figure 1. Maximum Output Power at 12VOUT with a

in Figure 2, until parasitic losses begin to dominate. transferred through the saturated core, leading to heating. minimize losses in the output diode. the current may need to be drawn from a third winding. information should be carefully considered. ratio tolerance of 1% or better . shows the details of several of these transformers.

3748 F02

Figure 2. Maximum Output Power at 12VOUT

Table 1. Pre-Designed T ransformers—Typical Specifications Unless Otherwise Noted *2.5k isolation, others are rated for 1.5kV isolation. †TARGET APPLICATION, NOT GUARANTEED. setting total turns and mutual inductance. decreasing turns ratio can deliver more power . There are two significant constraints on the turns ratio.

some amount of overshoot caused by leakage inductance.

• 1–D( ) For a more general analysis, Figure 5 illustrates a sweep of windings ratio on the x-axis while comparing output power and estimated efficiency for a 5V output using a 48V input. If the desired application required 20W, the maximum power curve indicates that a winding ratio of 12:1 would be sufficient at a current limit of 2A (R SENSE = 0.05Ω), while a winding ratio of 5:1 would deliver the same power at 3A. However , when examining the corre- sponding efficiency at max load for those two windings ratios and current limits, the 5:1, 3A selection is clearly the superior solution with an estimated efficiency of 85% compared to 78% for the 12:1, 2A application. There are several caveats to this evaluation. First, as the diode forward voltage becomes a smaller percentage of total loss at higher output voltages (>12V) the RMS cur- rent becomes less of a concern and minimizing it will have a much smaller impact on efficiency. More significantly, if a lower turns ratio forces the use of a diode with a larger forward drop to obtain a higher reverse voltage rating, any gains from minimizing current might be lost. For low out- put voltages (3.3V or 5V) or high input voltages (>48V), a turns ratio greater than one can be used with multiple primary windings relative to the secondary to maximize the transformer’s current gain. INPUT VOL TAGE (V) OUTPUT POWER (W)

3748 F03

3748 F04

3748 F05

Figure 3. Maximum Output Power at 12V Out Using Three Figure 4. Sources of Loss In 5V, 2A Out Typical Application Figure 5. Estimated Efficiency and Output Power at 5VOUT from

age inductance should be minimized. little margin for leakage inductance spiking. below, and the resultant waveforms are shown in Figure 6.

3748 F06

Figure 6. Observed Waveforms at MOSFET Drain when value of the capacitance will decrease the overshoot.

spike if it is desirable to use a lower reverse voltage diode. on fast enough to limit the leakage inductance spike. voltage with an RC snubber and RCD clamp, respectively. below the 200V VDS(MAX) rating of the Si7464DP MOSFET . Figure 7. RCD Clamp Figure 8. Waveform of MOSFET Drain During Normal Operation

3748 F07

Figure 9. Waveform of MOSFET Drain During Normal Operation

3748 F08

Rev. DFor more information www.analog.com APPLICATIONS INFORMATION (over manufacturing variations), this can be accommo - dated by adjusting the RFB/RREF resistor ratio. Winding Resistance Effects Resistance in either the primary or secondary will reduce overall efficiency (POUT/PIN). Good output voltage regula- tion will be maintained independent of winding resistance due to the boundary mode operation of the LT3748. Bifilar Winding A bifilar , or similar winding technique, is a good way to minimize troublesome leakage inductances. However , remember that this will also increase primary-to-sec - ondary capacitance and limit the primary-to-secondary breakdown voltage, so, bifilar winding is not always prac- tical. The Analog Devices Applications group is available and extremely qualified to assist in the selection and/or design of the transformer. Selecting a Current Sense Resistor The external current sense resistor allows the user to opti- mize the current limit behavior for the particular applica- tion under consideration. As the current sense resistor is varied from several ohms down to tens of milliohms, peak switch current goes from a fraction of an ampere to tens of amperes. Care must be taken to ensure proper circuit operation, especially with small current sense resistor values. For example, a peak MOSFET switch current of 4A requires a sense resistor of 0.025Ω. Note that the instantaneous peak power in the sense resistor is 1W, and it must be rated accordingly. The LT3748 has only a single sense line to this resistor . Therefore, any parasitic resistance in the ground side connection of the sense resistor will increase its apparent value. In the case of a 0.025Ω sense resistor , 1mΩ of parasitic resistance will cause a 4% reduction in peak switch current. Therefore, resistance of printed cir- cuit copper traces and vias cannot necessarily be ignored. Another issue for proper operation of the current sense circuitry is avoiding prematurely tripping the SENSE threshold while slewing the MOSFET drain when the GATE pin goes high. The LT3748 does not begin to compare the SENSE pin voltage with the target threshold until the GATE pin is near its final value, or until at least 150ns has passed, whichever occurs more slowly. This should be entirely sufficient for most applications but premature tripping of the SENSE comparator may occur in cases where a MOSFET with very high QG is used with a series resistor at the GATE pin. Output Short Circuits and SENSE Pin Over Current The LT3748 has an internal threshold to detect when pri- mary inductor current exceeds the programmed range. This can result from an inductive output short-circuit and an output voltage below zero, reflecting a voltage back to the primary side of the transformer which, in turn, causes the LT3748 to turn the external MOSFET on before the secondary current has discharged. When the voltage at the SENSE pin exceeds approximately 130mV—equiva - lent to 30% higher than the programmed I LIM(MAX) in the RSENSE resistor—the SS pin will be reset, stopping switching. Once the soft-start capacitor is recharged and the soft-start threshold is reached, switching will resume at the minimum current limit. High Drain Capacitance and Low Current Operation When designing applications with some combination of a low current limit (I LIM < 1A), a high secondary-to-pri - mary turns ratio (NPS << 1), multiple output windings, or very capacitive output diodes, it is important to minimize the capacitance reflected onto the primary winding and on the drain of the external MOSFET . After the MOSFET turns off during each switching cycle, the primary cur - rent charges that capacitance to slew the MOSFET drain until the secondary begins to deliver power , and if the drain node does not slew and remain above V IN within approximately 200ns once the GATE pin goes low and the MOSFET turns off, the LT3748 may detect that the cur - rent in the secondary is zero and turn the MOSFET back on prematurely, causing the LT3748 to switch continu - ously while delivering very little power to the output. The result will be droop of the output voltage at lighter loads and oscillation at the VC node. This problem can be pre- vented by maximizing NPS (minimizing ratio of secondary windings to primary windings), increasing the peak drain current (minimizing RSENSE), and minimizing the output diode and transformer capacitance.

draws 2.4µA when the voltage at the pin is below 1.223V. load without decreasing efficiency in normal operation. functions as the minimum load. cuitry and reduces power dissipation in the LT3748.

3748 F10

Figure 10. Undervoltage Lockout (UVLO)

temperatures the LT3748 will stop switching. designed to output a voltage between 7.2V and 20V.

3748 F09

Figure 11. INTVCC Pin Configurations

3748 F12

Figure 12. INTVCC Current at Low VIN Can Cause the LT3748

for lower output voltages and higher output currents. the applications section should be a good starting point. Figure 13. Waveforms at LT3748 Primary Side MOSFET Drain

with a design focus of maximizing efficiency.

1. Select T ransformer Turns Ratio

• NPS) IDIODE(RMS) = √(ILIM • NPS)2 • (1 – D)/3 The equation for output power can be rearranged to solve for the current limit, ILIM, which can be solved at the nomi- nal or the minimum VIN depending on application require- ments. In this application the 2A load requirement will be set at VIN = 7.5V to reduce operating stresses at higher input voltages. The results of the aforementioned equa - tions in this application are found in Table 2. APPLICATIONS INFORMATION Evaluating the results of the table, the 1:2 turns ratio looks demanding in terms of diode reverse-voltage require - ments (a diode with higher reverse bias capability gener- ally will have a larger forward drop and therefore lower application efficiency) and primary side currents and only decreases the output diode RMS current by 13% from the 1:1 case. However , on evaluating the minimum and maximum inductance requirements in Step 3, even the 1:1 case does not allow for enough on-time from maximum VIN for the range of inductance that provides sufficient off-time. For that reason, a 2: 1 turns ratio is selected, easing the requirement on the output diode reverse volt- age rating in the process. 2. Calculate Sense Resistor Value The sense resistor can be calculated by the following equation: RSENSE = 100mV ILIM The desired 5.8A current limit leads to an unusual value of 0.0172Ω, so the current limit is increased to use a more standard 0.016Ω value and ILIM of 6.25A. 3. Select a T ransformer Based on Inductance and Saturation Current Requirements The transformer in this application will be selected to optimize efficiency at a 80kHz minimum switching frequency at maxi- mum load from the nominal input voltage. In applications where transformer size is the primary requirement, reducing the current limit or increasing the switching frequency may be required. The following equations select the inductance required for a given switching frequency at max load and then verify that the inductance is large enough to satisfy the minimum on and minimum sampling times of the LT3748.

Table 2. Voltage Stresses, Output Capability and Diode Current vs Turns Ratio in 12VIN to 5V, 2A Application

Rev. D For more information www.analog.com L PRI ≤ VIN(MIN) • (VOUT + VF(DIODE)) • NPS/(fSW(MIN) • ILIM • ((V OUT + VF(DIODE)) • NPS + VIN(MIN))) LPRI ≥ (VOUT + VF(DIODE)) • RSENSE • 400ns • NPS/15mV LPRI ≥ VIN(MAX) • RSENSE • 200ns/15mV For this application, the primary inductance with a 2 :1 transformer and a 0.016Ω sense resistor for an 6.25A current limit is bounded by the minimum desired switch- ing frequency and the minimum off time requirement to be between 9.6µH and 11.5µH. Looking at Table 1, there are no transformers that fit that exact requirement. For the sake of prototyping, a transformer with slightly less than the desired primary inductance is selected with the PA3177NL. The application will need to be tested thor - oughly for stability at higher input voltages and when the current limit is at a minimum (in the middle of the output load range). The easiest solution to ease the requirement on minimum on-time is to reduce the maximum VIN volt- age although alternatively NPS could be increased at the expense of efficiency (and requiring a more thorough redesign). 4. Select a MOSFET Switch The selected 2: 1 transformer requires a nominal 55V rating on the MOSFET switch (from Table 2), assuming no leakage inductance. However , even a small amount of leakage inductance may cause the drain to ring to double the anticipated voltage, and generally this needs to be verified in the final design. However , at currents below 10A it is fairly easy to find a MOSFET with sufficiently low RDS(ON) to be a very small contributor to maximum load efficiency losses while similarly having a low enough QG to require minimum current and minimal losses when driving the MOSFET at lighter loads. Also, while consider- ing the efficiency gains and losses with a given MOSFET , it is important to realize that a trade-off in R DS(ON) for VDS(MAX) may backfire if a snubber needs to be added to the circuit to meet the voltage requirements and dis - sipates more energy than the difference in switch resis - tance. For that reason, a Vishay Si7738 is selected to give lots of margin with its 150V rating. The RMS current in the MOSFET can be calculated, squared and multiplied by the RDS(ON) to calculate losses and the current required to drive the FET at frequency can be determined, by the following equations: IMOSFET(RMS) = √ILIM2 • D/3 IINTVCC = fSW • QG PINTVCC = IINTVCC • (VIN – VINTVCC) In this application the MOSFET RMS current at maximum load is about 2.7A, which into the 0.038Ω RDS(ON) will be 0.28W, or on the order of 2% loss in efficiency. Assuming that the maximum operating frequency is around four times higher than the maximum load frequency (at about a quarter the output load) and reading the approximate QG at 7V operation from the Vishay data sheet, the approxi- mate INTVCC current is likely close to 8mA, dissipating 0.04W when the load is on the order of 2.5W, or less than 2%, and much less at maximum load. 5. Select the Output Diode The output diode reverse voltage, as calculated earlier , is the first important specification for the output diode. As with the MOSFET , choosing a diode with enough mar- gin should preclude the use of a snubber . The second criterion is the power requirement of the diode which is more difficult to correctly ascertain—some manufactur- ers give direct data about power dissipation versus duty cycle, which can be used with the data from the table to determine. To avoid using a snubber , a diode with a 60V reverse-bias capability and minimal forward drop was selected—in this case, the Diodes Inc. SBR 8U60P5. In this particular application where maximizing efficiency is the goal, minimizing the maximum voltage requirement on VIN may allow the use of a diode with a lower reverse bias rating and a lower forward drop which could further increase efficiency. Alternatively, if no efficient diode is available for a particular reverse bias rating, it may be more beneficial to increase the windings ratio until a diode with low forward drop can be selected and then reevalu- ate whether that solution with higher RMS diode current is beneficial. APPLICATIONS INFORMATION

Rev. DFor more information www.analog.com 6. Select the Feedback Resistor for Proper Output Voltage Using the iterative process laid out earlier in the Applications Information section, select the feedback resistor RFB and program the output voltage to 5V. Adjust the RTC resistor for temperature compensation of the out- put voltage. RREF is selected as 6.04k. 7. Select the Output Capacitor The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor . The following equation calculates the output voltage ripple: ΔVMAX = LPRI •ILIM2 2•COUT •VOUT 8. Add Snubber Circuitry as Necessary With the primary components selected, the application should be constructed to evaluate ringing at the drain of the MOSFET switch and to evaluate step response to opti- mize the compensation network. If using an RC snubber , the equations from the Applications Information section can be used or a rough estimate of component values may come from using the published leakage inductance of the transformer and selecting a snubber capacitor rang- ing from 1 to 3 times larger than the published MOSFET output capacitance. In this application, the peak MOSFET drain voltage was measured at maximum load from mini- mum VIN and exceeded the 150V rating of the Si7738. A DZ clamp was considered in order to maximize efficiency APPLICATIONS INFORMATION but was unable to turn on fast enough to sufficiently clamp the very fast leakage spike. The final solution is an RC snubber , implemented iteratively, that decreases effi- ciency by less than 1% across the majority of the output load range while reducing the worst-case drain voltage spike to just 80V. Similarly, the anode of the output diode is probed to look at potential ringing when the MOSFET switch turns on and a peak of 45V is measured across the diode. Therefore, no snubber circuitry is required. 9. Optimize the Compensation Network To set the compensation, the application is first config - ured with a 22nF capacitor and 10k resistor as a starting point. A load step is applied at both light and heavy loads at the 60V maximum input voltage and the capacitance is decreased until damping decreases to the desired limit, in this case with a compensation capacitance of 2.2nF and a response implying about 60˚ of phase margin. After veri- fying stability at the minimum input voltage, as well, the compensation capacitance is doubled for safety margin. The series resistance is varied from 5k to 50k but the optimal response is observed with 24.7k. For best ripple performance, select a compensation capacitor not less than 1nF, and select a compensation resistor not greater than 50k. 10. Soft-Start Capacitor and UVLO Resistor Divider A soft-start capacitor helps during the start-up of the flyback converter . Select the UVLO resistor divider for the intended input operation range. These equations are aforementioned.

Table 3. Voltage Stresses, Output Capability and Diode Current vs Turns Ratio in 48VIN to 12V, 2A Application the results of the initial step for selecting the turns ratio. using a snubber on either device. provides a compact and efficient solution. the LT3748 to just over 1/4W at 72V VIN. might be needed, but the B360 is small and inexpensive.

Figure 16. Automotive IGBT Controller Supply

3748 F16

Figure 17. Cross Regulation Performance of the Supply in Figure 16 with VO1 and VO3 Loaded with VO2 Swept

Figure 18. ±300V Isolated Flyback Converter

3748 F18

Figure 19. 48V, 0.5A Supply from 24V to 96V Input

3748 F20

Figure 20. Efficiency of 48V Supply of Figure 17

3748 F192nF

Figure 21. 5V, 8A Isolated Supply with Synchronous Secondary-Side Rectification Using LT8309 Figure 22. Efficiency of the Supply in Figure 21 as well as Performance Using a Conventional PDS760 Schottky Rectifier

3784 TA21

Figure 25. 3.3V, 10A Isolated, Synchronous Flyback Converter Figure 26. Efficiency of the Supply in Figure 25

3748 F25

Rev. D For more information www.analog.com PACKAGE DESCRIPTION MSOP (MS12) 0213 REV B 0.53 ±0.152 (.021 ±.006) SEATING PLANE 0.18 (.007) 1.10 (.043) MAX 0.17 –/uni00A00.27 (.007 – .011) TYP 0.86 (.034) REF 1.0 (.0394) BSC 0.50 (.0197) BSC 16 14 121110 1 3 5 6 7 8 NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.254 (.010) 0° – 6° TYP DETAIL “A” DETAIL “A” GAUGE PLANE 5.10 (.201) MIN 3.20 – 3.45 (.126 – .136) 0.889 ±0.127 (.035 ±.005) RECOMMENDED SOLDER PAD LAYOUT 0.305 ±0.038 (.0120 ±.0015) TYP 0.50 (.0197) BSC 1.0 (.0394) BSC 4.039 ±0.102 (.159 ±.004) (NOTE 3) 0.1016 ±0.0508 (.004 ±.002) 3.00 ±0.102 (.118 ±.004) (NOTE 4) 0.280 ±0.076 (.011 ±.003) REF 4.90 ±0.152 (.193 ±.006) 16 (12)-Lead Plastic MSOP with 4 Pins Removed (Reference LTC DWG # 05-08-1847 Rev B)

Rev. DFor more information www.analog.com Information furnished by Analog Devices is believed to be accurate and reliable. However , no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER A 10/10 Added H-grade information to Absolute Maximum Ratings, Pin Configuration, Order Information, and Electrical Characteristics sections. Revised text and Table 2 in the Applications Information section. Revised Figures 10 and 17 in the Applications Information section. Revised Typical Application drawing. 2, 3 15, 16, 20, 22 26, 27 B 2/15 Added MP-grade device. Added Synchronous Secondary Applications paragraphs. Added Figures 21, 22, 23, 24, 25 and 26. 2, 3 29, 30, 31 C 12/21 Added AEC-Q100 Qualified for Automotive Applications to Features section. Added “AUTOMOTIVE PRODUCTS**” to Ordering Information table and supplemental text. Corrected DODE to be DIODE. Corrected Figure 17 to be Figure 19. Corrected 5V to be 4.5V. Corrected RDRAIN to be RDRAIN and inserted Additional Output Voltage Error Sources section and equations. Corrected VOUT to be (VOUT + VF(DIODE)). Corrected (VOUT+… to be ((VOUT+… with extra parenthesis and added (from Table 2), after MOSFET switch. Corrected (R9, R5) to be (R5, R7). Corrected Z1: to be Z1-Z3. Removed 1 extra wire (and 2 dots) shorting VOUT+ to VOUT–. Corrected 8:1.4 to be 8:1:4, removed 2 extra wires (and 2 extra dots) shorting VOUT+ to VOUT–, removed lines/data below 76% and corrected axis labels in Figure 26. D 02/22 Removed E-Grade Automotive option. 2

Rev. D For more information www.analog.com  ANALOG DEVICES, INC. 2010-2022 www.analog.com RELATED PARTS TYPICAL APPLICATION 5V, 2A Output from Automotive Input with Continuous Operation from 6V to 45V EN/UVLO TC SS RFB RREF VC GND INTVCC L T3748 3748 TA0286.6k 47nF VIN 12V TYP VOUT+ 5V , 2A VOUT–VIN 2:1 825k 215k 10µF 8.3µH GATE SENSE 100µF 10V 18.2Ω 2.2nF 4.7µF 6.04k 330pF 48.7k 0.016/uni03A9 D1: DIODES INC. SBR8U60P5 D2: DIODES INC. BZT52C5V6 M1: Si7738DP T1: PULSE PA3177NL 24.7k PART NUMBER DESCRIPTION COMMENTS LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT-23 LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT-23 LT8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6A Switch Low IQ Monolithic No-Opto Flybacks, 8-Lead SO-8E LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 LT3573 40V Isolated Flyback Converter Monolithic No-Opto Flyback with Integrated 1.25A, 60V Switch LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 0.65A / 2.5A 60V Switch LT3757/LT3758 40V/100V Flyback, Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/100V Flyback, Boost Converters Monolithic with Integrated 5A/3.3A Switch LT1725 20V Isolated Flyback Controller Controller with Load Compensation Circuitry LT1737 20V Isolated Flyback Controller No Opto-Isolator or Third Winding Required, Up to 50W Output LT C 3803/LTC3803-3 LTC3803-5 200kHz/300kHz Flyback DC/DC Controllers VIN and VOUT Limited Only by External Components LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited Only by External Components