LT3511

LT8300 AD | Alldatasheet

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Rev. AFor more information www.analog.com Document Feedback TYPICAL APPLICATION FEATURES DESCRIPTION 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch The LT®8300 is a micropower high voltage isolated fly - back converter . By sampling the isolated output voltage directly from the primary-side flyback waveform, the part requires no third winding or opto-isolator for regulation. The output voltage is programmed with a single exter - nal resistor . Internal compensation and soft-start further reduce external component count. Boundary mode opera- tion provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 260mA, 150V DMOS power switch is integrated along with all high voltage circuitry and control logic into a 5-lead ThinSOT™ package. The LT8300 operates from an input voltages range of 6V to 100V and can deliver up to 2W of isolated output power . The high level of integration and the use of boundary and low ripple burst modes result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. 5V Micropower Isolated Flyback Converter

APPLICATIONS

n 6V to 100V Input Voltage Range n 260mA, 150V Internal DMOS Power Switch n Low Quiescent Current n 70µA in Sleep Mode n 330µA in Active Mode n Boundary Mode Operation at Heavy Load n Low-Ripple Burst Mode® Operation at Light Load n Minimum Load <0.5% (Typ) of Full Output n VOUT Set with a Single External Resistor n No T ransformer Third Winding or Opto-Isolator Required for Regulation n Accurate EN/UVLO Threshold and Hysteresis n Internal Compensation and Soft-Start n 5-Lead TSOT-23 Package n AEC-Q100 Qualified for Automotive Applications n Isolated Telecom, Automotive, Industrial, Medical Power Supplies n Isolated Auxiliary/Housekeeping Power Supplies Efficiency vs Load Current LT8300 4:1 RFB SW 300µH 19µH EN/UVLO 2.2µF 40.2k VIN VIN 36V TO 72V VOUT+ 1mA TO 300mA VOUT– GND 210k 47µF

8300 TA01a

LOAD CURRENT (mA) EFFICIENCY (%) 100

8300 TA01b

VIN = 48V VIN = 72V VIN = 36V All registered trademarks and trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, and 7471522.

Rev. A For more information www.analog.com PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS Operating Junction Temperature Range (Notes 3, 4) (Note 1) EN/UVLO 1 GND 2 TOP VIEW S5 PACKAGE 5-LEAD PLASTIC TSOT-23 TJMAX = 150°C, θJA = 150°C/W RFB 3

5 VIN

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8300ES5#PBF LT8300ES5#TRPBF L TGFF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8300IS5#PBF LT8300IS5#TRPBF L TGFF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8300HS5#PBF LT8300HS5#TRPBF L TGFF 5-Lead Plastic TSOT-23 –40°C to 150°C LT8300MPS5#PBF LT8300MPS5#TRPBF L TGFF 5-Lead Plastic TSOT-23 –55°C to 150°C AUTOMOTIVE PRODUCTS** LEAD FREE FINISH TAPE AND REEL TAPE AND REEL MINI PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8300ES5#WPBF LT8300ES5#WTRPBF LT8300ES5#WTRMPBF L TGFF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8300IS5#WPBF LT8300IS5#WTRPBF LT8300IS5#WTRMPBF L TGFF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8300HS5#WPBF LT8300HS5#WTRPBF LT8300HS5#WTRMPBF L TGFF 5-Lead Plastic TSOT-23 –40°C to 150°C Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container . Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. **Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for these models.

Rev. AFor more information www.analog.com

ELECTRICAL CHARACTERISTICS

Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 150V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 150V as shown in Figure 5. Note 3: The LT8300E is guaranteed to meet performance specifications from 0°C to 125°C operating junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT VIN Input Voltage Range 6 100 V VIN UVLO Threshold Rising Falling 5.8 3.2 6 V V IQ VIN Quiescent Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 1.2 200 330 2 µA µA µA µA EN/UVLO Shutdown Threshold For Lowest Off IQ l 0.3 0.75 V EN/UVLO Enable Threshold Falling Hysteresis l 1.199 1.223 0.016 1.270 V V IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.2 –0.1 2.5 0.1 2.8 0.1 µA µA µA fMAX Maximum Switching Frequency 720 750 780 kHz fMIN Minimum Switching Frequency 6 7.5 9 kHz tON(MIN) Minimum Switch-On Time 160 ns tOFF(MIN) Minimum Switch-Off Time 350 ns tOFF(MAX) Maximum Switch-Off Time Backup Timer 200 µs ISW(MAX) Maximum SW Current Limit l 228 260 292 mA ISW(MIN) Minimum SW Current Limit l 34 52 70 mA SW Over Current Limit To Initiate Soft-Start 520 mA RDS(ON) Switch On-Resistance ISW = 100mA 10 Ω ILKG Switch Leakage Current VIN = 100V, VSW = 150V 0.1 0.5 µA IRFB RFB Regulation Current l 98 100 102 µA RFB Regulation Current Line Regulation 6V ≤ VIN ≤ 100V 0.001 0.01 %/V tSS Soft-Start Timer 2.7 ms The LT8300I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8300H is guaranteed over the full –40°C to 150°C operating junction temperature range. The LT8300MP is guaranteed over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125°C. Note 4: The LT8300 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.

Rev. A For more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms VIN Shutdown Current VIN Quiescent Current, Sleep Mode VIN Quiescent Current, Active Mode Output Load and Line Regulation Output Temperature Variation Switching Frequency vs Load Current TA = 25°C, unless otherwise noted. LOAD CURRENT (mA) OUTPUT VOL TAGE (V) 5.20 4.85 5.15 4.90 5.00 5.05 5.10 4.95 4.80

8300 G01

VIN = 36V VIN = 48V VIN = 72V FRONT PAGE APPLICATION AMBIENT TEMPERATURE (°C) OUTPUT VOL TAGE (V) 5.5 4.6 5.4 4.7 5.0 5.1 5.2 5.3 4.8 4.9 4.5

8300 G02

1507550 125100250–25–50 FRONT PAGE APPLICATION VIN = 48V , IOUT = 200mA LOAD CURRENT (mA) FREQUENCY (kHz) 500 100 200 400 300

8300 G03

VIN = 48V 2µs/DIV FRONT PAGE APPLICATION VIN = 48V , IOUT = 300mA ILPRI 100mA/DIV VSW 50V/DIV VOUT 50mV/DIV

8300 G04

2µs/DIV FRONT PAGE APPLICATION VIN = 48V , IOUT = 60mA ILPRI 100mA/DIV VSW 50V/DIV VOUT 50mV/DIV

8300 G05 20µs/DIV

VIN = 48V , IOUT = 1mA ILPRI 100mA/DIV VSW 50V/DIV VOUT 50mV/DIV

8300 G06

VIN (V) IQ (µA)

8300 G07

TJ = –55°C TJ = 150°C TJ = 25°C VIN (V) IQ (µA) 100

8300 G08

TJ = 25°C TJ = –55°C TJ = 150°C VIN (V) IQ (µA) 380 300 320 360 340 280

8300 G09

TJ = 25°C TJ = –55°C TJ = 150°C

Rev. AFor more information www.analog.com TYPICAL PERFORMANCE CHARACTERISTICS RDS(ON) Switch Current Limit Maximum Switching Frequency Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time EN/UVLO Enable Threshold EN/UVLO Hysteresis Current RFB Regulation Current TA = 25°C, unless otherwise noted. TEMPERATURE (°C) VEN/UVLO (V) 1.240 1.205 1.210 1.225 1.230 1.235 1.215 1.220 1.200

8300 G10

1507550 125100250–25–50 TEMPERATURE (°C) IHYS (µA)

8300 G11

1507550 125100250–25–50 TEMPERATURE (°C) IRFB (µA) 105 101 102 103 104 100

8300 G12

1507550 125100250–25–50 TEMPERATURE (°C) RESISTANCE (Ω)

8300 G13

1507550 125100250–25–50 TEMPERATURE (°C) ISW (mA) 300 100 150 200 250

8300 G14

1507550 125100250–25–50 MAXIMUM CURRENT LIMIT MINIMUM CURRENT LIMIT TEMPERATURE (°C) FREQUENCY (kHz) 1000 200 400 600 800

8300 G15

1507550 125100250–25–50 TEMPERATURE (°C) FREQUENCY (kHz)

8300 G16

1507550 125100250–25–50 TEMPERATURE (°C) TIME (ns) 400 100 200 300

8300 G17

1507550 125100250–25–50 TEMPERATURE (°C) TIME (ns) 400 100 200 300

8300 G18

1507550 125100250–25–50

Rev. A For more information www.analog.com PIN FUNCTIONS EN/UVLO (Pin 1) : Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8300. Pull the pin below 0.3V to shut down the LT8300. This pin has an accurate 1.223V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resis- tor divider from VIN to ground. A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. GND (Pin 2): Ground. Tie this pin directly to local ground plane. RFB (Pin 3) : Input Pin for External Feedback Resistor . Connect a resistor from this pin to the transformer primary SW pin. The ratio of the RFB resistor to the inter- nal trimmed 12.23k resistor , times the internal bandgap reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. SW (Pin 4) : Drain of the 150V Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. VIN (Pin 5) : Input Supply. The V IN pin supplies current to internal circuitry and serves as a reference voltage for the feedback circuitry connected to the R FB pin. Locally bypass this pin to ground with a capacitor . BLOCK DIAGRAM 8300 BD OSCILLATOR 1:4 S R Q1.223V 25µA M2M3 BOUNDARY DETECTOR DRIVER RSENSE 0.3Ω gmRREF 12.23kΩ RFB 2.5µA EN/UVLO 3 45 +1.223V REFERENCE REGULATORS VIN 2GND RFB SWVIN VIN NPS:1 DOUT LSECLPRI VOUT+ VOUT– COUT CIN

Rev. AFor more information www.analog.com OPERATION The LT8300 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation bound- ary. Opto-isolator circuits waste output power , and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unit-to-unit variation and aging over lifetime. Circuits employing extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT8300 samples the isolated output voltage through the primary-side flyback pulse waveform. In this man - ner , neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8300 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method improves load regulation without the need of external load compensation components. The LT8300 is a simple to use micropower isolated fly - back converter housed in a 5-lead TSOT-23 package. The output voltage is programmed with a single external resis- tor . By integrating the loop compensation and soft-start inside, the part further reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switch- ing regulators including reference, regulators, oscillator , logic, current amplifier , current comparator , driver , and power switch. The novel sections include a flyback pulse sense circuit, a sample-and-hold error amplifier , and a boundary mode detector , as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Boundary Conduction Mode Operation The LT8300 features boundary conduction mode opera- tion at heavy load, where the chip turns on the primary power switch when the secondary current is zero. Boundary conduction mode is a variable frequency, vari- able peak-current switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around V IN. A boundary mode detector senses this event and turns the power switch back on. Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduc - tion mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit subharmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter , boundary conduction mode increases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switch- ing and gate charge losses. To avoid this scenario, the LT8300 has an additional internal oscillator , which clamps the maximum switching frequency to be less than 750kHz. Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8300 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch cur- rent limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8300 starts to fold back the switching frequency while keeping the mini - mum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sample-and-hold error amplifier . Meanwhile, the part switches between sleep mode and active mode, thereby

Rev. A For more information www.analog.com OPERATION reducing the effective quiescent current to improve light load efficiency. In this condition, the LT8300 operates in low ripple Burst Mode. The typical 7.5kHz minimum Output Voltage The RFB resistor as depicted in the Block Diagram is the only external resistor used to program the output voltage. The LT8300 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier , which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the V IN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: V FLBK = (VOUT + VF + ISEC • ESR) • NPS V F = Output diode forward voltage I SEC = T ransformer secondary current ESR = Total impedance of secondary circuit N PS = T ransformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current IRFB by the flyback pulse sense circuit (M2 and M3). This current IRFB also flows through the internal trimmed 12.23k RREF resistor to generate a ground-referred voltage. The result- ing voltage feeds to the inverting input of the sample- and-hold error amplifier . Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the VFLBK equation can be assumed to be zero. The bandgap reference voltage VBG, 1.223V, feeds to the non-inverting input of the sample-and-hold error ampli - fier . The relatively high gain in the overall loop causes the voltage across RREF resistor to be nearly equal to the APPLICATIONS INFORMATION switching frequency determines how often the output volt- age is sampled and also the minimum load requirement. bandgap reference voltage VBG. The resulting relationship between VFLBK and VBG can be expressed as: VFLBK RFB ⎟•RREF =VBG or VFLBK = VBG RREF ⎟•RFB =IRFB •RFB V BG = Bandgap reference voltage I RFB = RFB regulation current = 100µA Combination with the previous V FLBK equation yields an equation for V OUT, in terms of the R FB resistor , trans- former turns ratio, and diode forward voltage: VOUT =100µA • RFB NPS ⎟−VF Output Temperature Coefficient The first term in the V OUT equation does not have tem - perature dependence, but the output diode forward volt- age VF has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coef- ficient produces approximately 200mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible effect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however , the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. For customers requiring tight output voltage regulation across temperature, please refer to other L TC parts with integrated temperature compensation features.

Rev. AFor more information www.analog.com APPLICATIONS INFORMATION Selecting Actual RFB Resistor Value The LT8300 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evalua- tion of the RFB resistor value. Therefore, a simple two-step process is required to choose feedback resistor RFB. Rearrangement of the expression for VOUT in the Output Voltage section yields the starting value for RFB: RFB = NPS • VOUT +VF( ) 100µA V OUT = Output voltage V F = Output diode forward voltage = ~0.3V N PS = T ransformer effective primary-to-secondary turns ratio Power up the application with the starting RFB value and other components connected, and measure the regulated output voltage, V OUT(MEAS). The final R FB value can be adjusted to: RFB(FINAL) = VOUT VOUT(MEAS)

RFB Once the final RFB value is selected, the regulation accu- racy from board to board for a given application will be very consistent, typically under ±5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ±1%). However , if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. Output Power A flyback converter has a complicated relationship between the input and output currents compared to a buck or a boost converter . A boost converter has a rela- tively constant maximum input current regardless of input voltage and a buck converter has a relatively constant maximum output current regardless of input voltage. This is due to the continuous non-switching behavior of the two currents. A flyback converter has both discontinu - ous input and output currents which make it similar to a non-isolated buck-boost converter . The duty cycle will affect the input and output currents, making it hard to predict output power . In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. The graphs in Figures 1 to 4 show the typical maximum output power possible for the output voltages 3.3V, 5V, 12V, and 24V. The maximum output power curve is the calculated output power if the switch voltage is 120V dur- ing the switch-off time. 30V of margin is left for leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 120V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 36V and a maximum input voltage of 72V. A six-to-one winding ratio fits this design example perfectly and outputs equal to 2.44W at 72V but lowers to 1.87W at 36V. The following equations calculate output power: POUT =η• VIN •D •ISW(MAX) • 0.5 η=Efficiency =85% D =Duty Cycle = VOUT +VF( )•NPS VOUT +VF( )•NPS +VIN I SW(MAX) = Maximum switch current limit = 260mA

Figure 4. Output Power for 24V Output Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output Figure 3. Output Power for 12V Output

8300 F01

8300 F02

8300 F03

8300 F04

the power switch shorter than approximately 160ns.

and may cause instability at light load. following information should be carefully considered. shows the details of these transformers. Table 5. Predesigned T ransformers — Typical Specifications

All the transformers are rated for 1.5kV Isolation.

Rev. A For more information www.analog.com Typically, choose the transformer turns ratio to maximize available output power . For low output voltages (3.3V or 5V), a larger N:1 turns ratio can be used with multiple pri- mary windings relative to the secondary to maximize the transformer’s current gain (and output power). However , remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 150V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure: NPS < 150V −VIN(MAX) −VLEAKAGE VOUT +VF For lower output power levels, choose a smaller N:1 turns ratio to alleviate the SW pin voltage stress. Although a 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch, the multiplied parasitic capacitance through turns ratio coupled with the relatively resistive 150V internal power switch may cause the switch turn- on current spike ringing beyond 160ns leading-edge blanking, thereby producing light load instability in cer - tain applications. So any 1 :N turns ratio should be fully evaluated before its use with the LT8300. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer speci- fies turns ratio accuracy within ±1%. APPLICATIONS INFORMATION Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8300, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output volt - age regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction mode operation of the LT8300. Leakage Inductance and Snubbers T ransformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. It is very impor- tant to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus VIN should be kept below 120V. This leaves at least 30V margin for the leakage spike across line and load condi - tions. A larger voltage margin will be required for poorly wound transformers or for excessive leakage inductance. In addition to the voltage spikes, the leakage inductance also causes the SW pin ringing for a while after the power switch turns off. To prevent the voltage ringing falsely trig- ger boundary mode detector , the LT8300 internally blanks the boundary mode detector for approximately 250ns. Any remaining voltage ringing after 250ns may turn the power switch back on again before the secondary cur - rent falls to zero. So the leakage inductance spike ringing should be limited to less than 250ns.

A snubber circuit is recommended for most applications. Zener) snubber and the RC (resistor-capacitor) snubber . compromise is to choose the largest voltage breakdown. around 72V and below the 78V maximum. highest at maximum load and minimum input voltage. Figure 5. Maximum Voltages for SW Pin Flyback Waveform Figure 6. Snubber Circuits

8300 F05

8300 F06b8300 F06a

Table 2. Recommended Zener Diodes Table 3. Recommended Diodes Figure 7. Undervoltage Lockout (UVLO)

8300 F07

converted to heat and will not be delivered to the load. may need to be sized for thermal dissipation. falling threshold is set at 1.223V with 16mV hysteresis. LT8300 in shutdown with quiescent current less than 2µA.

serve as a minimum load if pre-loading is not acceptable. input range from 36V to 72V. Step 1: Select the T ransformer Turns Ratio. bility at different transformer turns ratio. Table 4. Switch Voltage Stress and Output Current Capability

Rev. A For more information www.analog.com APPLICATIONS INFORMATION Step 2: Determine the Primary Inductance. Primary inductance for the transformer must be set above a minimum value to satisfy the minimum switch-off and switch-on time requirements: LPRI ≥ tOFF(MIN) •NPS • VOUT +VF( ) ISW(MIN) LPRI ≥ tON(MIN) • VIN(MAX) ISW(MIN) t OFF(MIN) = 350ns t ON(MIN) = 160ns I SW(MIN) = 52mA Example: LPRI ≥ 350ns • 2 •(12V +0.3V) 52mA =166µH LPRI ≥160ns • 72V 52mA =222µH Most transformers specify primary inductance with a tol- erance of ±20%. With other component tolerance consid- ered, choose a transformer with its primary inductance 20% to 40% larger than the minimum values calculated above. LPRI = 300µH is then chosen in this example. Once the primary inductance has been determined, the maximum load switching frequency can be calculated as: fSW = 1 tON +tOFF = 1 LPRI •ISW VIN + LPRI •ISW NPS •(VOUT +VF ) ISW = VOUT •IOUT • 2 η• VIN •D Example: D = (12V +0.3V)• 2 (12V +0.3V)• 2+48V =0.34 ISW = 12V • 0.12A • 2 0.85 • 48V • 0.34 =0.21A fSW =260kHz The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 400mA is necessary to work with the LT8300. The 10396-T022 from Sumida is chosen as the flyback transformer . Step 3: Choose the Output Diode. T wo main criteria for choosing the output diode include forward current rating and reverse voltage rating. The maximum load requirement is a good first-order guess as the average current requirement for the output diode. A conservative metric is the maximum switch current limit multiplied by the turns ratio, I DIODE(MAX) = ISW(MAX) • NPS Example: I DIODE(MAX) = 0.52A Next calculate reverse voltage requirement using maxi - mum VIN: VREVERSE =VOUT + VIN(MAX) NPS Example: VREVERSE =12V +72V =48V The SB R0560S1 ( 0.5A, 60V diode) from Diodes Inc. is chosen.

Rev. AFor more information www.analog.com Step 4: Choose the Output Capacitor . The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor . Use the equation below to calculate the output capacitance: COUT = LPRI •ISW 2 • VOUT • ΔVOUT Example: Design for output voltage ripple less than 1% of V OUT, i.e., 120mV. COUT = 300µH •(0.21A)2 2 •12V • 0.12V =4.6µF Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop to 40% of quoted capacitance at the maximum voltage rating. So a 10uF , 16V rating ceramic capacitor is chosen. Step 5: Design Snubber Circuit. The snubber circuit protects the power switch from leak- age inductance voltage spike. A DZ snubber is recom - mended for this application because of lower leakage inductance and larger voltage margin. The Zener and the diode need to be selected. The maximum Zener breakdown voltage is set according to the maximum VIN: V ZENER(MAX) ≤ 150V – VIN(MAX) Example: V ZENER(MAX) ≤ 150V – 72V = 78V APPLICATIONS INFORMATION A 68V Zener with a maximum of 72V will provide optimal protection and minimize power loss. So a 68V, 0.5W Zener from On Semiconductor (MMSZ5266BT1G) is chosen. Choose a diode that is fast and has sufficient reverse volt- age breakdown: V REVERSE > VSW(MAX) V SW(MAX) = VIN(MAX) + VZENER(MAX) Example: V REVERSE > 144V A 150V, 0.6A diode from Diodes Inc. (BAV20W) is chosen. Step 6: Select the RFB Resistor . Use the following equation to calculate the starting value for RFB: RFB = NPS •(VOUT +VF ) 100µA Example: RFB = 2 •(12V +0.3V) 100µA =246k Depending on the tolerance of standard resistor values, the precise resistor value may not exist. For 1% standard values, a 243k resistor in series with a 3.01k resistor should be close enough. As discussed in the Application Information section, the final RFB value should be adjusted on the measured output voltage.

Rev. A For more information www.analog.com APPLICATIONS INFORMATION Step 7: Select the EN/UVLO Resistors. Determine the amount of hysteresis required and calcu - late R1 resistor value: V IN(HYS) = 2.5µA • R1 Example: Choose 2.5V of hysteresis, R1 = 1M Determine the UVLO thresholds and calculate R2 resistor value: VIN(UVLO+) =1.239V •(R1+R2) +2.5µA •R1 Example: Set VIN UVLO rising threshold to 34.5V, R2 = 40.2k V IN(UVLO+) = 34.1V V IN(UVLO–) = 31.6V Step 8: Ensure minimum load. The theoretical minimum load can be approximately esti- mated as: ILOAD(MIN) = 300µH •(52mA)2 • 7.5kHz 2 •12V =0.25mA Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the converter delivers more energy than what is consumed at the out - put. The real minimum load for this application is about 0.6mA, 0.5% of 120mA maximum load. In this example, a 20k resistor is selected as the minimum load.

Rev. AFor more information www.analog.com TYPICAL APPLICATIONS LT8300 6:1 RFB SW 300µH 8µH EN/UVLO 2.2µF 40.2k VIN VIN 36V TO 72V VOUT+ 1mA TO 330mA VOUT– GND 316k T1: WURTH 750312557 D1: DIODES INC. SBR2A30P1 47µF

8300 TA02

5V Micropower Isolated Flyback Converter 12V Micropower Isolated Flyback Converter LT8300 2:1 RFB SW 300µH 75µH EN/UVLO 2.2µF 40.2k VIN VIN 36V TO 72V VOUT+ 12V 0.6mA TO 120mA VOUT– GND 243k T1: SUMIDA 10396-TO22 D1: DIODES INC. SBR0560S1 10µF

8300 TA03

Rev. A For more information www.analog.com TYPICAL APPLICATIONS LT8300 8:1 RFB SW 400µH 6µH EN/UVLO 2.2µF 40.2k VIN VIN 36V TO 72V VOUT+ 3.3V 2mA TO 440mA VOUT– GND 287k T1: WURTH 750312367 D1: NXP PMEG2020EH 100µF

8300 TA05

24V Micropower Isolated Flyback Converter 3.3V Micropower Isolated Flyback Converter LT8300 1:1 RFB SW 300µH 300µH EN/UVLO 2.2µF 40.2k VIN VIN 36V TO 72V VOUT+ 24V 0.3mA TO 60mA VOUT– GND 243k T1: WURTH 750311559 D1: DIODES DFLS 1200-7 4.7µF

8300 TA04

Rev. AFor more information www.analog.com TYPICAL APPLICATIONS VIN to (VIN + 10V) Micropower Converter VIN to (VIN – 10V) Micropower Converter LT8300 D1 RFB SW 330µH EN/UVLO 1µF 118k VIN VIN 15V TO 80V VOUT+ 10V 100mA VOUT– GND 102k L1: COILTRONICS DR73-331-R D1: DIODES INC. SBR1U150SA Z1: CENTRAL CMDZ12L

8300 TA07

4.7µF LT8300 D1 RFB SW 330µH EN/UVLO 1µF 118k VIN VIN 15V TO 80V VOUT+ 10V 50mA VOUT– GND 102k L1: COILTRONICS DR73-331-R D1: DIODES INC. SBR1U150SA Z1: CENTRAL CMDZ12L

8300 TA06

4.7µF

Rev. A For more information www.analog.com PACKAGE DESCRIPTION 5-Lead Plastic TSOT-23 (Reference L TC DWG # 05-08-1635 Rev B) 1.50 – 1.75 (NOTE 4)2.80 BSC 0.30 – 0.45 TYP

5 PLCS (NOTE 3)

DATUM ‘A’ 0.09 – 0.20 (NOTE 3) S5 TSOT-23 0302 REV B PIN ONE

2.90 BSC

(NOTE 4)

0.95 BSC

1.90 BSC

0.80 – 0.90

1.00 MAX

0.01 – 0.100.20 BSC 0.30 – 0.50 REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193

3.85 MAX

0.62 MAX 0.95 REF RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 1.4 MIN2.62 REF

1.22 REF

Rev. AFor more information www.analog.com Information furnished by Analog Devices is believed to be accurate and reliable. However , no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER A 06/19 Added Automotive Part Numbers 2

Rev. A For more information www.analog.com  ANALOG DEVICES, INC. 2012-2019 www.analog.com RELATED PARTS TYPICAL APPLICATION 3.3V Isolated Converter (Conforming to DEF-STAN61-5) PART NUMBER DESCRIPTION COMMENTS LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing LT3798 Off-Line Isolated No Opto-Coupler Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3757/LT3759/LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch LTC3803/LTC3803-3/ LTC3803-5 200kHz/300kHz Flyback Controllers in SOT-23 VIN and VOUT Limited by External Components LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited by External Components VIN (V) IVIN (µA) 400 200 300 100

8300 TA08b

1:1 RFB SW 150µH 150µH EN/UVLO 1µF 93.1k VIN VIN 18V TO 32V VOUT+ 3.3V 0mA TO 20mA VOUT– GND 42.2k D1: DIODES INC. SBR0560S1-7 L1: DRQ73-151-R Z1: CENTRAL CMDZ4L7 1µF 1µFZ1

8300 TA08a

LT3009-3.3 GND SHDN IN OUT Input Current with No Load

Summary

100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch 06/19-REV.A

Description