LT1941EFE

LT1941 LINER | Alldatasheet

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, LTC and LT are registered trademarks of Linear Technology Corporation. step-down converters with 3A and 2A power switches. ing with microcontrollers and DSPs. batteries and distributed-power supplies. Figure 1. Triple Output Power Supply: 3.3V, 1.8V, –12V

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V IN VIN SW1 SW1 BOOST1 PGOOD1 VC1 FB1 PGOOD2 VC2 FB2 RUN/SS1 RUN/SS2 RUN/SS3 BIAS2 SW3 PGND V IN BOOST2 SW2 V IN PGOOD3 FB3 NFB V 5GOOD 12GOOD BIAS1 ORDER PART NUMBER TJMAX = 125°C, qJA = 25°C/ W EXPOSED PAD (PIN 29) IS GND MUST BE SOLDERED TO PCB The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN, VBIAS1, VBIAS2 = 5V, VBOOST1, VBOOST2 = 8V, unless otherwise noted. (Note 2) (Note 1) RUN/SS, V Operating Ambient Temperature Range LT1941EFE PARAMETER CONDITIONS MIN TYP MAX UNITS Minimum Operating Voltage l 3.1 3.5 V VIN Quiescent Current Not Switching 2 3.5 mA BIAS1 Quiescent Current Not Switching 5 7.5 mA BIAS2 Quiescent Current Not Switching 1.6 2.2 mA Shutdown Current V RUNSS1,2,3 = 0V 50 75 mA Reference Voltage Line Regulation 5V < V IN < 25V 0.01 %/V VC Source Current V C = 0.6V 100 mA VC Sink Current V C = 0.6V 100 mA VC Clamp Voltage 1.7 V Switching Frequency l 0.9 1.1 1.35 MHz Switching Phase SW1 to SW2 150 180 210 Deg SW1 to SW3 –30 0 30 Deg Foldback Frequency V FB = 0V 200 kHz RUN/SS Current 123 mA RUN/SS Threshold 0.4 0.6 V 5GOOD Threshold V IN Rising 4.5 V 5GOOD Voltage Output Low I 5GOOD = 125mA, VIN = 4V 0.2 0.4 V 5GOOD Leakage V 5GOOD = 2V 10 400 nA 12GOOD Threshold V IN Rising 10.8 V ABSOLUTE AXI U RATI GSW WW U PACKAGE/ORDER I FOR ATIOUU W

ELECTRICAL CHARACTERISTICS

Consult LTC Marketing for parts specified with wider operating temperature ranges.

The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN, VBIAS1, VBIAS2 = 5V, VBOOST1, VBOOST2 = 8V, unless otherwise noted. (Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS 12GOOD Voltage Output Low I 12GOOD = 125mA 0.2 0.4 V 12GOOD Leakage V 12GOOD = 2V, VIN = 12V 10 400 nA PGOOD Voltage Output Low I PGOOD = 200mA 0.2 0.4 V PGOOD Pin Leakage V PGOOD = 2V 10 400 nA 3A Step-Down FB1 Voltage 618 628 638 mV l 613 638 mV FB1 Pin Bias Current l 50 500 nA PGOOD1 Threshold Offset V FB Rising 54 mV Frequency Shift Threshold on FB1 0.35 V Error Amplifier Transconductance 1700 mMhos Error Amplifier Voltage Gain 500 V/V VC Switching Threshold 0.9 V VC1 to Switch Current Gain 5A / V Switch 1 Current Limit (Note 3) V IN = 12V, VBOOST1, VBOOST2 = 15V l 3 4.3 6 A Switch 1 VCESAT (Note 7) I SW = 2.5A 400 600 mV BOOST1 Pin Current I SW = 2.5A 40 60 mA Switch 1 Leakage Current 0.01 10 mA Minimum Boost Voltage Above Switch (Note 4) 1.8 2.5 V Maximum Duty Cycle l 78 88 % 2A Step-Down FB2 Voltage 618 628 638 mV l 613 638 mV FB2 Pin Bias Current l 50 500 nA PGOOD2 Threshold Offset V FB Rising 54 mV Frequency Shift Threshold on FB2 0.35 V Error Amplifier Transconductance 1700 mMhos Error Amplifier Voltage Gain 500 V/V VC Switching Threshold 0.9 V VC2 to Switch Current Gain 3.6 A/V Switch 2 Current Limit (Note 3) V IN = 12V, VBOOST1, VBOOST2 = 15V l 2 2.9 4.1 A Switch 2 VCESAT (Note 7) I SW = 1.5A 450 600 mV BOOST2 Pin Current I SW = 1.5A 26 40 mA Switch 2 Leakage Current 0.01 10 mA Minimum Boost Voltage Above Switch (Note 4) 1.8 2.5 V Maximum Duty Cycle l 78 88 % 1.5A Inverting/Boost FB3 Voltage 1.23 1.25 1.27 V l 1.22 1.27 V FB3 Pin Bias Current l 800 1400 nA NFB Voltage l –15 0 15 mV NFB Pin Bias Current l 60 500 nA

The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN, VBIAS1, VBIAS2 = 5V, VBOOST1, VBOOST2 = 8V, unless otherwise noted. (Note 2) PARAMETER CONDITIONS MIN TYP MAX UNITS NFB4 Voltage (VFB4-VNVB4) 1.212 1.24 1.258 V l 1.205 1.260 V FB3 Pin Output Current V FB3 = 1.35V, VNFB = –0.1V l 150 350 mA PGOOD3 Threshold Offset V FB Rising 120 mV Error Amplifier Transconductance 800 mMhos Error Amplifier Voltage Gain 150 V/V VC Switching Threshold 1.1 V VC3 to Switch Current Gain 5 A/V Frequency Shift Threshold on FB3 0.65 V Switch 3 Current Limit (Note 5) l 1.5 2 2.9 A Switch 3 VCESAT ISW = 1A 240 320 mV BIAS2 Pin Current I SW = 1A 30 45 mA Switch 3 Leakage Current 0.01 10 mA Maximum Duty Cycle l 77 86 % Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LT1941E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 5: Current limit is guaranteed by design and/or correlation to static test. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 7: Guaranteed by design, not 100% tested. TYPICAL PERFOR A CE CHARACTERISTICS UW Efficiency, VOUT1 = 1.8V LOAD CURRENT (A) EFFICIENCY (%)

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0.5 1 1.5 2.5

90 VIN = 5V

TA = 25°C Efficiency, VOUT2 = 3.3V Efficiency, V OUT3 = –12V LOAD CURRENT (A) EFFICIENCY (%) 0.25 0.5 0.75 1

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1.25 1.5 VIN = 5V TA = 25°C LOAD CURRENT (mA) EFFICIENCY (%) 50 100 150 200

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VIN = 5V TA = 25°C

TYPICAL PERFOR A CE CHARACTERISTICS UW BOOST1 Pin Current V FB3 vs Temperature SW2 Current Limit vs Duty Cycle SW1 PIN CURRENT (A) BOOST CURRENT (mA) 0.5 1 1.5 2

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2.5 3 TA = 25°C TEMPERATURE (°C) –50

1.220 VFB (V)

1.235 1.250 1.265 1.280 –25 0 25 50

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DUTY CYCLE (%) CURRENT LIMIT (A) 0.5 1.0 1.5 2.0 2.5 3.0 20 40 60 80

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SWITCH CURRENT (A) SWITCH VOLTAGE (mV) 100 200 300 400 600 500 0.5 1

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1.5 2 TA = 25°C SW3 VCESAT BOOST2 Pin Current SWITCH CURRENT (A) SWITCH VOLTAGE (mV) 100 200 300 400 500

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TA = 25°C SW2 PIN CURRENT (A) BOOST CURRENT (mA)

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0.5 1 1.5

40 TA = 25°C

VFB1, VFB2 vs Temperature TEMPERATURE (°C) –50

0.605 VFB (V)

0.615 0.625 0.635 0.645 –25 0 25 50

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SW1 Current Limit vs Duty Cycle SW1 VCESAT SWITCH CURRENT (A) SWITCH VOLTAGE (mV) 100 200 300 400 500 0.5 1 1.5 2

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2.5 3 TA = 25°C DUTY CYCLE (%) CURRENT LIMIT (A) 3.0 4.0 5.0

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2.0 1.0 2.5 3.5 4.5 1.5 0.5 20 40 60 100 TYPICAL MINIMUM

TYPICAL PERFOR A CE CHARACTERISTICS UW Switching Frequency vs % of Feedback Voltage TEMPERATURE (°C) –50 FREQUENCY (MHz) 1.3 1.2 1.1 1.0 0.9 –25 0 25 50

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% OF FEEDBACK VOLTAGE SWITCHING FREQUENCY (MHz)0.2 0.4 0.6 0.8 1.0 1.2 20 40 60 80

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TA = 25°C TEMPERATURE (°C) –50 RUN/SS CURRENT (µA) 0.5 1.0 1.5 2.0 3.0 –25 02 5 5 0

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2.5 IRUN/SS vs Temperature RUN/SS Thresholds vs Temperature Minimum Input Voltage VOUT2 = 5V Minimum Input Voltage VOUT2 = 3.3V Frequency vs Temperature 1.4 1.2 1.0 0.8 0.6 0.4 0.2 RUN/SS THRESHOLDS (V) TEMPERATURE (°C) –50 25 75

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–25 0 50 100 125 TO SWITCH TO RUN LOAD CURRENT (mA) 5.0 MINIMUM INPUT VOLTAGE (V) 7.0 7.5 8.0 10 100 1000

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6.5 6.0 5.5 VIN TO START DBOOST = CMDSH-3 TA = 25°C VIN TO RUN BOOST DIODE TIED TO OUTPUT BOOST DIODE TIED TO INPUT LOAD CURRENT (mA) 3.0 MINIMUM INPUT VOLTAGE (V) 5.0 5.5 6.0 10 100 1000

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4.5 4.0 3.5 VIN TO START VIN TO RUN BOOST DIODE TIED TO OUTPUT BOOST DIODE TIED TO INPUT DBOOST = CMDSH-3 TA = 25°C

VIN (Pins 1, 2, 22, 25): The VIN pins supply current to the LT1941’s internal circuitry and to the internal power switches. These pins must be tied to the same source and locally bypassed. SW1, SW2, SW3 (Pins 3, 4, 23, 27): The SW pins are the outputs of the internal power switches. Connect these pins to the inductors and switching diodes. BOOST1, BOOST2 (Pins 5, 24): The BOOST pins are used to provide drive voltages, higher than the input voltage, to the internal bipolar NPN power switches. Tie through a diode from V OUT or from VIN. PGOOD1, PGOOD2, PGOOD3 (Pins 6, 9, 21): The PGOOD pins are the open-collector outputs of an internal com- parator. PGOOD remains low until the FB pin is within 10% of the final regulation voltage. As well as indicating output regulation, the PGOOD pins can sequence the switching regulators. Leave these pins unconnected if unused. The PGOOD outputs are valid when V IN is greater than 3.5V and any of the RUN/SS pins are high. They are not valid when all RUN/SS pins are low. VC1, VC2, VC3 (Pins 7, 10, 18): The VC pins are the outputs of the internal error amps. The voltages on these pins control the peak switch currents. These pins are normally used to compensate the control loops. Each switching regulator can be shut down by pulling its respective V C pin to ground with an NMOS or NPN transistor. FB1, FB2, FB3 (Pins 8, 11, 20): The LT1941 regulates each feedback pin to either 0.628V (FB1, FB2) or 1.25V (FB3). Connect the feedback resistor divider taps to these pins. RUN/SS1, RUN/SS2, RUN/SS3 (Pins 12, 13, 14): The RUN/SS pins are used to shut down the individual switch- ing regulators and the internal bias circuits. They also provide a soft-start function. To shut down either regula- tor, pull the RUN/SS pin to ground with an open drain or collector. Tie a capacitor from this pin to ground to limit switch current during start-up. If neither feature is used, leave these pins unconnected. BIAS1 (Pin 15): The BIAS1 pin supplies the current to the LT1941’s internal regulator. Tie this pin to the lowest available voltage source above 2.35V (Either V IN, VOUT or any other available supply). 12GOOD (Pin 16): The 12GOOD pin is the open-collector output of an internal comparator. 12GOOD remains low until VIN is within 10% of 12V. The pin pulls low when the part is in shutdown. Leave this pin unconnected if unused. 5GOOD (Pin 17): The 5GOOD pin is the open-collector output of an internal comparator. 5GOOD remains low until V IN is within 10% of 5V. The pin pulls low when the part is in shutdown. Leave this pin unconnected if unused. NFB (Pin 19): The LT1941 contains an op amp configured with an output at FB3, noninverting terminal at GND and an inverting terminal at NFB. Connect the feedback resistor network virtual ground at this node if regulating negative voltages. Otherwise, tie this node to FB3. PGND (Pin 26): Tie directly to local ground plane. BIAS2 (Pin 28): The BIAS2 pin supplies the current to the driver of SW3. Tie this pin to the lowest available voltage source above 2.5V (Either V IN, VOUT or any other available supply). Exposed Pad (Pin 29): Ground. The underside Exposed Pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be soldered to the circuit board for proper operation.

The LT1941 is a constant frequency, current mode, triple output regulator with internal power switches. The three regulators share common circuitry including input source, voltage reference and oscillator, but are otherwise inde- pendent. Operation can be best understood by referring to the Block Diagram. If the RUN/SS pins are tied to ground, the LT1941 is shut down and draws 50mA from the input source tied to V IN. Internal 2mA current sources charge external soft-start capacitors, generating voltage ramps at these pins. If any of the RUN/SS pins exceed 0.6V, the internal bias circuits turn on, including the internal regulator, reference and 1.1MHz master oscillator. Each switching regulator will only begin to operate when its corresponding RUN/SS pin reaches »1V. The master oscillator generates three clock signals, with the two signals for the step-down regulators out of phase by 180°. The three switchers are current mode regulators. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and pro- vides cycle-by-cycle current limit. The Block Diagram shows only one of the two step-down switching regulators. A pulse from the slave oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at V C, current comparator C1 resets the flip-flop, turning off the switch. The current in the inductor flows through the external Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the V C pin controls the current through the inductor to the output. The internal error amplifier regulates the output voltage by continually adjusting the V C pin voltage. The threshold for switching on the VC pin is »1V and an active clamp of 1.8V limits the output current. The RUN/SS pin voltage also clamps the V C pin voltage. As the internal current source charges the external soft-start capacitor, the current limit increases slowly. An internal op amp allows the part to regulate negative voltages using only two external resistors. Each switcher contains an extra, independent oscillator to perform frequency foldback during overload conditions. This slave oscillator is normally synchronized to the master oscillator. A comparator senses when V FB is less than 50% of its regulated value and switches the regulator from the master oscillator to a slower slave oscillator. The VFB pin is less than 50% of its regulated value during start- up, short circuit and overload conditions. Frequency foldback helps limit switch current power under these conditions. The switch drivers for SW1 and SW2 operate either from VIN or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. The BIAS1 pin allows the internal circuitry to draw its current from a lower voltage supply than the input, also reducing power dissipation and increasing efficiency. If the voltage on the BIAS1 pin falls below 2.35V, then its quiescent current will flow from V IN. The BIAS2 pin allows the driver for SW3 to draw its current from a lower voltage supply than the input. This reduces power dissipation within the part and increases efficiency. If the voltage on the BIAS2 pin falls below »2V, then SW3 will lock out and will not be able to turn on until BIAS2 rises above »2.1V. A power good comparator trips when the FB pin is at 90% of its regulated value. The PGOOD output is an open- collector transistor that is off when the output is in regulation, allowing an external resistor to pull the PGOOD pin high. Power good is valid when the LT1941 is enabled and V IN > 3.5V. Input power good comparators monitor the input supply. The 5GOOD and 12GOOD pins are open-collector outputs of internal comparators. The 5GOOD pin remains low until the input is within 10% of 5V. The 12GOOD pin remains low until the input is within 10% of 12V. The 5GOOD and 12GOOD pins are valid as long as V IN is greater than 1.1V. Both the 5GOOD and 12GOOD pins will sink current when the part is in shutdown, independent of the voltage at V IN. BLOCK DIAGRA W

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Figure 2. Block Diagram of the LT1941 with Associated External Components

R2 should be 10k or less to avoid bias current errors. lists several vendors and types that are suitable. Table 1. Inductors

APPLICATIO S I FOR ATIOWU UU The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a slightly higher maximum load current and will reduce the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note AN44. Finally, for duty cycles greater than 50% (V OUT/VIN > 0.5), a minimum inductance is required to avoid subharmonic oscillations. See AN19. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to- peak inductor ripple current. The LT1941 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT1941 will deliver depends on the switch current limit, the inductor value and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor: DI L = (1 – DC)(VOUT + VF)/(L • f) where f is the switching frequency of the LT1941 and L is the value of the inductor. The peak inductor and switch current is: ISWPK = ILPK = IOUT + DIL/2 To maintain output regulation, this peak current must be less than the LT1941’s switch current limit ILIM. For SW1, ILIM is at least 3A at low duty cycles and decreases linearly to 2.4A at DC = 0.8. For SW2, ILIM is at least 2A for at low duty cycles and decreases linearly to 1.6A at DC = 0.8. The maximum output current is a function of the chosen inductor value: I OUT(MAX) = ILIM – DIL/2 = 3 • (1 – 0.25 • DC) – DIL/2 for SW1 = 2 • (1 – 0.25 • DC) – DIL/2 for SW2 Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors and choose one to meet cost or space goals. Then use these equations to check that the LT1941 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when I OUT is less than DIL/2. Output Capacitor Selection For 5V and 3.3V outputs, a 10mF, 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. For lower voltages, 10mF is adequate for ripple requirements but increasing C OUT will improve transient performance. Other types and values will also work; the following discusses tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to gener- ate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilize the LT1941’s control loop. Because the LT1941 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option.

with applied voltage and at temperature extremes. Table 2. Low ESR Surface Mount Capacitors will then increase to the typical peak switch current. Table 3. Schottky Diodes

BOOST and SW pins is less than 25V. will present a load to the switcher that will allow it to start. Figure 3. Generating the Boost Voltage

1941 F03

down mode and the SW pin current drops to ~50mA. the system from a shorted or reversed input. Figure 4. Diode D4 Prevents a Shorted Input from

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R4 should be 10k or less to avoid bias current errors. Use 10k or larger, up to 20k for R3.

total switch current so that 0.75A per inductor is sufficient. the VBIAS2 • IBIAS2 power loss inside the part. describe the input capacitor considerations in more detail. Table 4. Inductors

sources will charge these pins to ~2V. three outputs. Figure 6 shows several circuits to do this. Table 5. Converter Equivalent Model Parameters Figure 5. Model for Loop Response

1941 F05

switch current and therefore input current during start-up. where COUT is the value of the output capacitor.

final values for these capacitors. with both channels sharing a single soft-start capacitor. nel␣ 2 until output 1 is in regulation. gets high enough to allow channel 2 to start switching. external control signals and output 1 is in regulation.

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Figure 6. Several Methods of Sequencing Two Ouputs. Channel 1 Starts First

Figure 7. Subtracting the Current when the Switch is ON (a) From the Current when the Switch is OFF (b) Reveals the Path

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Figure 8. Power Path Components and Topside Layout

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5 CHANNEL 3

1941 F09b

2.5 CHANNEL 3

Figure 9. Power Output Capability of an Individual Channel Depends on the Output Power of the Other Channels Channel 1 output power can be up to 5W. output supply using a buck regulator.

–21.6V, –65V, 3.3V and 1.8V with Soft-Start VIN LT1941 GNDPGND 0.22µF 22µF C13 1000pF C15 1.5nF C14 1.5nF C12 3300pF 10.2k R15 R13 178k R11 2.49k R10 10k V IN 5GOOD 12GOOD VOUT1 1.8V 2.4A VOUT3 –21.6V 72mA VOUT4 –65V 30mA L2 3.3µHL1 3µH 2.7µH R12 10.7k 13.7k 7.32k 3.3k R14 15k

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V OUT2 3.3V 1.4A 0.22µF 33µF 1µF 35VC6 10µF 4.7µF 25V 1µF 35V C10 1µF 35V C16 4700pF PGOOD1 PGOOD2 PGOOD3 BOOST2 FB2 V FB1 V RUNSS2 BIAS1 BIAS2 V RUNSS3 RUNSS1 SW3 NFB FB3 5GOOD 12GOOD BOOST1 C17 1.5nF NOTE: TOTAL OUTPUT POWER OF VOUT3 AND VOUT4 NOT TO EXCEED 1.9W C1 TO C11: X5R OR X7R D1, D2: CMDSH-3 D3: B220A D4: MBRM120L D5 TO D7: BAV99 OR EQUIVALENT SW2SW1 VOUT2VOUT1 4.7µF 25V C11 4.7µF 25V

Quadruple Output Power Supply –12V, 3.3V and 2.5V with Soft-Start VIN VOUT3VOUT2VOUT1 LT1941 GNDPGND 0.22µF 22µF C12 1000pF C13 1.5nF C11 1.5nF C10 1000pF 13.7k R13 118k R11 2.49k R10 10k V IN 5GOOD 12GOOD VOUT1 2.5V 2.3A VOUT3 12V 100mA VOUT4 –12V 100mA L2 3.3µHL1 3µH L3 10µH R12 10.7k 10.2k 3.4k 10k R14 2.2k C1 TO C9: X5R OR X7R D1, D2: CMDSH-3 D3: B220A D4: MBRM120L D5 TO D8: MBR0540

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V OUT2 3.3V 1.4A 0.22µF 33µF 4.7µF 4.7µFD5 10µF 10µF C10 10µF C14 6800pF PGOOD1 PGOOD2 PGOOD3 BOOST2 FB2 V FB1 V RUNSS2 BIAS1 BIAS2 V RUNSS3 RUNSS1 SW3 NFB FB3 5GOOD 12GOOD BOOST1 C15 1.5nF SW2SW1

28-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation EB Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. FE28 (EB) TSSOP 0204 0.09 – 0.20 (.0035 – .0079) 0° – 8° 0.25 REF 0.50 – 0.75 (.020 – .030) 4.30 – 4.50* (.169 – .177) 13 4 5 6 7 89 1 0 11 12 13 14 192022 21 15 1618 17 9.60 – 9.80* (.378 – .386) 4.75 (.187) 2.74 (.108) 28 2726 25 24 23 1.20 (.047) MAX 0.05 – 0.15 (.002 – .006) 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 2RECOMMENDED SOLDER PAD LAYOUT EXPOSED PAD HEAT SINK ON BOTTOM OF PACKAGE0.45 –0.05

0.65 BSC

4.50 –0.10 6.60 –0.10 1.05 –0.10 4.75 (.187) 2.74 (.108) MILLIMETERS (INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS 2. DIMENSIONS ARE IN 3. DRAWING NOT TO SCALE SEE NOTE 4 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT 6.40 (.252) BSC

LT/TP 0504 1K • PRINTED IN USA ª LINEAR TECHNOLOGY CORPORATION 2004 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1613 550mA (I SW), 1.4MHz, High Efficiency Step-Up DC/DC Converter V IN: 0.9V to 10V, VOUT(MAX) = 34V, IQ = 3mA, ISD < 1mA, ThinSOT TM Package LT1615/LT1615-1 300mA/80mA (I SW), High Efficiency Step-Up DC/DC Converter V IN: 1V to 15V, VOUT(MAX) = 34V, IQ = 20mA, ISD < 1mA, ThinSOT Package LT1617/LT1617-1 300mA/100mA (I SW), 1.2MHz/2.2MHz, High Efficiency Inverting V IN: 1.2V to 15V, VOUT(MAX) = –34V, IQ = 20mA, ISD < 1mA, DC/DC Converter ThinSOT Package LT1618 1.5A (I SW), 1.25MHz, High Efficiency Step-Up DC/DC Converter V IN: 1.6V to 18V, VOUT(MAX) = 35V, IQ = 1.8mA, ISD < 1mA, DC/DC Converter ThinSOT Package LT1931/LT1931A 1A (I SW), 1.2MHz/2.2MHz, High Efficiency Inverting V IN: 2.6V to 16V, VOUT(MAX) = –34V, IQ = 5.8mA, ISD < 1mA, DC/DC Converter ThinSOT Package LT1943 Quad Output, 2.6A Buck, 2.6A Boost, 0.3A Boost, V IN: 4.5V to 22V, VOUT(MAX) = 40V, IQ = 10mA, ISD < 35mA, 0.4A Inverter 1.2MHz TFT DC/DC Converter TSSOP28E Package LT1944-1 Dual Output 150mA (I SW), Constant Off-Time, High Efficiency V IN: 1.2V to 15V, VOUT(MAX) = 34V, IQ = 20mA, ISD < 1mA, Step-Up DC/DC Converter MS10 Package LT1944 Dual Output 350mA (I SW), Constant Off-Time, High Efficiency V IN: 1.2V to 15V, VOUT(MAX) = 34V, IQ = 20mA, ISD < 1mA, Step-Up DC/DC Converter MS10 Package LT1945 Dual Output Pos/Neg 350mA (I SW), Constant Off-Time, V IN: 1.2V to 15V, VOUT(MAX) = –34V, IQ = 20mA, ISD < 1mA, High Efficiency Step-Up DC/DC Converter MS10 Package DC/DC Converter MS8 Package LT1961 1.5A (I SW), 1.25MHz, High Efficiency Step-Up DC/DC Converter V IN: 3V to 25V, VOUT(MAX) = 35V, IQ = 0.9mA, ISD < 6mA, LT3436 3A (I SW), 1MHz, 34V Step-Up DC/DC Converter V IN: 3V to 25V, VOUT(MAX) = 34V, IQ = 0.9mA, ISD < 6mA, LT3461/LT3461A 300mA (I SW), High Efficiency Step-Up DC/DC Converter with V IN: 2.5V to 16V, VOUT(MAX) = 38V, IQ = 2.8mA, ISD < 1mA, Integrated Schottky and Soft-Start ThinSOT Package LT3463 Dual Output Pos/Neg 250mA (I SW), Constant Off-Time, High V IN: 2.4V to 15V, VOUT(MAX) = –40V, IQ = 40mA, ISD < 1mA, Efficiency Step-Up DC/DC Converter with Integrated Schottkys 3mm · 3mm DFN10 Package LT3464 85mA (I SW), High Efficiency Step-Up DC/DC Converter with V IN: 2.3V to 10V, VOUT(MAX) = 34V, IQ = 25mA, ISD < 1mA, Integrated Schottky and PNP Disconnect ThinSOT Package ThinSOT is a trademark of Linear Technology Corporaton. Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 l FAX: (408) 434-0507 l www.linear.com