LT1507I

LT1507 LINER | Alldatasheet

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Buck Mode Switching Regulator 5V to 3.3V Efficiency LOAD CURRENT (A) EFFICIENCY (%) 100 1.00 LT1507 • TA02 0.25 0.50 0.75 1.25 VIN = 5V VOUT = 3.3V TYPICAL APPLICATIONU 5V to 3.3V Volt Down Converter VIN SHDN VSW SENSE BOOST GND V C LT1507-3.3 D2† 1N914 0.1µF L1* 5µH 1N5818CC 3.3nF +C1 100µF 10V TANTALUM OUTPUT 3.3V 1.25A DEFAULT (OPEN) = ON +C3* 47µF 16V TANTALUM *** AVX TPSD477M016R0150 OR SPRAGUE 593D EQUIVALENT. RIPPLE CURRENT RATING ‡ 0.6A AVX TPSD108M010R0100 OR SPRAGUE 593D EQUIVALENT COILTRONICS CTX5-1. SUBSTITUTION UNITS SHOULD BE RATED AT ‡ 1.25A, USING LOW LOSS CORE MATERIAL SEE BOOST PIN CONSIDERATIONS IN APPLICATIONS INFORMATION SECTION FOR ALTERNATIVE D2 CONNECTION with all the necessary oscillator, control and logic cir- cuitry. High switching frequency allows a considerable reduction in the size of external components. The topology is current mode for fast transient response and good loop stability. Both fixed output voltage (3.3V) and adjustable parts are available. A special high speed bipolar process and new design techniques allow this regulator to achieve high efficiency at a high switching frequency. Efficiency is maintained over a wide output current range by keeping quiescent supply current to 4mA and by utilizing a supply boost capacitor to allow the NPN power switch to saturate. A shutdown signal will reduce supply current to 20µA. The LT1507 can be externally synchronized from 570kHz to 1MHz with logic level inputs. The LT1507 fits into standard 8-pin SO and PDIP pack- ages. Temperature rise is kept to a minimum by the high efficiency design. Full cycle-by-cycle short-circuit protec- tion and thermal shutdown are provided. Standard surface mount external parts are used including the inductor and capacitors. , LTC and LT are registered trademarks of Linear Technology Corporation. n Portable Computers n Battery-Powered Systems n Battery Charger n Distributed Power APPLICATIONSU

FEATURES

n Constant 500kHz Switching Frequency n Uses All Surface Mount Components n Operates with Inputs as Low as 4V n Saturated Switch Design (0.3Ω ) n Cycle-by-Cycle Current Limiting n Easily Synchronizable n Inductor Size as Low as 2µH n Shutdown Current: 20µA The LT®1507 is a 500kHz monolithic buck mode switching regulator, functionally identical to the LT1375 but opti- mized for lower input voltage applications. It will operate over a 4V to 15V input range, compared with 5.5V to 25V for the LT1375. A 1.5A switch is included on the die along DESCRIPTIONU

Operating Ambient Temperature Range ABSOLUTE MAXIMUM RATINGSW WW U PACKAGE/ORDER INFORMATIONW UU TJMAX = 125°C, θJA = 80°C/W TO 120°C/ W (N) TJMAX = 125°C, θJA = 120°C/W TO 170°C/ W (S) DEPENDING ON PC BOARD LAYOUT TOP VIEW VC FB/SENSE GND SYNC BOOST V IN VSW SHDN N8 PACKAGE 8-LEAD PDIP S8 PACKAGE 8-LEAD PLASTIC SO Consult factory for Military grade parts. ORDER PART NUMBER LT1507CN8 LT1507CN8-3.3 LT1507CS8 LT1507CS8-3.3 LT1507IN8 LT1507IN8-3.3 LT1507IS8 LT1507IS8-3.3 S8 PART MARKING

ELECTRICAL CHARACTERISTICS

TJ = 25°C, VIN = 5V, VC = 1.5V, boost open, switch open unless otherwise specified. PARAMETER CONDITIONS MIN TYP MAX UNITS Reference Voltage (Adjustable) 2.39 2.42 2.45 V All Conditions l 2.36 2.48 V Sense Voltage (3.3V) 3.25 3.3 3.35 V All Conditions l 3.23 3.37 V Sense Pin Resistance l 4.0 6.6 9.5 k Ω Reference Voltage Line Regulation 4.3V ≤ VIN ≤ 15V l 0.01 0.03 %/V FB Input Bias Current l 0.5 2 µA Error Amplifier Voltage Gain (Note 8) (Note 1) 150 400 Error Amplifier Transconductance (Note 8) ΔI(VC) = ±10µA 1500 2000 2700 µmho l 1100 3000 µmho VC Pin to Switch Current Transconductance 2 A/V Error Amplifier Source Current V FB = 2.1V or VSENSE = 2.9V l 150 225 320 µA Error Amplifier Sink Current V FB = 2.7V or VSENSE = 3.7V 2 mA VC Pin Switching Threshold Duty Cycle = 0 0.9 V VC Pin High Clamp V FB = 2.1V or VSENSE = 2.9V 2.1 V Switch Current Limit V C Open, VFB = 2.1V or VSENSE = 2.9V DC ≤ 50% l 1.50 2 3 A VIN ≥ 5V, VBOOST = VIN + 5V DC = 80% l 1.35 3 A Switch On Resistance (Note 6) I SW = 1.5A, VBOOST = VIN + 5V 0.3 0.4 Ω l 0.5 Ω Maximum Switch Duty Cycle V FB = 2.1V or VSENSE = 2.9V 90 93 % l 86 93 % 1507I 1507I3 1507 15073

PARAMETER CONDITIONS MIN TYP MAX UNITS Switch Frequency V C Set to Give 50% Duty Cycle 460 500 540 kHz –25 °C ≤ TJ ≤ 125°C 440 560 kHz TJ ≤ –25°C 440 570 kHz Switch Frequency Line Regulation 4.3V ≤ VIN ≤ 15V l 0.05 0.15 %/V Frequency Shifting Threshold on FB Pin Δf = 10kHz l 0.8 1.0 1.3 V Minimum Input Voltage (Note 2) l 4 4.3 V Minimum Boost Voltage (Note 3) I SW ≤ 1.5A l 3 3.5 V Boost Current (Note 4) V BOOST = VIN + 5V I SW = 500mA, –25°C ≤ TJ ≤ 125°C1 2 2 2 m A TJ ≤ –25°C2 5 m A ISW = 1.5A, –25 °C ≤ TJ ≤ 125°C2 5 3 5 m A TJ ≤ –25°C4 0 m A Input Supply Current (Note 5) l 3.8 5.4 mA Shutdown Supply Current V SHDN = 0V, VIN ≤ 12V 15 50 µA VSW = 0V, VC Open l 75 µA Lockout Threshold V C Open l 2.3 2.38 2.46 V Shutdown Threshold V C Open Device Shutting Down l 0.15 0.37 0.70 V Device Starting Up l 0.25 0.45 0.70 V Minimum Synchronizing Amplitude l 1.5 2.2 V Synchronizing Frequency Range (Note 7) 580 1000 kHz The l denotes specifications which apply over the operating temperature range. Note 1: Gain is measured with a VC swing equal to 200mV above the low clamp level to 200mV below the upper clamp level. Note 2: Minimum input voltage is not measured directly, but is guaranteed by other tests. It is defined as the voltage where internal bias lines are still regulated, so that the reference voltage and oscillator frequency remain constant. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. Note 3: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 4: Boost current is the current flowing into the BOOST pin with the pin held 5V above input voltage. It flows only during switch ON time. Note 5: Input supply current is the bias current drawn by the V IN pin when the SHDN pin is held at 1V (switching disabled). Note 6: Switch ON resistance is calculated by dividing VIN to VSW voltage by the forced current (1.5A). See Typical Performance Characteristics for the graph of switch voltage at other currents. Note 7: For synchronizing frequency above 700kHz, with duty cycles above 50%, external slope compensation may be needed. See Applications Information. Note 8: Transconductance and voltage gain refer to the internal amplifier exclusive of the voltage divider. To calculate gain and transconductance refer to SENSE pin on fixed voltage parts. Divide values shown by the ratio V OUT/2.42. TJ = 25°C, VIN = 5V, VC = 1.5V, boost open, switch open unless otherwise specified. VC Pin Shutdown Threshold JUNCTION TEMPERATURE (°C) –50 2.44 2.43 2.42 2.41 2.40 100 LT1507 • TPC03 –25 0 25 50 75 125 FEEDBACK VOLTAGE (V) CURRENT (µA) 2.0 1.5 1.0 0.5 VOLTAGE CURRENT LOAD CURRENT (A) EFFICIENCY (%) 100 1.00 LT1507 • TA02 0.25 0.50 0.75 1.25 VIN = 5V VOUT = 3.3V Switch Peak Current Limit Feedback Pin Voltage and Current JUNCTION TEMPERATURE (°C) –50 1.4 1.2 1.0 0.8 0.6 0.4 100 LT1507 • TPC01 –25 0 25 50 75 125 THRESHOLD VOLTAGE (V) TYPICAL PERFORMANCE CHARACTERISTICS UW

JUNCTION TEMPERATURE (°C) –50 2.40 2.36 2.32 0.8 0.4 25 75 LT1507 • TPC05 –25 0 50 100 125 SHUTDOWN PIN VOLTAGE (V) STANDBY STARTUP SHUTDOWN Standby and Shutdown Thresholds TYPICAL PERFORMANCE CHARACTERISTICS UW INPUT VOLTAGE (V) INPUT SUPPLY CURRENT (µA) 369 1 2 LT1507 • TPC06 VSHDN = 0V Shutdown Supply Current Shutdown Supply Current Error Amplifier Transconductance TEMPERATURE (°C) –50 500 400 300 200 25 75 L11507 • TPC04 –25 0 50 100 125 CURRENT (µA) CURRENT REQUIRED TO FORCE SHUTDOWN (FLOWS OUT OF PIN). AFTER SHUTDOWN, CURRENT DROPS TO A FEW µA AT 2.38V STANDBY THRESHOLD (CURRENT FLOWS OUT OF PIN) JUNCTION TEMPERATURE (°C) –50 TRANSCONDUCTANCE (µmho) 2500 2000 1500 1000 500 0 50 75 LT1507 • TPC08 –25 25 100 125 SHUTDOWN VOLTAGE (V) INPUT SUPPLY CURRENT (µA) 150 125 100 0.1 0.2 0.3 0.4 LT1507 • TPC07 0.5 VIN = 10V FREQUENCY (Hz) GAIN (µmho) PHASE (DEG) 3000 2500 2000 1500 1000 500 200 150 100 –50 100 10k 100k 10M LT1507 • TPC09 1k 1M GAIN PHASE ROUT 200k COUT 12pF VC VFB × 2e–3 ERROR AMPLIFIER EQUIVALENT CIRCUIT RLOAD = 50Ω Error Amplifier Transconductance Minimum Input Voltage with 3.3V OutputSwitching Frequency FEEDBACK PIN VOLTAGE (V) SWITCHING FREQUENCY (kHz) OR CURRENT (µA)500 400 300 200 100 2.0 LT1507 • TPC10 0.5 1.0 1.5 2.5 SWITCHING FREQUENCY FEEDBACK PIN CURRENT Frequency Foldback JUNCTION TEMPERATURE (°C) –50 600 550 500 450 400 100 LT1507 • TPC11 –25 0 25 50 75 125 FREQUENCY (kHz) LOAD CURRENT (mA) 5.0 INPUT VOLTAGE (V) 5.5 6.0 6.5 10 100 1000 LT1507 • TPC12 4.5 4.0 3.5 3.0 MINIMUM VOLTAGE TO START WITH STANDARD CIRCUITMINIMUM VOLTAGE TO RUN WITH STANDARD CIRCUIT MINIMUM INPUT VOLTAGE CAN BE REDUCED BY ADDING A SMALL EXTERNAL PNP. SEE APPLICATIONS INFORMATION

TYPICAL PERFORMANCE CHARACTERISTICS UW Inductor Core Loss for 3.3V Output INDUCTANCE (µH)

0.001 CORE LOSS (W)

0.1 0.01 1.0 24 6 8 1 0 LT1507 • TPC17 TYPE 52 POWDERED IRON Kool Mµ® PERMALLOY µ = 125 Metglas® CORE LOSS IS INDEPENDENT OF LOAD CURRENT UNTIL LOAD CURRENT FALLS LOW ENOUGH FOR CIRCUIT TO GO INTO DISCONTINUOUS MODE VOUT = 3.3V VIN = 5V IOUT = 1A Kool Mµ is a registered trademark of Magnetics, Incorporated. Metglas is a registered trademark of AlliedSignal Incorporated. BOOST (Pin 1): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. Without this added voltage the typical switch voltage loss would be about 1.5V. The additional boost voltage allows the switch to saturate and voltage loss approximates that of a 0.3 Ω FET structure, but with a much smaller die area. Efficiency improves from PIN FUNCTIONSUU U 70% for conventional bipolar designs to greater than 85% for these new parts. VIN (Pin 2): Input Pin. The LT1507 is designed to operate with an input voltage between 4.5V and 15V. Under certain conditions, input voltage may be reduced down to 4V. Actual minimum operating voltage will always be higher than the output voltage. It may be limited by switch OUTPUT VOLTAGE (%) OUTPUT CURRENT (A) 2.5 2.0 1.5 1.0 0.5 LT1507 • TPC13 20 40 60 100 MOS LOAD RESISTOR LOAD CURRENT SOURCE LOAD FOLDBACK CHARACTERISTICS *POSSIBLE UNDESIRED STABLE POINT FOR CURRENT SOURCE LOAD *SEE "MORE THAN JUST VOLTAGE FEEDBACK" IN APPLICATIONS INFORMATION SECTION Current Limit Foldback INPUT VOLTAGE (V) CURRENT (A) 0.25 0.50 0.75 1.00 1.25 1.50 6 8 10 12 LT1507 • TPC14 L = 10µH L = 5µH L = 3µH L = 2µH VOUT = 3.3V Maximum Load Current at VOUT = 3.3V INPUT VOLTAGE (V) CURRENT (A) 0.25 0.50 0.75 1.00 1.25 1.50 369 1 2 LT1507 • TPC15 L = 10µH L = 5µH L = 20µH Maximum Load Current at VOUT = 5V Boost Pin Current SWITCH CURRENT (A) BOOST PIN CURRENT (mA) 0.25 0.50 0.75 1.00 LT1507 • TPC16 1.25 TJ = 25°C Switch Voltage Drop SWITCH CURRENT (A) SWITCH VOLTAGE (V) 0.8 0.6 0.4 0.2 0.25 0.50 0.75 1.00 LT1507 • TPC18 1.25 1.50 TJ = 25°C

saturation voltage and maximum duty cycle. A typical value for minimum input voltage is 1V above output voltage. Start-up conditions may require more voltage at light loads. See Minimum Input Voltage for details. V SW (Pin 3): The switch pin is driven up to the input voltage in the ON state and is an open circuit in the OFF state. At higher load currents, pin voltage during the off condition will be one diode drop below ground as set by the external catch diode. At lighter loads the pin will assume an intermediate state equal to output voltage during part of the switch OFF time. Maximum negative voltage on the switch pin is 1V with respect to the GND pin, so it must always be clamped with a catch diode to the GND pin. SHDN (Pin 4): The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. Actually this pin has two separate thresh- olds, one at 2.38V to disable switching and a second at 0.4V to force complete micropower shutdown. The 2.38V threshold functions as an accurate undervoltage lockout (UVLO). This is sometimes used to prevent the regulator from delivering power until the input voltage has reached a predetermined level. BLOCK DIAGRAMW The LT1507 is a constant frequency, current mode buck converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on. When switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error amplifier commands current to be delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor and output capacitor, then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency, but will not have the additional 90° shift until well beyond the LC resonant frequency. This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. High switch efficiency is attained by using the BOOST pin to provide a voltage to the switch driver which is higher than the input voltage, allowing the switch to be saturated. This boosted voltage is generated with an external capaci- tor and diode. Two comparators are connected to the shutdown pin. One has a 2.38V threshold for undervoltage lockout and the second has a 0.4V threshold for complete shutdown. SYNC (Pin 5): The SYNC pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 10% and 90% duty cycle. The synchronizing range is equal to initial operating frequency up to 1MHz. See Sychronizing section for details. FB/SENSE (Pin 7): The feedback pin is used to set output voltage using an external voltage divider that generates 2.42V at the pin with the desired output voltage. The fixed voltage (– 3 .3V) parts have the divider included on the chip and the feedback pin is used as a sense pin connected directly to the 5V output. Two additional functions are performed by the feedback pin. When the pin voltage drops below 1.7V, switch current limit is reduced. Below 1V, switching frequency is also reduced. See More Than Just Voltage Feedback. V C (Pin 8): The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation but can do double duty as a current clamp or control loop override. This pin sets at about 1V for very light loads and 2V at maximum load. It can be driven to ground to shut off the regulator, but if driven high, current must be limited to 4mA.

complete version found in the LT1375/LT1376 data sheet. Figure 1. Block Diagram

SENSE, connected directly to the output. frequency during start-up as the output voltage rises. diode to defeat foldback current limit. AVX TPSD337M020R0200 OR SPRAGUE 593 EQUIVALENT. Figure 2. Typical Schematic for LT1507 Adjustable Application

APPLICATIONS INFORMATIONWU UU pin (RDIV = R1/R2 ≤ 4k). The net result is that reductions in frequency and current limit are affected by output voltage divider impedance. Although divider impedance is not critical, caution should be used if resistors are increased beyond the suggested values and short-circuit conditions will occur with high input voltage. High frequency pickup will also increase and the protection accorded by frequency and current foldback will decrease. CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR For most applications the value of the inductor will fall in the range of 2 µH to 10 µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the LT1507 switch, which has a 1.5A limit. Higher values also reduce output ripple voltage and reduce core loss. Graphs in the Typical Performance Characteristics section show maximum output load current versus induc- tor size and input voltage. A second graph shows core loss versus inductor size for various core materials. When choosing an inductor you might have to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, EMI, fault cur- rent in the inductor, saturation and, of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Choose a value in microhenries from the graphs of Maximum Load Current and Inductor Core Loss for 3.3V Output. If you want to double check that the chosen inductor value will allow sufficient load current, go to the next section, Maximum Output Load Current. Choosing a small inductor with lighter loads may result in discontinuous mode of operation, but the LT1507 is designed to work well in either mode. Keep in mind that lower core loss means higher cost, at least for closed- core geometries like toroids. Type 52 powdered iron, Kool Mµ and Molypermalloy are old standbys for tor- oids in ascending order of price. A newcomer, Metglas, gives very low core loss with high saturation current. Assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions. If maxi- mum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 1.5A overload condition. Dead shorts (V OUT ≤ 1V) will actually be more gentle on the inductor because the LT1507 has foldback current limiting (see graph in Typical Performance Character- istics). 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, espe- cially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall in between somewhere. The following formula assumes a con- tinuous mode of operation, but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. II VV V fLVPEAK OUT OUT IN OUT IN =+ (± ) () ( ) ( )2 VIN = Maximum input voltage f = Switching frequency = 500kHz 3. Decide if the design can tolerate an “open” core geom- etry like ferrite rods or barrels, which have high mag- netic field radiation or whether it needs a closed core like a toroid to prevent EMI problems. One would not want an open core next to a magnetic storage media for instance! This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. The following is an example of just how subtle the “B” field problems can be with open geometry cores. We had selected an open drum shaped ferrite core for the LTC1376 demonstration board because the induc- tor was extremely small and inexpensive. It met all the requirements for current and the ferrite core gave low core loss. When the boards came back from assembly, many of them had somewhat higher than expected output ripple voltage. We removed the inductors and output capacitors and found them to be no different than the good boards. After much head scratching and hours of delicate low level ripple measurements on the good and bad boards, I realized that the problem must

tor caused the induced field to reduce output ripple. have on regulator or system performances.

1. Look for an inductor (see Table 1) which meets the

cost, sometimes dramatically.

1. After making an initial choice, consider secondary things

Table 1. Representative Surface Mount Units needed for typical LT1507 applications is 0.05Ω to 0.5Ω . typical solid tantalum surface mount capacitors.

ripple current ratings and tolerance of turn-on surges. Table 2. Surface Mount Solid Tantalum Capacitor ESR are prone to failure if they undergo high surge currents. dead shorted, do not harm the capacitors.

APPLICATIONS INFORMATIONWU UU Example: with VOUT = 3.3V, VIN = 5V; VOUT/VIN = 3.3/5 = 0.67 Maximum load current would be equal to maximum switch current for an infinitely large inductor, but with finite inductor size, maximum load current is reduced by one half peak-to-peak inductor current. The following formula assumes continuous mode operation; the term on the right must be less than one half of I Continuous mode: II VV V LfVOUT MAX P OUT IN OUT IN () ± () ( ± ) () () ( )= 2 For the conditions above, with L = 5µH and f = 500kHz; I A .± . . ()  () 14 2 33 5 33 2 5 10 500 10 5 14 2 02 2 12 At VIN = 8V, VOUT/V IN = 0.41, so IP is equal to 1.5A and IOUT(MAX) is equal to; 15 33 8 33 2 5 10 500 10 8 15 03 9 11 1 .± ( . )( ± . ) .±. . ()  () == A Note that there is less load current available at the higher input voltage because inductor ripple current increases. This is not always the case. Certain combinations of inductor value and input voltage range may yield lower available load current at the lowest input voltage due to reduced peak switch current at high duty cycles. If load current is close to the maximum available, please check maximum available current at both input voltage extremes. To calculate actual peak switch current with a given set of conditions, use: II VV V LfVSWITCH PEAK OUT OUT IN OUT IN (± ) () ( ) ( )=+ 2 For lighter loads where discontinuous mode operation can be used, maximum load current is equal to: Discontinuous mode: I If L V VV VOUT MAX PI N OUT IN OUT () ( ) ( ) ( ) () ( ± )= Example: with L = 2µH, VOUT = 5V and VIN(MAX) = 15V; I mA OUT MAX() (.) () ( ± )= ()  () −1 5 500 10 2 10 15 2 5 15 5 338 23 6 The main reason for using such a tiny inductor is that it is physically very small, but keep in mind that peak-to-peak inductor current will be very high. This will increase output ripple voltage. If the output capacitor has to be made larger to reduce ripple voltage, the overall circuit could actually be larger. CATCH DIODE The suggested catch diode (D1) is a 1N5818 Schottky or its Motorola equivalent, MBR130. It is rated at 1A average forward current and 30V reverse voltage. Typical forward voltage is 0.42V at 1A. The diode conducts current only during switch OFF time. Peak reverse voltage is equal to regulator input voltage. Average forward current in normal operation can be calculated from: I IV V VD AVG OUT IN OUT IN (± )= This formula will not yield values higher than 1A with maximum load current of 1.25A unless the ratio of input to output voltage exceeds 5:1. The only reason to consider a larger diode is the worst-case condition of a high input voltage and overloaded (not shorted) output. Under short- circuit conditions, foldback current limit will reduce diode current to less than 1A, but if the output is overloaded and does not fall to less than 1/3 of nominal output voltage, foldback will not take effect. With the overloaded condi- tion, output current will increase to a typical value of 1.8A, determined by peak switch current limit of 2A. With V IN = 10V, VOUT = 2V (3.3V overloaded) and IOUT = 1.8A: IAD AVG() .( ±) .==1 8 10 2 10 14 4

ous operation under these conditions must be tolerated. of film or ceramic capacitor will work fine. Figure 3. Suggested Layout anode is connected to the output.

APPLICATIONS INFORMATIONWU UU LT1507 • F04 VIN HIGH FREQUENCY CIRCULATING PATH LOAD SWITCH NODE C3 C1 INPUT BYPASSING AND VOLTAGE RANGE Input Bypass Capacitor Stepdown converters draw current from the input supply in pulses. The average height of these pulses is equal to load current and the duty cycle is equal to VOUT/VIN. Rise and fall time of the current is very fast. A local bypass capacitor across the input supply is necessary to ensure proper operation of the regulator and minimize the ripple current fed back into the input supply. The capacitor also forces switching current to flow in a tight local loop, minimizing EMI. Do not cheat on the ripple current rating of the input bypass capacitor, but also don’t get hung up on the value in microfarads. The input capacitor is intended to absorb all the switching current ripple, which can have an RMS value as high as one half of load current. Ripple current ratings on the capacitor must be observed to ensure reliable operation. The actual value of the capacitor in microfarads is not particularly important because at 500kHz, any value above 5µF is essential resistive. Ripple current rating is the critical parameter. RMS ripple current can be calculated from: I RMS I VV V V RIPPLE OUT OUT IN OUT IN () (± )= 2 high speed circulating current path shown in Figure 4 and to make connections to the output capacitor in a manner that minimizes output ripple and noise. For more details, see Applications Information section in the LT1376 data sheet. Figure 4. High Speed Switching Path The term inside the radical has a maximum value of 0.5 when input voltage is twice output and stays near 0.5 for a relatively wide range of input voltages. It is common practice, therefore, to simply use the worst-case value and assume that RMS ripple current is one half of load current. At maximum output current of 1.5A for the LT1507, the input bypass capacitor should be rated at 0.75A ripple current. Note however, that there are many secondary considerations in choosing the final ripple current rating. These include ambient temperature, average versus peak load current, equipment operating schedule and required product lifetime. For more details see Application Notes 19 and 46. Input Capacitor Type Some caution must be used when selecting the type of capacitor used at the input of regulators. Aluminum electrolytics are lowest cost, but are physically large to achieve adequate ripple current rating, and size con- straints (especially height) may preclude their use. Ceramic capacitors are now available in larger values and their high ripple current and voltage rating make them ideal for input bypassing. Cost is slightly higher and footprint may also be somewhat larger. Solid tantalum capacitors are a good choice except that they have a history of occasional spectacular failures when they are subjected to very large current surges during power-up. The capacitors can short and then burn with a brilliant white light and lots of nasty smoke. This phenomenon occurs in only a small percentage of units, but it has led some OEM companies to forbid their use in high surge applications. The input bypass capacitor of regulators can see such high surges when a battery or high capacitance source is connected. Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (AVX TPS series for instance, see Table 2). Even these units may fail if the input current surge exceeds a value equal to the voltage rating of the capacitor divided by 1Ω (10A for a 10V capacitor). For this reason, AVX recom- mends using the highest voltage rating possible for the input capacitor. For equal case size, this means that lower values of capacitance must be used. As stated above, this

hot plugged to typical regulator systems. Sanyo but there are now additional sources. input voltage drops below the minimum specification. differential is near minimum. loads, see the next section. be added or the circuit in Figure 6 can be used. Figure 5. Minimum Input Voltage for VOUT = 3.3V

APPLICATIONS INFORMATIONWU UU For VIN = 4.7, VOUT = 3.3V, f = 1MHz, L = 5µH and DCS = 25%: Vm VP-P ≥ −− (. .) ( . ) 66 47 1 02 5 2 1 10 5 10 1 8 To avoid small values of RS, the compensation capacitor (CC) should be made as small as possible. 2000pF will work in most situations. If we increase V PP to 90mV for a little cushion, RS will be: Rk Cp F S = ()  () () () () (. ) (. ) .5 0 25 0 75 0 09 2 10 1 10 2 1 10 5200 612 THERMAL CALCULATIONS Power dissipation in the LT1507 chip comes from four sources: switch DC loss, switch AC loss, boost circuit current and input quiescent current. The formulas below show how to calculate each of these losses. These formu- las assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. Switch loss: P RI V V ns I V fSW SW OUT OUT IN Boost current loss: P V V I BOOST OUT IN OUT=+    0 008 75. Quiescent current loss: PV VQ IN OUT=+ (. ) (. )0 003 0 005 RSW = Switch resistance (≈ 0.4Ω ) 16ns = Equivalent switch current/voltage overlap time f = Switching frequency works by prematurely tripping the oscillator before it reaches its normal peak value. For instance, if the oscilla- tor is synchronized at twice its nominal frequency, oscil- lator amplitude will drop by half. A ramp which previously started at the 40% point now starts at the 80% point! This effectively blocks slope compensation and the regulator may respond with fluctuating pulse widths, a “phase oscillation” if you will. The regulator output stays in regulation but subharmonic frequencies are generated at the switch node. The solution to this problem is to generate an external ramp that replaces the missing internal ramp. As it turns out, this is not difficult if the sync signal can be arranged to have a fairly low duty cycle (< 35%). The ramp is created by AC coupling a resistor from the sync signal to the compensation capacitor as shown in Figure 7. This gener- ates a negative ramp on the V C pin during switch ON time that emulates the missing internally generated ramp. Amplitude of the ramp should be about 100mV to 200mV peak-to-peak. The formulas for calculating the values of R S and C S are shown below. Note that the C S value is unimportant as long as it exceeds the value given. The formula assures that the impedance of C S will be small compared to RS. R VD C D C VC f C fR S SYNC S S C S S = − () ( ) () ( ) () ( ) P-P π VSYNC = Peak-to-peak value of sync signal DCS = Duty cycle of incoming sync signal VP-P = Desired amplitude of ramp f = Sync frequency Theoretical minimum amplitude for the ramp, assuming no internal ramp, is: V VV D C fLg OUT IN S mP P-P ≥ −−() ( ) () ( ) ( ) gmP = Transconductance from V C pin to switch current (1.8A/V for the LT1507).

See Application Information in LT1376 data sheet. *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. for CF to limit switching frequency ripple at the VC pin. Figure 10. Overall Loop Phase and Gain Information furnished by Linear Technology Corporation is believed to be accurate and reliable. tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

PART NUMBER DESCRIPTION COMMENTS LT1371 3A 500kHz Step-Up Switching Regulator High Current DC/DC Conversion Uses Small Power Components LT1372 1.5A 500kHz Step-Up Switching Regulator Includes Positive and Negative Output Voltage Regulation LT1375 1.5A 500kHz Step-Down Switching Regulator Includes Synchronization Capability LT1376 1.5A 500kHz Step-Down Switching Regulator Output Biasing Yields 90% Efficiency LT1377 1.5A 1MHz Step-Up Switching Regulator Highest Frequency Monolithic Switching Regulator 1507f LT/TP 0697 4K • PRINTED IN USA  LINEAR TECHNOLOGY CORPORA TION 1996 Dimensions in inches (millimeters) unless otherwise noted.PACKAGE DESCRIPTIONU 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 1 2 3 4 0.150 – 0.157** (3.810 – 3.988) 8 7 6 5 0.189 – 0.197* (4.801 – 5.004) 0.228 – 0.244 (5.791 – 6.197) 0.016 – 0.050 0.406 – 1.270 0.010 – 0.020 0°– 8° TYP 0.008 – 0.010 (0.203 – 0.254) SO8 0695 0.053 – 0.069 (1.346 – 1.752) 0.014 – 0.019 (0.355 – 0.483) 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSCDIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE RELATED PARTS Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 l (408) 432-1900 FAX: (408) 434-0507 l TELEX: 499-3977 l www.linear-tech.com