LT1506IR

LT1506 LINER | Alldatasheet

Document overview

• Manufacturer or author: Provided By ALLDATASHEET.COM(FREE DATASHEET DOWNLOAD SITE)
• PDF pages: 24

Technical content

4.5A, 500kHz Step-Down Switching Regulator n Constant 500kHz Switching Frequency n Easily Synchronizable n Operates with Input as Low as 4V n Uses All Surface Mount Components n Inductor Size Reduced to 1.8µH n Saturating Switch Design: 0.07Ω n Shutdown Current: 20µA n Cycle-by-Cycle Current Limiting

FEATURES

The LT®1506 is a 500kHz monolithic buck mode switching regulator functionally identical to the LT1374 but optimized for lower input voltage applications. It will operate over a 4V to 15V input range compared with 5.5V to 25V for the LT1374. A 4.5A switch is included on the die along with all the necessary oscillator, control and logic circuitry. High switching frequency allows a considerable reduction in the size of external components. The topology is current mode for fast transient response and good loop stability. Both fixed output voltage and adjustable parts are available. A special high speed bipolar process and new design tech- niques achieve high efficiency at high switching frequency. Efficiency is maintained over a wide output current range by keeping quiescent supply current to 4mA and by utiliz- ing a supply boost capacitor to saturate the power switch. The LT1506 fits into standard 7-pin DD and fused lead SO-8 packages. Full cycle-by-cycle short-circuit protection and thermal shutdown are provided. Standard surface mount external parts are used, including the inductor and capacitors. There is the optional function of shutdown or synchronization. A shutdown signal reduces supply current to 20µA. Synchronization allows an external logic level sig- nal to increase the internal oscillator from 580kHz to 1MHz. DESCRIPTIONU n Portable Computers n Battery-Powered Systems n Battery Charger n Distributed Power APPLICATIONSU , LTC and LT are registered trademarks of Linear Technology Corporation. TYPICAL APPLICATIONU LOAD CURRENT (A) EFFICIENCY (%) 2.0 2.5 3.0 3.5

1506 TA02

0.5 1.0 1.5 4.0 VOUT = 3.3V VIN = 5V L = 10µH Efficiency vs Load Current 5V to 3.3V Down Converter BOOST LT1506-3.3 VIN OUTPUT 3.3V INPUT

1506 TA01

0.68µF CC 1.5nF MBRS330T3 100µF, 10V SOLID TANTALUM 10µF TO 50µF CERAMIC 1N914 5µH V SW SENSESHDN OPEN OR HIGH = ON GND VC

ABSOLUTE MAXIMUM RATINGSW WW U Operating Junction Temperature Range

ELECTRICAL CHARACTERISTICS

PARAMETER CONDITIONS MIN TYP MAX UNITS Feedback Voltage (Adjustable) 2.39 2.42 2.45 V All Conditions l 2.36 2.48 V Sense Voltage (Fixed 3.3V) 3.25 3.3 3.35 V All Conditions l 3.23 3.37 V SENSE Pin Resistance 4 6.6 9.5 k Ω Reference Voltage Line Regulation 4.3V ≤ VIN ≤ 15V 0.01 0.03 %/V Feedback Input Bias Current l 0.5 2 µA Error Amplifier Voltage Gain (Notes 2, 8) 200 400 Error Amplifier Transconductance ΔI (VC) = ±10µA (Note 8) 1500 2000 2700 µMho l 1000 3100 µMho VC Pin to Switch Current Transconductance 5.3 A/V Error Amplifier Source Current V FB = 2.1V or VSENSE = 2.9V l 140 225 320 µA Error Amplifier Sink Current V FB = 2.7V or VSENSE = 3.7V l 140 225 320 µA VC Pin Switching Threshold Duty Cycle = 0 0.9 V VC Pin High Clamp 2.1 V Switch Current Limit V C Open, VFB = 2.1V or VSENSE = 2.9V, DC ≤ 50% l 4.5 6 8.5 A Slope Compensation DC = 80% 0.8 A TJ = 25°C, VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted. PACKAGE/ORDER INFORMATIONW UU ORDER PART NUMBER ORDER PART NUMBER LT1506CR LT1506CR-3.3 LT1506CR-SYNC LT1506CR-3.3 SYNC LT1506IR LT1506IR-3.3 LT1506IR-SYNC LT1506IR-3.3 SYNC TJMAX = 125°C, θJA = 30°C/W WITH PACKAGE SOLDERED TO 0.5 SQUARE INCH COPPER AREA OVER BACKSIDE GROUND PLANE OR INTERNAL POWER PLANE. θ JA CAN VARY FROM 20°C/W TO >40 °C/W DEPENDING ON MOUNTING TECHNIQUES 1506I 506I33 1506 150633 S8 PART MARKING LT1506CS8 LT1506CS8-3.3 LT1506IS8 LT1506IS8-3.3 *Default is the adjustable output voltage device with FB pin and shutdown function. Option -3.3 replaces FB with SENSE pin for fixed 3.3V output applications. -SYNC replaces SHDN with SYNC pin for applications requiring synchronization. Consult factory for Military grade parts. FB OR SENSE* BOOST V IN GND V SW SYNC OR SHDN* V C R PACKAGE 7-LEAD PLASTIC DD PAK FRONT VIEW TAB IS GND TOP VIEW S8 PACKAGE 8-LEAD PLASTIC SO VIN BOOST GND** VSW SYNC SHDN V C FB OR SENSE* θJA = 80°C/ W **WITH FUSED (GND) GROUND PIN CONNECTED TO GROUND PLANE OR LARGE LANDS (Note 1)

PARAMETER CONDITIONS MIN TYP MAX UNITS Switch On Resistance (Note 7) I SW = 4.5A 0.07 0.1 Ω l 0.13 Ω Maximum Switch Duty Cycle V FB = 2.1V or VSENSE = 2.9V 90 93 % l 86 93 % Switch Frequency V C Set to Give 50% Duty Cycle 460 500 540 kHz l 440 560 kHz Switch Frequency Line Regulation 4.3V ≤ VIN ≤ 15V l 0 0.15 %/V Frequency Shifting Threshold on FB Pin Δf = 10kHz l 0.8 1.0 1.3 V Minimum Input Voltage (Note 3) l 4.0 4.3 V Minimum Boost Voltage (Note 4) I SW ≤ 4.5A l 2.3 3.0 V Boost Current (Note 5) I SW = 1A l 20 35 mA ISW = 4.5A l 90 140 mA Input Supply Current (Note 6) l 3.8 5.4 mA Shutdown Supply Current V SHDN = 0V, VSW = 0V, VC Open 15 50 µA l 75 µA Lockout Threshold V C Open l 2.3 2.38 2.46 V Shutdown Thresholds V C Open Device Shutting Down l 0.13 0.37 0.60 V Device Starting Up l 0.25 0.45 0.7 V Synchronization Threshold l 1.5 2.2 V Synchronizing Range 580 1000 kHz SYNC Pin Input Resistance 40 k Ω TJ = 25°C, VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted. The l denotes specifications which apply over the full operating temperature range. Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Gain is measured with a VC swing equal to 200mV above the switching threshold level to 200mV below the upper clamp level. Note 3: Minimum input voltage is not measured directly, but is guaranteed by other tests. It is defined as the voltage where internal bias lines are still regulated so that the reference voltage and oscillator frequency remain constant. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 5: Boost current is the current flowing into the boost pin with the pin held 5V above input voltage. It flows only during switch on time. Note 6: Input supply current is the bias current drawn by the input pin with switching disabled. Note 7: Switch on resistance is calculated by dividing V IN to VSW voltage by the forced current (4.5A). See Typical Performance Characteristics for the graph of switch voltage at other currents. Note 8: Transconductance and voltage gain refer to the internal amplifier exclusive of the voltage divider. To calculate gain and transconductance refer to SENSE pin on fixed voltage parts. Divide values shown by the ratio V OUT/2.42.

TYPICAL PERFORMANCE CHARACTERISTICS UW Feedback Pin Voltage TEMPERATURE (°C) –50 2.430 2.425 2.420 2.415 2.410 100

1506 G03

–25 0 25 50 75 125 FEEDBACK VOLTAGE (V) Switch Peak Current Limit DUTY CYCLE (%) SWITCH PEAK CURRENT (A) 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0

1506 G02

ThresholdsShutdown Pin Bias Current Shutdown Supply Current JUNCTION TEMPERATURE (°C) –50 2.40 2.36 2.32 0.8 0.4 25 75

1506 G05

–25 0 50 100 125 SHUTDOWN PIN VOLTAGE (V) LOCKOUT START-UP SHUTDOWN INPUT VOLTAGE (V) INPUT SUPPLY CURRENT (µA) 51 0 1 5

1506 G06

VSHDN = 0V Shutdown Supply Current SHUTDOWN VOLTAGE (V) INPUT SUPPLY CURRENT (µA) 0.1 0.2 0.3 0.4

1506 G07

VIN = 10V FREQUENCY (Hz) GAIN (µMho) PHASE (DEG) 3000 2500 2000 1500 1000 500 200 150 100 –50 100 10k 100k 10M

1506 G09

ERROR AMPLIFIER EQUIVALENT CIRCUIT ROUT 200k COUT 12pF VC RLOAD = 50Ω VFB 2 × 10–3)( Error Amplifier TransconductanceError Amplifier Transconductance JUNCTION TEMPERATURE (°C) –50 TRANSCONDUCTANCE (µMho) 2500 2000 1500 1000 500 0 50 75

1506 G08

–25 25 100 125 Minimum Input Voltage with 3.3V Output LOAD CURRENT (mA) 4.1 INPUT VOLTAGE (V) 4.3 4.5 4.7 10 100 1000

1506 G12

3.9 3.7 3.5 3.3 TEMPERATURE (°C) –50 –500 –400 –300 –200 25 75

1506 G04

–25 0 50 100 125 CURRENT (µA) AT 0.37V SHUTDOWN THRESHOLD. AFTER SHUTDOWN, CURRENT DROPS TO A FEW µA AT 2.38V LOCKOUT THRESHOLD

TYPICAL PERFORMANCE CHARACTERISTICS UW Frequency Foldback FEEDBACK PIN VOLTAGE (V) SWITCHING FREQUENCY (kHz) OR CURRENT (µA)500 400 300 200 100 2.0

1506 G10

0.5 1.0 1.5 2.5 SWITCHING FREQUENCY FEEDBACK PIN CURRENT Inductor Core Loss for 3.3V Output INDUCTANCE (µH) CORE LOSS (W) 1.0 0.1 0.01 0.001 468 1 0

1506 G01

Metglas® Kool Mµ® PERMALLOY µ = 125 Switching Frequency TEMPERATURE (°C) –5 0 550 540 530 520 510 500 490 480 470 460 450 100

1506 G11

–25 0 25 50 75 125 FREQUENCY (kHz) BOOST Pin Current SWITCH CURRENT (A) BOOST PIN CURRENT (mA) 12 3 45

1506 G14

DUTY CYCLE = 100% Maximum Load Current at VOUT = 3.3V INPUT VOLTAGE (V) 4.0 4.2 4.4

1506 G13

3.8 3.6 61 0 81 4 3.4 3.2 3.0 LOAD CURRENT (A) L= 10µH L= 5µH L= 3µH L= 1.8µH Maximum Load Current at VOUT = 5V INPUT VOLTAGE (V) 2.6 LOAD CURRENT (A) 2.8 3.2 3.4 3.6 9 13 15 4.4

1506 G17

3.0 71 1 3.8 4.0 4.2 L= 10µH L= 5µH L= 3µH L= 1.8µH Switch Voltage Drop SWITCH CURRENT (A) SWITCH VOLTAGE (mV) 250 200 400 125°C 25°C 350 300

1506 G16

–40 °C VC Pin Shutdown Threshold JUNCTION TEMPERATURE (°C) –50 1.4 1.2 1.0 0.8 0.6 0.4 100

1506 G15

–25 0 25 50 75 125 THRESHOLD VOLTAGE (V) SHUTDOWN OUTPUT VOLTAGE (%)

1506 G18

OUTPUT CURRENT (A) FOLDBACK CHARACTERISTICS CURRENT SOURCE LOAD RESISTOR LOAD MOS LOAD POSSIBLE UNDESIRED STABLE POINT FOR CURRENT SOURCE LOAD* Current Limit Foldback Kool Mµ is a registered trademark of Magnetics, Inc. Metglas is a registered trademark of AlliedSignal Inc. *See “More Than Just Voltage Feedback” in the Applications Information section.

FB/SENSE: The feedback pin is used to set output voltage using an external voltage divider that generates 2.42V at the pin with the desired output voltage. The fixed voltage (-3.3) parts have the divider included on the chip and the FB pin is used as a SENSE pin, connected directly to the 3.3V output. Three additional functions are performed by the FB pin. When the pin voltage drops below 1.7V, switch current limit is reduced. Below 1.5V the external sync function is disabled. Below 1V, switching frequency is also reduced. See Feedback Pin Function section in Applica- tions Information for details. BOOST: The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. Without this added voltage, the typical switch voltage loss would be about 1.5V. The additional boost voltage allows the switch to saturate and voltage loss approximates that of a 0.07Ω FET structure, but with much smaller die area. Efficiency improves from 75% for conventional bipolar designs to > 89% for these new parts. V IN: This is the collector of the on-chip power NPN switch. This pin powers the internal circuitry and internal regula- tor. At NPN switch on and off, high dI/dt edges occur on this pin. Keep the external bypass and catch diode close to this pin. All trace inductance on this path will create a voltage spike at switch off, adding to the V CE voltage across the internal NPN. GND: The GND pin connection needs consideration for two reasons. First, it acts as the reference for the regulated output, so load regulation will suffer if the “ground” end of the load is not at the same voltage as the GND pin of the IC. This condition will occur when load current or other currents flow through metal paths between the GND pin and the load ground point. Keep the ground path short between the GND pin and the load and use a ground plane when possible. The second consideration is EMI caused by GND pin current spikes. Internal capacitance between the V SW pin and the GND pin creates very narrow (<10ns) current spikes in the GND pin. If the GND pin is connected to system ground with a long metal trace, this trace may radiate excess EMI. Keep the path between the input bypass and the GND pin short. The GND pin of the SO-8 package is directly attached to the internal tab. This pin should be attached to a large copper area to improve thermal resistance. V SW: The switch pin is the emitter of the on-chip power NPN switch. This pin is driven up to the input pin voltage during switch on time. Inductor current drives the switch pin negative during switch off time. Negative voltage is clamped with the external catch diode. Maximum negative switch voltage allowed is – 0.8V. SYNC: The sync pin is used to synchronize the internal oscillator to an external signal. It is directly logic compat- ible and can be driven with any signal between 10% and 90% duty cycle. The synchronizing range is equal to initial operating frequency, up to 1MHz. This pin replaces SHDN on -SYNC option parts. See Synchronizing section in Applications Information for details. When not in use, this pin should be grounded. SHDN: The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. Actually, this pin has two separate thresholds, one at 2.38V to disable switching, and a second at 0.4V to force complete micropower shutdown. The 2.38V threshold functions as an accurate undervoltage lockout (UVLO). This is sometimes used to prevent the regulator from operating until the input votlage has reached a predeter- mined level. V C: The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. This pin sits at about 1V for very light loads and 2V at maximum load. It can be driven to ground to shut off the regulator, but if driven high, current must be limited to 4mA.

and output capacitor, then an abrupt 180° shift will occur. and also gives much quicker transient response. than the input voltage, allowing switch to be saturated. S flip-flop to turn the switch on. Figure 1. Block Diagram

voltage and provide several overload protection features. Foldback graph in Typical Performance Characteristics). load current with output voltage less than 50% of final value. the LT1506 to lose control of current limit. Figure 2. Frequency and Current Limit Foldback

1506 F02

APPLICATIONS INFORMATIONWU UU The internal circuitry which forces reduced switching frequency also causes current to flow out of the feedback pin when output voltage is low. The equivalent circuitry is shown in Figure 2. Q1 is completely off during normal operation. If the FB pin falls below 1V, Q1 begins to conduct current and reduces frequency at the rate of approximately 5kHz/µA. To ensure adequate frequency foldback (under worst-case short-circuit conditions), the external divider Thevinin resistance must be low enough to pull 150µA out of the FB pin with 0.6V on the pin (R DIV ≤ 4k). The net result is that reductions in frequency and current limit are affected by output voltage divider imped- ance. Although divider impedance is not critical, caution should be used if resistors are increased beyond the suggested values and short-circuit conditions will occur with high input voltage . High frequency pickup will increase and the protection accorded by frequency and current foldback will decrease. MAXIMUM OUTPUT LOAD CURRENT Maximum load current for a buck converter is limited by the maximum switch current rating (I P) of the LT1506. This current rating is 4.5A up to 50% duty cycle (DC), decreasing to 3.7A at 80% duty cycle. This is shown graphically in Typical Performance Characteristics and as shown in the formula below: I P = 4.5A for DC ≤ 50% DC = Duty cycle = VOUT/VIN Example: with VOUT = 5V, VIN = 8V; DC = 5/8 = 0.625, and; Current rating decreases with duty cycle because the LT1506 has internal slope compensation to prevent cur- rent mode subharmonic switching. For more details, read regard because it has nonlinear slope compensation which gives better compensation with less reduction in current limit. Maximum load current would be equal to maximum switch current for an infinitely large inductor , but with finite inductor size, maximum load current is reduced by one-half peak-to-peak inductor current. The following formula assumes continuous mode operation, implying that the term on the right is less than one-half of I IOUT(MAX) = Continuous Mode For the conditions above and L = 3.3µH, I A OUT MAX( ) − =− () −() () =− = 58 5 2 3 3 10 500 10 8 43 05 7 37 3 .· · .. . At VIN = 15V, duty cycle is 33%, so IP is just equal to a fixed 4.5A, and IOUT(MAX) is equal to: 51 5 5 2 3 3 10 500 10 15 45 10 1 34 9 .· · .. . − () −() () =− = A Note that there is less load current available at the higher input voltage because inductor ripple current increases. This is not always the case. Certain combinations of inductor value and input voltage range may yield lower available load current at the lowest input voltage due to reduced peak switch current at high duty cycles. If load current is close to the maximum available, please check maximum available current at both input voltage ex- tremes. To calculate actual peak switch current with a given set of conditions, use: II VV V LfV SW PEAK OUT OUT IN OUT IN ( ) =+ −() () ( ) ( )2 I P − () −() () ( ) ( ) VV V LfV OUT IN OUT IN2

APPLICATIONS INFORMATIONWU UU CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR For most applications the output inductor will fall in the range of 3µH to 20µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the LT1506 switch, which has a 4.5A limit. Higher values also reduce output ripple voltage, and reduce core loss. Graphs in the Typical Performance Characteristics section show maximum output load current versus inductor size and input voltage. A second graph shows core loss versus inductor size for various core materials. When choosing an inductor you might have to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, EMI, fault cur- rent in the inductor, saturation, and of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Choose a value in microhenries from the graphs of maximum load current and core loss. Choosing a small inductor with lighter loads may result in discontinuous mode of operation, but the LT1506 is designed to work well in either mode. Keep in mind that lower core loss means higher cost, at least for closed core geometries like toroids. The core loss graphs show absolute loss for a 3.3V output, so actual percent losses must be calculated for each situation. Assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions. If maxi- mum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 4.5A overload condition. Dead shorts will actually be more gentle on the induc- tor because the LT1506 has foldback current limiting. 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, espe- cially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall in between somewhere. The following formula assumes continu- ous mode of operation, but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. II VV V fLV PEAK OUT OUT IN OUT IN −() () ( ) ( )2 VIN = Maximum input voltage f = Switching frequency, 500kHz 3. Decide if the design can tolerate an “open” core geom- etry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent EMI problems. One would not want an open core next to a magnetic storage media, for instance! This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radia- tion will be a problem. 4. Start shopping for an inductor (see representative surface mount units in Table 2) which meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heating), and fault current (if the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts). Keep in mind that all good things like high efficiency, low profile, and high temperature operation will increase cost, sometimes dramatically. Get a quote on the cheapest unit first to calibrate yourself on price, then ask for what you really want. 5. After making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. Use the experts in the Linear Technology’s applica- tions department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest devel- opments in low profile, surface mounting, etc.

tantalum surface mount capacitors. Table 3. Surface Mount Solid Tantalum Capacitor ESR are prone to failure if they undergo high surge currents. dead shorted, do not harm the capacitors.

APPLICATIONS INFORMATIONWU UU and does not fall to less than 1/3 of nominal output voltage, foldback will not take effect. With the overloaded condi- tion, output current will increase to a typical value of 5.7A, determined by peak switch current limit of 6A. With V IN = 15V, VOUT = 4V (5V overloaded) and IOUT = 5.7A: IAD AVG() = −() = 5 7 15 4 41 8 This is safe for short periods of time, but it would be prudent to check with the diode manufacturer if continu- ous operation under these conditions must be tolerated. BOOST PIN CONSIDERATIONS For most applications, the boost components are a 0.27µF capacitor and a 1N914 or 1N4148 diode. The anode is connected to the regulated output voltage and this gener- ates a voltage across the boost capacitor nearly identical to the regulated output. In certain applications, the anode may instead be connected to the unregulated input volt- age. This could be necessary if the regulated output voltage is very low (< 3V) or if the input voltage is less than 5V. Efficiency is not affected by the capacitor value, but the capacitor should have an ESR of less than 1Ω to ensure that it can be recharged fully under the worst-case condi- tion of minimum input voltage. Almost any type of film or ceramic capacitor will work fine. For nearly all applications, a 0.27µF boost capacitor works just fine, but for the curious, more details are provided here. The size of the boost capacitor is determined by switch drive current requirements. During switch on time, drain current on the capacitor is approximately I OUT/ 50. At peak load current of 4.25A, this gives a total drain of 85mA. Capacitor ripple voltage is equal to the product of on time and drain current divided by capacitor value; ΔV = (t ON)(85mA/C). To keep capacitor ripple voltage to less than 0.6V (a slightly arbitrary number) at the worst- case condition of tON = 1.8µs, the capacitor needs to be 0.27µF. Boost capacitor ripple voltage is not a critical parameter, but if the minimum voltage across the capaci- tor drops to less than 3V, the power switch may not saturate fully and efficiency will drop. An approximate formula for absolute minimum capacitor value is: C IV V fV V MIN OUT OUT IN OUT = () ( ) () −() f = Switching frequency VOUT = Regulated output voltage VIN = Minimum input voltage This formula can yield capacitor values substantially less than 0.27µF, but it should be used with caution since it does not take into account secondary factors such as capacitor series resistance, capacitance shift with tem- perature and output overload. SHUTDOWN FUNCTION AND UNDERVOLTAGE LOCKOUT Figure 4 shows how to add undervoltage lockout (UVLO) to the LT1506. Typically, ULVO is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. ULVO prevents the regulator from operating at source voltages where these problems might occur. Threshold voltage for lockout is about 2.38V, slightly less than the internal 2.42V reference voltage. A 3.5 µA bias current flows out of the pin at threshold. This internally generated current is used to force a default high state on the shutdown pin if the pin is left open. When low shut- down current is not an issue, the error due to this current can be minimized by making R LO 10k or less. If shutdown current is an issue, RLO can be raised to 100k, but the error due to initial bias current and changes with temperature should be considered. Rk R RV V VR A LO HI LO IN LO = () −() − () 23 8 23 8 35 to 100k 25k suggested .. µ VIN = Minimum input voltage

1506 F04

Figure 4. Undervoltage Lockout input rises back to 7.5V. ΔV is therefore 1.5V and VIN = 6V.

very short and separate from the analog ground line. 500kHz triwave, so are much less critical. Figure 5. Suggested Layout (Topside Only Shown) Figure 6. High Speed Switching Path

1506 F05

1506 F06

APPLICATIONS INFORMATIONWU UU The term inside the radical has a maximum value of 0.5 when input voltage is twice output, and stays near 0.5 for a relatively wide range of input voltages. It is common practice therefore to simply use the worst-case value and assume that RMS ripple current is one half of load current. At maximum output current of 4.5A for the LT1506, the input bypass capacitor should be rated at 2.25A ripple current. Note however, that there are many secondary considerations in choosing the final ripple current rating. These include ambient temperature, average versus peak load current, equipment operating schedule, and required product lifetime. For more details, see Application Notes 19 and 46, and Design Note 95. Input Capacitor Type Some caution must be used when selecting the type of capacitor used at the input to regulators. Aluminum electrolytics are lowest cost, but are physically large to achieve adequate ripple current rating, and size con- straints (especially height), may preclude their use. Ceramic capacitors are now available in larger values, and their high ripple current and voltage rating make them ideal for input bypassing. Cost is fairly high and footprint may also be somewhat large. Solid tantalum capacitors would be a good choice, except that they have a history of occasional spectacular failures when they are subjected to large current surges during power-up. The capacitors can short and then burn with a brilliant white light and lots of nasty smoke. This phenomenon occurs in only a small percentage of units, but it has led some OEM companies to forbid their use in high surge applications. The input bypass capacitor of regulators can see these high surges when a battery or high capacitance source is connected. Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (AVX TPS series for instance, see Table 3), but even these units may fail if the input voltage surge approaches the maximum voltage rating of the capacitor. AVX recom- mends derating capacitor voltage by 2:1 for high surge applications. Larger capacitors may be necessary when the input volt- age is very close to the minimum specified on the data sheet. Small voltage dips during switch on time are not normally a problem, but at very low input voltage they may cause erratic operation because the input voltage drops below the minimum specification. Problems can also occur if the input-to-output voltage differential is near minimum. The amplitude of these dips is normally a function of capacitor ESR and ESL because the capacitive reactance is small compared to these terms. ESR tends to be the dominate term and is inversely related to physical capacitor size within a given capacitor type. SYNCHRONIZING (-SYNC Option for DD Package) The SYNC pin, is used to synchronize the internal oscilla- tor to an external signal. The SYNC input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 10% and 90%. The input can be driven directly from a logic level output. The synchronizing range is equal to initial operating frequency up to 1MHz. This means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (560kHz), not the typical operating frequency of 500kHz. Caution should be used when synchronizing above 700kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs at input voltages less than twice output voltage. Higher inductor values will tend to eliminate this problem. See Frequency Compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. Application Note 19 has more details on the theory of slope compensation. At power-up, when V C is being clamped by the FB pin (see Figure 2, Q2), the sync function is disabled. This allows the frequency foldback to operate in the shorted output con- dition. During normal operation, switching frequency is controlled by the internal oscillator until the FB pin reaches 1.5V, after which the SYNC pin becomes operational. THERMAL CALCULATIONS Power dissipation in the LT1506 chip comes from four sources: switch DC loss, switch AC loss, boost circuit current, and input quiescent current. The following

formulas show how to calculate each of these losses. Figure 9. Figure 10 shows a Bode plot of the phase and gain quency (≈16kHz) set by capacitor ESR (0.1Ω ). Figure 9. Model for Loop Response

1506 F09

Figure 10. Response from VC Pin to Output

1506 F11

Figure 11. Error Amplifier Gain and Phase Figure 12. Overall Loop Characteristics

1505 F10

1505 F12

impedance of the amplifier at frequencies above 500Hz. dominate all other effects with respect to loop response.

APPLICATIONS INFORMATIONWU UU cases, the resistor may have to be larger to get acceptable phase response, and some means must be used to control ripple voltage at the VC pin. The suggested way to do this is to add a capacitor (CF) in parallel with the RC/CC network on the VC pin. Pole frequency for this capacitor is typically set at one-fifth of switching frequency so that it provides significant attenuation of switching ripple, but does not add unacceptable phase shift at loop unity-gain frequency. With R C = 3k, C fR k pFF C () ( ) () =  () 2 500 10 3 531 3π π · How Do I Test Loop Stability? The “standard” compensation for LT1506 is a 1.5nF capacitor for CC, with RC = 0. While this compensation will work for most applications, the “optimum” value for loop compensation components depends, to various extent, on parameters which are not well controlled. These include inductor value (± 30% due to production tolerance, load current and ripple current variations), output capacitance (± 20% to ± 50% due to production tolerance, tempera- ture, aging and changes at the load), output capacitor ESR (± 200% due to production tolerance, temperature and aging), and finally, DC input voltage and output load current . This makes it important for the designer to check out the final design to ensure that it is “robust” and tolerant of all these variations. I check switching regulator loop stability by pulse loading the regulator output while observing transient response at the output, using the circuit shown in Figure 13. The regulator loop is “hit” with a small transient AC load current at a relatively low frequency, 50Hz to 1kHz. This causes the output to jump a few millivolts, then settle back to the original value, as shown in Figure 14. A well behaved loop will settle back cleanly, whereas a loop with poor phase or gain margin will “ring” as it settles. The number of rings indicates the degree of stability, and the frequency of the ringing shows the approximate unity-gain fre- quency of the loop. Amplitude of the signal is not particu- larly important, as long as the amplitude is not so high that the loop behaves nonlinearly. GMP = Transconductance of power stage = 5.3A/V GMA = Error amplifier transconductance = 2(10–3) ESR = Output capacitor ESR 2.42 = Reference voltage With VOUT = 5V and ESR = 0.03Ω , a value of 6.5k for RC would yield zero gain margin, so this represents an upper limit. There is a second limitation however which has nothing to do with theoretical small signal dynamics. This resistor sets high frequency gain of the error amplifier, including the gain at the switching frequency. If switching frequency gain is high enough, output ripple voltage will appear at the V C pin with enough amplitude to muck up proper operation of the regulator. In the marginal case, subharmonic switching occurs, as evidenced by alternat- ing pulse widths seen at the switch node. In more severe cases, the regulator squeals or hisses audibly even though the output voltage is still roughly correct. None of this will show on a theoretical Bode plot because Bode is an amplitude insensitive analysis. Tests have shown that if ripple voltage on the VC is held to less than 100mVP-P, the LT1506 will be well behaved. The formula below will give an estimate of VC ripple voltage when RC is added to the loop, assuming that RC is large compared to the reactance of CC at 500kHz. V R G V V ESR VL f C RIPPLE C MA IN OUT IN () ( ) ( ) 24. GMA = Error amplifier transconductance (2000µMho) If a computer simulation of the LT1506 showed that a series compensation resistor of 3k gave best overall loop response, with adequate gain margin, the resulting VC pin ripple voltage with V IN = 10V, V OUT = 5V, ESR = 0.1 Ω , L = 10µH, would be: V k VC RIPPLE( ) ()   −() ( ) ( ) ()  3 2 10 10 5 0 1 2 4 10 10 10 500 10 0 144

• . . This ripple voltage is high enough to possibly create subharmonic switching. In most situations a compromise value (< 2k in this case) for the resistor gives acceptable phase margin and no subharmonic problems. In other

Figure 13. Loop Stability Test Circuit

1506 F13

Figure 14. Loop Stability Check this wouldn’t matter because amplitude is not critical. temperature-dependent characteristics. the output capacitor in production. tolerance but phase margin generally hangs in there. larger switcher. The inductor size is considerably reduced.

rating, size and cost of filter capacitors. ductance prevents current hogging. loading is shared between inputs. Figure 15. Current Sharing 12A Supply

1506 F15

mum input ripple current is a 4A square wave at 600kHz. increased to 22nF (× 3) for the ceramic output capacitor. must be ripple rated for the total output current.

increasing switch current of the two remaining devices. used for sequencing multiple regulator outputs. energy is stroed in L1A only, since no current flows in L1B. Figure 16. Buck Converter with Adjustable Soft Start

1506 F17

1506 F16

Figure 17. Dual Output SEPIC Converter Information furnished by Linear Technology Corporation is believed to be accurate and reliable. tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

1506f LT/TP 1198 4K • PRINTED IN USA  LINEAR TECHNOLOGY CORPORA TION 1998 Dimensions in inches (millimeters) unless otherwise noted.PACKAGE DESCRIPTIONU RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1074/LT1076 Step-Down Switching Regulators 40V Input, 100kHz, 5A and 2A LTC

1148 High Efficiency Synchronous Step-Down Switching Regulator External FET Switches

LTC1149 High Efficiency Synchronous Step-Down Switching Regulator External FET Switches LTC1174 High Efficiency Step-Down and Inverting DC/DC Converter 0.5A, 150kHz Burst Mode TM Operation LT1176 Step-Down Switching Regulator PDIP LT1076 LT1370 High Efficiency DC/DC Converter 42V, 6A, 500kHz Switch LT1371 High Efficiency DC/DC Converter 35V, 3A, 500kHz Switch LT1372/LT1377 500kHz and 1MHz High Efficiency 1.5A Switching Regulators Boost Topology LT1374 High Efficiency Step-Down Switching Regulator 25V, 4.5A, 500kHz Switch LT1435/LT1436 High Efficiency Step-Down Converter External Switches, Low Noise Burst Mode is a trademark of Linear Technology Corporation. R Package 7-Lead Plastic DD Pak (LTC DWG # 05-08-1462) R (DD7) 0396 0.026 – 0.036 (0.660 – 0.914) 0.143 +0.012 –0.020 ()3.632 +0.305 –0.508 0.040 – 0.060 (0.330 – 0.584) 0.095 – 0.115 (2.413 – 2.921) 0.004 +0.008 –0.004 ()0.102 +0.203 –0.102 0.050 – 0.012 (1.270 – 0.305) 0.059 (1.499) TYP 0.045 – 0.055 (1.143 – 1.397) 0.165 – 0.180 (4.191 – 4.572) 0.330 – 0.370 (8.382 – 9.398) 0.060 (1.524) TYP 0.390 – 0.415 (9.906 – 10.541) 15° TYP 0.300 (7.620) 0.075 (1.905) 0.183 (4.648) 0.060 (1.524) 0.060 (1.524) 0.256 (6.502) BOTTOM VIEW OF DD PAK HATCHED AREA IS SOLDER PLATED COPPER HEAT SINK 1 2 3 4 0.150 – 0.157** (3.810 – 3.988) 8 7 6 5 0.189 – 0.197* (4.801 – 5.004) 0.228 – 0.244 0.406 – 1.270 0.010 – 0.020 0°– 8° TYP 0.008 – 0.010 (0.203 – 0.254) SO8 0996 0.053 – 0.069 (1.346 – 1.752) 0.014 – 0.019 (0.355 – 0.483) 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) TYP DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 l FAX: (408) 434-0507 l www.linear-tech.com