LMR62014

LMR62014 NSC | Alldatasheet

Document overview

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Technical content

Features

■ Input voltage range of 2.7V to 14V ■ Output voltage up to 20V ■ Switch current up to 1.4A ■ 1.6 MHz switching frequency ■ Low shutdown Iq, <1 µA ■ Cycle-by-cycle current limiting ■ Internally compensated ■ SOT23-5 packaging (2.92 x 2.84 x 1.08mm) ■ Fully enabled for WEBENCH® Power Designer Performance Benefits ■ Extremely easy to use ■ Tiny overall solution reduces system cost

Applications

■ Boost Conversions from 3.3V, 5V, and 12V Rails ■ Space Constrained Applications ■ Embedded Systems ■ LCD Displays ■ LED Applications System Performance Efficiency vs Load Current VIN = 3.3V, VOUT = 12V 30167439 Efficiency vs Load Current VIN = 5V, VOUT = 12V 30167457 © 2011 National Semiconductor Corporation 301674 www.national.com LMR62014 SIMPLE SWITCHER® 20Vout, 1.4A Step-Up Voltage Regulator in SOT-23

See NS Package Number MF05A

Ordering Information

Order Number Package Type Package Drawing Supplied As Package ID LMR62014XMFE SOT23-5 MF05A

250 Units, Tape and Reel SH1B

LMR62014XMF 1000 Units, Tape and Reel LMR62014XMFX 3000 Units, Tape and Reel Pin Descriptions Pin Name Function 1 SW Drain of the internal FET switch. 2 GND Analog and power ground. 3 FB Feedback point that connects to external resistive divider. 4 SHDN Shutdown control input. Connect to Vin if the feature is not used. 5 VIN Analog and power input. www.national.com 2 LMR62014

Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Storage Temperature Range −65°C to +150°C Operating Junction Temperature Range −40°C to +125°C Lead Temp. (Soldering, 5 sec.) 300°C Power Dissipation (Note 2) Internally Limited FB Pin Voltage −0.4V to +6V SW Pin Voltage −0.4V to +22V Input Supply Voltage −0.4V to +14.5V SHDN Pin Voltage −0.4V to VIN + 0.3V θJ-A (SOT23-5) 265°C/W ESD Rating (Note 3) Human Body Model 2 kV For soldering specifications: see product folder at www.national.com and www.national.com/ms/MS/MS- SOLDERING.pdf

Electrical Characteristics

Limits in standard typeface are for TJ = 25°C, and limits in boldface type apply over the full operating temperature range (−40°C ≤ TJ ≤ +125°C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A. Symbol Parameter Conditions Min (Note 4) Typical (Note 5) Max (Note 4) Units VIN Input Voltage 2.7 14 V VOUT (MIN) Minimum Output Voltage Under Load RL = 43Ω (Note 8) VIN = 2.7V 5.4 7 V VIN = 3.3V 8 10 VIN = 5V 13 17 RL = 15Ω (Note 8) VIN = 2.7V 3.75 5 VIN = 3.3V 5 6.5 VIN = 5V 8.75 11 ISW Switch Current Limit (Note 6) 1.8 1.4 2 A RDS(ON) Switch ON Resistance ISW = 100 mA Vin = 5V 260 400 500 mΩ ISW = 100 mA Vin = 3.3V 300 450 550 SHDNTH Shutdown Threshold Device ON 1.5 VDevice OFF 0.50 ISHDN Shutdown Pin Bias Current VSHDN = 0 0 µAVSHDN = 5V 0 2 VFB Feedback Pin Reference Voltage VIN = 3V 1.205 1.230 1.255 V IFB Feedback Pin Bias Current VFB = 1.23V 60 500 nA IQ Quiescent Current VSHDN = 5V, Switching 2 3.0 mA VSHDN = 5V, Not Switching 400 500 µAVSHDN = 0 0.024 1 FB Voltage Line Regulation 2.7V ≤ VIN ≤ 14V 0.02 %/V FSW Switching Frequency (Note 7) 1 1.6 1.85 MHz DMAX Maximum Duty Cycle (Note 7) 86 93 % IL Switch Leakage Not Switching VSW = 5V 1 µA Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions. Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, TJ(MAX) = 125° C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature for designs using this device can be calculated using the formula: 3 www.national.com LMR62014

If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as required to maintain a safe junction temperature. Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Note 4: Limits are guaranteed by testing, statistical correlation, or design. Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value of the parameter at room temperature. Note 6: Switch current limit is dependent on duty cycle (see Typical Performance Characteristics). Note 7: Guaranteed limits are the same for Vin = 3.3V input. Note 8: L = 10 µH, COUT = 4.7 µF, duty cycle = maximum www.national.com 4 LMR62014

Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN. Iq Vin (Active) vs Temperature 30167402 Oscillator Frequency vs Temperature 30167405 Max. Duty Cycle vs Temperature 30167407 Iq Vin (Idle) vs Temperature 30167425 Feedback Bias Current vs Temperature 30167426 Feedback Voltage vs Temperature 30167427 5 www.national.com LMR62014

RDS(ON) vs Temperature 30167428 Current Limit vs Temperature 30167429 RDS(ON) vs VIN 30167452 Efficiency vs Load Current VIN = 2.7V, VOUT = 5V 30167435 Efficiency vs Load Current VIN = 3.3V, VOUT = 5V 30167436 Efficiency vs Load Current VIN = 4.2V, VOUT = 5V 30167437 www.national.com 6 LMR62014

Efficiency vs Load Current VIN = 2.7V, VOUT = 12V 30167438 Efficiency vs Load Current VIN = 3.3V, VOUT = 12V 30167439 Efficiency vs Load Current VIN = 5V, VOUT = 12V 30167440 Efficiency vs Load Current VIN = 5V, VOUT = 18V 30167441 7 www.national.com LMR62014

The LMR62014 switching regulator is a current-mode boost converter operating at fixed frequency of 1.6 MHz. The use of SOT-23 package, made possible by the minimal power loss of the internal 1.4A switch, and use of small in- ductors and capacitors result in the industry's highest power density. The LMR62014 is capable of greater than 90% duty cycle, making it ideal for boosting to voltages up to 20V. These parts have a logic-level shutdown pin that can be used to reduce quiescent current and extend battery life. Protection is provided through cycle-by-cycle current limiting and thermal shutdown. Internal compensation simplifies de- sign and reduces component count. Theory of Operation The LMR62014 is a switching converter IC that operates at a fixed frequency (1.6 MHz) for fast transient response over a wide input voltage range and incorporates pulse-by-pulse current limiting protection. Because this is current mode con- trol, a 33 mΩ sense resistor in series with the switch FET is used to provide a voltage (which is proportional to the FET current) to both the input of the pulse width modulation (PWM) comparator and the current limit amplifier. At the beginning of each cycle, the S-R latch turns on the FET. As the current through the FET increases, a voltage (propor- tional to this current) is summed with the ramp coming from the ramp generator and then fed into the input of the PWM comparator. When this voltage exceeds the voltage on the other input (coming from the Gm amplifier), the latch resets and turns the FET off. Since the signal coming from the Gm amplifier is derived from the feedback (which samples the voltage at the output), the action of the PWM comparator constantly sets the correct peak current through the FET to keep the output voltage in regulation. Q1 and Q2 along with R3 - R6 form a bandgap voltage refer- ence used by the IC to hold the output in regulation. The currents flowing through Q1 and Q2 will be equal, and the feedback loop will adjust the regulated output to maintain this. Because of this, the regulated output is always maintained at a voltage level equal to the voltage at the FB node "multiplied up" by the ratio of the output resistive divider. The current limit comparator feeds directly into the flip-flop that drives the switch FET. If the FET current reaches the limit threshold, the FET is turned off and the cycle terminated until the next clock pulse. The current limit input terminates the pulse regardless of the status of the output of the PWM com- parator. Application Hints SELECTING THE EXTERNAL CAPACITORS The best capacitors for use with the LMR62014 are multi-lay- er ceramic capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency which makes them optimum for use with high frequency switching converters. When selecting a ceramic capacitor, only X5R and X7R di- electric types should be used. Other types such as Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of rated capacitance in many typical applica- tions. Always consult capacitor manufacturer’s data curves before selecting a capacitor. SELECTING THE OUTPUT CAPACITOR A single ceramic capacitor of value 4.7 µF to 10 µF will provide sufficient output capacitance for most applications. If larger amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Os- con can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500 kHz due to significant ringing and temperature rise due to self-heating from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability. www.national.com 8 LMR62014

In general, if electrolytics are used, it is recommended that they be paralleled with ceramic capacitors to reduce ringing, switching losses, and output voltage ripple. SELECTING THE INPUT CAPACITOR An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nomi- nal value of 2.2 µF, but larger values can be used. Since this capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other circuitry. FEED-FORWARD COMPENSATION Although internally compensated, the feed-forward capacitor Cf is required for stability (see Basic Application Circuit ). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for the zero fz should be approximately 6 kHz. Cf can be calculated using the for- mula: Cf = 1 / (2 X π X R1 X fz) SELECTING DIODES The external diode used in the typical application should be a Schottky diode.The diode must be rated to handle the max- imum output voltage and load current. A 20V diode such as the MBR0520 is recommended. The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceed- ing 0.5A average, a Toshiba CRS08 can be used. LAYOUT HINTS High frequency switching regulators require very careful lay- out of components in order to get stable operation and low noise. All components must be as close as possible to the LMR62014 device. It is recommended that a 4-layer PCB be used so that internal ground planes are available. As an example, a recommended layout of components is shown: 30167416 Recommended PCB Component Layout Some additional guidelines to be observed: 1. Keep the path between L1, D1, and C2 extremely short. Parasitic trace inductance in series with D1 and C2 will increase noise and ringing. 2. The feedback components R1, R2 and CF must be kept close to the FB pin of U1 to prevent noise injection on the FB pin trace. 3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1, as well as the negative sides of capacitors C1 and C2. SETTING THE OUTPUT VOLTAGE The output voltage is set using the external resistors R1 and R2 (see Basic Application Circuit). A value of approximately 13.3 kΩ is recommended for R2 to establish a divider current of approximately 92 µA. R1 is calculated using the formula: R1 = R2 X (VOUT/1.23 − 1) 9 www.national.com LMR62014

The maximum duty cycle of the switching regulator deter- mines the maximum boost ratio of output-to-input voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is defined as: This applies for continuous mode operation. INDUCTANCE VALUE The first question we are usually asked is: “How small can I make the inductor?” (because they are the largest sized com- ponent and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean more load power can be delivered because the energy stored during each switching cycle is: E = L/2 X (lp)2 Where “lp” is the peak inductor current. An important point to observe is that the LMR62014 will limit its switch current based on peak current. This means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load. Conversely, using too little inductance may limit the amount of load current which can be drawn from the output. Best performance is usually obtained when the converter is operated in “continuous” mode at the load current range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing the in- ductor current to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous op- eration as the output load is reduced far enough, but a larger inductor stays “continuous” over a wider load current range. To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10 µH inductor) will be ana- lyzed. We will assume: VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V Since the frequency is 1.6 MHz (nominal), the period is ap- proximately 0.625 µs. The duty cycle will be 62.5%, which means the ON time of the switch is 0.390 µs. It should be noted that when the switch is ON, the voltage across the in- ductor is approximately 4.5V. Using the equation: V = L (di/dt) We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON time. Using these facts, we can then show what the inductor current will look like dur- ing operation: 30167418 10 µH Inductor Current, 5V–12V Boost (LMR62014X) During the 0.390 µs ON time, the inductor current ramps up 0.176A and ramps down an equal amount during the OFF time. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to about 33 mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode. A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and continuous operation will be main- tained at the typical load current values. MAXIMUM SWITCH CURRENT The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is illustrated in the graphs below which show typical val- ues of switch current as a function of effective (actual) duty cycle: www.national.com 10 LMR62014

Switch Current Limit vs Duty Cycle CALCULATING LOAD CURRENT As shown in the figure which depicts inductor current, the load current is related to the average inductor current by the rela- tion: ILOAD = IIND(AVG) x (1 - DC) Where "DC" is the duty cycle of the application. The switch current can be found by: ISW = IIND(AVG) + ½ (IRIPPLE) Inductor ripple current is dependent on inductance, duty cy- cle, input voltage and frequency: IRIPPLE = DC x (VIN-VSW) / (f x L) combining all terms, we can develop an expression which al- lows the maximum available load current to be calculated: The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses of the FET and diode. For actual load current in typical applications, we took bench data for various input and output voltages that displayed the maximum load current available for a typical device in graph form: 30167448 Max. Load Current (typ) vs VIN DESIGN PARAMETERS VSW AND ISW The value of the FET "ON" voltage (referred to as VSW in the equations) is dependent on load current. A good approxima- tion can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current. FET on resistance increases at V IN values below 5V, since the internal N-FET has less gate voltage in this input voltage range (see Typical performance Characteristics curves). Above VIN = 5V, the FET gate voltage is internally clamped to 5V. The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see Typical performance Characteristics curves. THERMAL CONSIDERATIONS At higher duty cycles, the increased ON time of the FET means the maximum output current will be determined by power dissipation within the LMR62014 FET switch. The switch power dissipation from ON-state conduction is calcu- lated by: P(SW) = DC x IIND(AVE)2 x RDS(ON) There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation. INDUCTOR SUPPLIERS Recommended suppliers of inductors for this product include, but are not limited to Sumida, Coilcraft, Panasonic, TDK and Murata. When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation at peak currents. A suitable core type must be used to mini- mize core (switching) losses, and wire power losses must be considered when selecting the current rating. SHUTDOWN PIN OPERATION The device is turned off by pulling the shutdown pin low. If this function is not going to be used, the pin should be tied directly to VIN. If the SHDN function will be needed, a pull-up resistor must be used to VIN (approximately 50k-100kΩ recommend- ed). The SHDN pin must not be left unterminated. 11 www.national.com LMR62014

FIGURE 1. Flash LED Application

Physical Dimensions inches (millimeters) unless otherwise noted 5-Lead SOT-23 Package Order Number LMR62014XMF, LMR62014XMFX 13 www.national.com LMR62014

LMR62014 SIMPLE SWITCHER® 20Vout, 1.4A Step-Up Voltage Regulator in SOT-23 For more National Semiconductor product information and proven design tools, visit the following Web sites at: www.national.com Products Design Support Amplifiers www.national.com/amplifiers WEBENCH® Tools www.national.com/webench Audio www.national.com/audio App Notes www.national.com/appnotes Clock and Timing www.national.com/timing Reference Designs www.national.com/refdesigns Data Converters www.national.com/adc Samples www.national.com/samples Interface www.national.com/interface Eval Boards www.national.com/evalboards LVDS www.national.com/lvds Packaging www.national.com/packaging Power Management www.national.com/power Green Compliance www.national.com/quality/green Switching Regulators www.national.com/switchers Distributors www.national.com/contacts LDOs www.national.com/ldo Quality and Reliability www.national.com/quality LED Lighting www.national.com/led Feedback/Support www.national.com/feedback Voltage References www.national.com/vref Design Made Easy www.national.com/easy PowerWise® Solutions www.national.com/powerwise Applications & Markets www.national.com/solutions Serial Digital Interface (SDI) www.national.com/sdi Mil/Aero www.national.com/milaero Temperature Sensors www.national.com/tempsensors SolarMagic™ www.national.com/solarmagic PLL/VCO www.national.com/wireless PowerWise® Design University www.national.com/training THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS, IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS DOCUMENT. TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS. EXCEPT AS PROVIDED IN NATIONAL’S TERMS AND CONDITIONS OF SALE FOR SUCH PRODUCTS, NATIONAL ASSUMES NO LIABILITY WHATSOEVER, AND NATIONAL DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY RELATING TO THE SALE AND/OR USE OF NATIONAL PRODUCTS INCLUDING LIABILITY OR WARRANTIES RELATING TO FITNESS FOR A PARTICULAR PURPOSE, MERCHANTABILITY, OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY RIGHT. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness. National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation. All other brand or product names may be trademarks or registered trademarks of their respective holders. Copyright© 2011 National Semiconductor Corporation For the most current product information visit us at www.national.com National Semiconductor Americas Technical Support Center Email: support@nsc.com Tel: 1-800-272-9959 National Semiconductor Europe Technical Support Center Email: europe.support@nsc.com National Semiconductor Asia Pacific Technical Support Center Email: ap.support@nsc.com National Semiconductor Japan Technical Support Center Email: jpn.feedback@nsc.com www.national.com