LC5910S

LC5910S SANKEN | Alldatasheet

Document overview

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Technical content

Datasheet sections

• 9.1 Operation in Critical Current Mode (CRM)---------------------------------------------------
• 9.3 Reference Voltage Select Function for LED Current Limit --------------------------------
• 9.4 LED Current Setting -----------------------------------------------------------------------------
• 9.5 Bottom Detection Function (Off-time Period Termination Detection)------------------
• 9.6 Maximum On-time --------------------------------------------------------------------------------
• 9.7 PWM Dimming ------------------------------------------------------------------------------------
• 9.8 External Power MOSFET Gate Drive --------------------------------------------------------
• 9.9 Overvoltage Protection (OVP) ------------------------------------------------------------------
• 9.11 Thermal Shutdown (TSD) -----------------------------------------------------------------------
• 10.1 Inductor Setting -----------------------------------------------------------------------------------
• 10.2 Input Smoothing Electrolytic Capacitor, CIN, Setting --------------------------------------
• 10.3 Output Smoothing Capacitor, COUT, Setting -------------------------------------------------
• 10.4 Current Detection Resistor, RCS, Setting -----------------------------------------------------
• 11.1 External Components ----------------------------------------------------------------------------
• 11.1.1 Inductor ---------------------------------------------------------------------------------------
• 11.1.2 Input and Output Smoothing Electrolytic Capacitor --------------------------------
• 11.1.3 Current Detection Resistor ----------------------------------------------------------------
• 11.1.4 Freewheeling Diode -------------------------------------------------------------------------
• 11.1.5 External Power MOSFET -----------------------------------------------------------------
• 11.2 PCB Trace Layout --------------------------------------------------------------------------------

Features

• Critical Current Mode (CRM) Control (Low switching loss, low noise, reduction in the number of components)
• Enable Function
• Bottom Detection Voltage 100 mV (typ.) LED Control Parts
• PWM Dimming Function
• Reference Voltage Selection Function for LED Current Limiting (three levels, 75% to 100%)
• Current Reference Accuracy: ±1% Protections
• LED Output Short Circuit Protection: Pulse-by-pulse
• Current Det ection Short Circuit Protection : Aut o- restart
• Input Voltage Drop Protection: Auto-restart
• Overvoltage Protection ( OVP): Auto -restart, fault output
• Thermal Shutdo wn (TSD) with Hysteresis: Auto - restart Typical Application CS VCC GND OUT SEL PWM FAULT BD RBD1 CBD CO UTDS VIN CIN CVCC RBD2 RCS LED Fault Signal PWM Signal LED Current Switch Signal VCC Package SOP8 Not to scale Specification
• Recommended Supply Voltage : 12 V (min.) to 17 V (max.)
• Circuit Current at Standby: 100 μA (typ.) Application
• LED Back Light
• LED Lighting Equipment
• LED Bulbs

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 3 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 1. Absolute Maximum Ratings Current polarities are defined as follows: current going into the IC (sinking) is positive current (+); and current coming out of the IC (sourcing) is negative current (−). Unless specifically noted, TA = 25 °C. Parameter Symbol Pins Rating Unit CS Pin Voltage VCS 1−3 −0.3 to 18.0 V VCC Pin Voltage VCC 2−3 −0.3 to 18.0 V OUT Pin Voltage VOUT 4−3 −0.3 to 18.0 V BD Pin Voltage VBD 5−3 −1.0 to 18.0 V FAULT Pin Voltage VFAULT 6−3 −0.3 to 18.0 V PWM Pin Voltage VPWM 7−3 −0.3 to 18.0 V SEL Pin Voltage VSEL 8−3 −0.3 to 18.0 V OUT Pin Sink/Source Current IOUT 4−3 −1.5 to 1.5 A BD Pin Sink/Source Current IBD 5−3 −4.0 to 4.0 mA Allowable Dissipation(1)(2) PD ― 1.2 W Thermal Resistance between Junction and Lead (pin 3) θj- Pin ― 65 °C/W Thermal Resistance between Junction and Ambient (2) θj-A ― 95 °C/W Junction Temperature(3) Tj ― 150 °C Operating Ambient Temperature(1) Top ― −40 to 125 °C Storage Temperature Tstg ― −40 to 150 °C (1) Limited by junction temperature. (2) The IC is mounted on the glass-epoxy board (40 × 40 mm) with copper area (25 × 25 mm). (3) The temperature of thermal shutdown operation is 150 °C (typ.). 2. Recommended Operating Range The recommended operating range shows the operating conditions that are required for maintaining the normal circuit function shown in the electrical characteristics. The IC should be used within the recommended operating range. Unless specifically noted, TA = 25 °C. Parameter Symbol Min. Typ. Max. Unit Input Voltage Range VCC 12 ― 17 V Operating Ambient Temperature Range TOP −40 ― 85 °C

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 4 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 3. Electrical Characteristics Current polarities are defined as follows: current going into the IC (sinking) is positive current (+); and current coming out of the IC (sourcing) is negative current (−). Unless specifically noted, TA = 25 °C, VCC = 12 V. Parameter Symbol Conditions Pins Min. Typ. Max. Unit Operation Start Voltage VCC(ON) 2−3 10.0 11.0 12.0 V Operation Stop Voltage VCC(OFF) 2−3 9.0 10.0 11.0 V Operation Start Stop Hysteresis VCC(HYS) 2−3 0.5 1.0 3.0 V Circuit Current in Operation ICC(ON) 2−3 ― 1.1 1.8 mA Circuit Current in Non-operation ICC(OFF) VPWM = 0 V 2−3 ― 100 170 μA PWM Pin On Threshold Voltage VPWM(ON) 7−3 1.6 2.0 2.4 V PWM Pin Off Threshold Voltage VPWM(OFF) 7−3 0.8 1.1 1.4 V PWM Pin Sink Current IPWM VPWM = 3.3 V 7−3 0.50 1.25 3.00 μA Standby Mode Switching Time tST_BY VPWM = 3.3 V → 0 V 7−3 20 36 57 ms PWM Disable VSEL Pin Voltage VSEL_DIS VPWM = 3.3 V 8−3 0 ― 0.4 V CS Pin Reference Voltage 1 VCS1 1−3 742.5 750 757.5 mV CS Pin Reference Voltage 2 VCS2 1−3 990 1000 1010 mV CS Pin Reference Voltage 3 VCS3 1−3 1089 1100 1111 mV CS Pin Overvoltage Protection Threshold Voltage 1 VCSOVP1 VPWM = 3.3 V 1−3 2.4 2.7 3.1 V CS Pin Overvoltage Protection Threshold Voltage 2 VCSOVP2 VPWM = 0 V 1−3 0.40 0.72 1.00 V CS Pin Blanking Time tLEB 1−3 320 ns Overvoltage Protection Restart Time tRESTART 1−3 5.0 11.0 19.0 ms Maximum On-time tONMAX VCS = 0 V 4−3 15.0 20.0 29.0 μs BD Pin Detection Voltage VBD(L) 5−3 70 100 170 mV BD Pin Detection Hysteresis Voltage VBD(HYS) 5−3 ― 100 ― mV BD Pin Time Out Time 1 tBD(TOUT1) VCS = 1.5 V 5−3 15.0 20.0 29.0 μs BD Pin Time Out Time 2 tBD(TOUT2) VCS = 0 V 5−3 300 570 900 μs BD Pin Detection Mask Time tBDMSK 5−3 0.3 0.62 1.10 μs OUT Pin On-resistance (High)* ROUT(HIGH) IOUT = 30 mA 4−3 ― 30 ― Ω OUT Pin On-resistance (Low)* ROUT(LOW) IOUT = −30 mA 4−3 ― 15 ― Ω FAULT Pin Output Resistance RFAULT 6−3 300 610 1000 Ω Thermal Shutdown Operation Temperature* TSD — ― 150 ― °C Thermal Shutdown Temperature Hysteresis* TSD(HYS) — ― 33 ― °C * Ensured by design, not product tested.

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 5 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 4. Block Diagram PWM VCC SEL CS OUT GND BD FAULT UVLO

11 V / 10 V

Reg. UVLO WDT PWMo = L → 36 ms Standby Reg. EN T SD 150 °C / 117 °C Maximum On-time 20 µs 1.25 µA 1.25 µA CS Reference Selector VSEL ≤ 0.4 V, Disable 0.75V ≤ VSEL ≤ 1.25 V, VCS1 = 0.75 V 1.75V ≤ VSEL ≤ 2.25 V, VCS2 = 1.00 V 2.75V ≤ VSEL ≤ 3.25 V, VCS3 = 1.10 V VCS 2.7 V 0.72 V LEB R S Q 100 mV / 200 mV BD Time-out 20 µs / 570 µs BD Blanking OUT H→L GAT E tON_MAX VCC PWMo PWMo CS_OVP Figure 4-1. LC5910S Block Diagram 5. Pin Configuration Definitions CS VCC GND OUT SEL PWM FAULT BD Pin Name Descriptions 1 CS Current Detection Signal Input.

2 VCC

Power supply voltage input (within the recommended operating range, 12 V to 17 V). A capacitor is connected between the VCC and GND pins in order to supply stable power supply voltage to the IC.

3 GND Ground

4 OUT Signal output for the gate drive of the power

MOSFET.

5 BD The resistor for adjusting the off -time is

connected.

6 FAULT

Fault signal output. This pin outputs the signal ( the signal can be used for the setting to stop the pr e converters such as PFC.)

7 PWM

PWM dimming signal input. This pin inputs the PWM signal for dimming the LED string.

8 SEL

Reference voltage select signal input. This pin inputs the DC voltage signal to select the internal reference voltage that contr ols the LED current.

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 6 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 6. Typical Applications CS VCC GND OUT SEL PWM FAULT BD RBD1 CBD CO UTDS Power Supply Input, VIN CIN CVCC RBD2 RCS LED Fault Signal PWM Signal LED Current Switch Signal VCC IL Figure 6-1. LC5910S Typical Application

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 7 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 7. Physical Dimension

• SOP8 Package
• SOP8 Land Pattern Example 1.27 0.95 0.64 5.25 Dimensions in millimeters NOTES: - Dimensions in millimeters - Bare lead frame: Pb-free (RoHS compliant)

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 8 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 8. Marking Diagram Part Number LC5910 S K Y M D Control Number Lot Number: Y is the last digit of the year of manufacture (0 to 9) M is the month of the year (1 to 9, O, N or D) D is a period of days, 1: the first 10 days of the month (1st to 10th) 2: the second 10 days of the month (11th to 20th) 3: the last 10-11 days of the month (21st to 31st)

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 9 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 9. Operational Description All the characteristic values given in this section are typical values, unless they are specified as minimum or maximum. Current polarities are defined as follows: current going in to the IC (sinking) is positive current (+); and current coming out of the IC (sourcing) is negative current (−).

9.1 Operation in Critical Current Mode

(CRM) Figure 9-1 shows the LED drive circuit example using the LC5910. CS VCC GND OUT SEL PWM FAULT BD RBD1 CBD CO UT DS VIN CIN CVCC RBD2 RCS LED Fault Signal PWM Signal LED Current Switch Signal VCC IL ID IS VDS Figure 9-1. LED Drive Circuit Example using the LC5910S The IC uses the Critical Current Mode (CRM) control. To drive the LED string, the IC controls the on -time and off-time of the external power MOSFET as follows:

• On-time period: When the CS pin detects the peak of the extern al inductor current, IL, the on-time period is terminated..
• Off-time period: When the BD pin detects decrease of the voltage, VDS, between drain and source of the external power MOSFET, the off-time period is terminated. As a result, the waveform of the in ductor current, I L, becomes triangular as shown in Figure 9 -2. T he LED current, ILED, is half of the peak value of the inductor current, IL. The IC controls the peak value of the inductor current, IL, and stabilizes the LED current,ILED. See Figure 4-1 and Figure 6-1 for circuit symbols. The inductor current, IL, is converted into a voltage by the current detection resistor, R CS, and input to the CS pin. When the voltage becomes equal to the internal reference voltage, VCS, the OUT pin output is turned off. The off-time period is set by the V DS waveform that is input to the BD pin (see Section 9.5 ). When the BD pin voltage decreases to the BD pin detection voltage, VBD(L), the off-time period is terminated and the turn -on of the next cycle is started. As described above, the IC always turns on at the VDS of the power MOSFET de creasing in the CRM control. Therefore, the CRM control reduces the sw itching loss and noise more than the PWM continuous current mode (CCM) which switches hard. The CRM control requires no phase compensation and fewer components, resulting in the cost reduction in system. The oscillation frequency of the CRM control depends on the input voltage. VDS ID IS IL t ILED VBD(L) Turn-on timming tOFF tON VCS Turn-off timming Figure 9-2. CRM Control of the LC5910S

9.2 Enable

The IC has the enable function . When the SEL pin voltage decreases 0.4 V or lower, the OUT pin signal is fixed low. A pull -down resistor is connected internally between the SEL and the GND pins.

9.3 Reference Voltage Select Function for

The LED current, I LED, is controll ed using the peak value o f the inductor current, I L. IL is converted into a voltage by the current detection resistor, R CS, and input to the CS pin. The IC controls the voltage to be equal to the reference voltage of the internal comparator, V CS. The refer ence voltage of the comparator can be set to three levels according to the DC voltage applied to the SEL pin (see Table 9 -1). The function enables altering the LED current with the resistance of the current detection resistor, R CS, fixed.

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 10 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 If the different frequency switching noises of the other converter on PCB superimposes on the SEL pin , the RC filter shown in Figure 9-3 is required. The RC filter must be connected close to the SEL pin . Due to the delay caused by the combination of RSEL and CSEL, it take s time for the SEL pin voltage to settle at the determined VSEL after the DC voltage is applied to the SEL pin. Be sure to co nfirm the actual operation and adjust the constants. Table 9-1. The Relation between SEL Pin Voltage and Internal Reference Voltage SEL Pin Voltage, VSEL Internal Reference Voltage, VCS (typ.) VSEL ≤ 0.40 V (Enable state) 0.75V ≤ VSEL ≤ 1.25 V VCS1 = 750 mV 1.75V ≤ VSEL ≤ 2.25 V VCS2 = 1000 mV 2.75V ≤ VSEL ≤ 3.25 V VCS3 = 1100 mV GND SEL RSEL DC Voltage CSEL LC5910S VSEL Figure 9-3. RC filter of SEL Pin

9.4 LED Current Setting

As shown in Figure 9 -2, t he LED current, I LED, is controlled so that the peak of the inductor current, I L, at the power MOSFET turn- on is equal to the reference voltage, V CS, set by the SEL pin. When t he power MOSFET turns off, the energy stored in the inductor is regenerated through the f reewheeling diode, DS, and the power MOSFET turns on at the timing of the end of energy regeneration. By repeating this, the waveform of the IL becomes triangular with the current determined by the RCS and the VCS at the top, and the I LED is controlled to be half of the peak value of I L. The ILED is calculated by the Equation (1). ILED = 0.5 × VCS RCS (1) where: VCS is the reference voltage set by the SEL pin (see Section 9.2), and RCS is the resistance of the current detection resistor.

9.5 Bottom Detection Function (Off-time

Period Termination Detection) For the off -time period setting, the voltage between drain and source, V DS, is used. The V DS is input to the BD pin from the drain pin through a coupling capacitor and a voltage dividing resistor. As shown in Figure 9 -3, when the power MOSFET turns off, the energy stored in the inductor is regenerated through the f reewheeling diode , D S. When the current flowing in the D S becomes zero, the V DS decreases due to the free oscillation by the inductor, L, and the voltage resonance of parasitic capacitance between drain and source, Cds. When the BD pin voltage decreases to the BD pin detection voltage, VBD(L) = 100 mV, or lower, the function terminates the off -time period and turn on the power MOSFET. The BD pin detection mask time, tBDMASK = 0.62 μs (typ.), prevents misdetection due to surge and ringing waveform immediately after the power MOSFET turns off. If the BD pin voltage maintains above VBD(L) during ≥tBD(TOUT1) of 20 μs or more after the power MO SFET turns off, the power MOSFET is forcibly turned on. The standard constant for the BD pin is as follows: When VIN = 160 V and VLED = 130 V, CBD = 15 to 100 pF, RBD1 = 63 kΩ, and RBD2 = 2.2 kΩ. If RBD1 decreases with fixing R BD2, the turn- on delay increases.. CBDF, which is f or noise countermeasure, should be connected as needed. However, adding the CBDF increases the turn-on delay. Since the bottom detect ion voltage of V DS of the BD pin is V BD(L) = 100 mV , it is strictly detected a little earlier than the bottom point. ( Depending on the constant of the BD pin detection circuit, the turn -on is delayed and the bottom point of V DS may be adjusted to approximately 0 V). GND BD RBD1 CBDF LC5910S CBD RCS 100 mV / 200 mV BD Time-out 20 µs / 600 µs BD Blanking OUT H→L RBD2 Figure 9-4. BD Pin Connection

9.6 Maximum On-time

In case the current detection resistor, R CS, is shorted or the supply voltage, V IN, decreases, the OUT pin

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 11 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 output signal of the is kept in the logic high state ; and the power MOSFET may be destroyed . To prevent this event, the OUT pin output signal is lim ited by the maximum on-time. When the on- time period exceeds the maximum on - time, t ONMAX = 20 μs (typ.), the OU T pin output signal becomes logic low, and outputs the logic high signal again in tBD(TOUT2) = 570 μs.

9.7 PWM Dimming

The PWM pin is the input pin of the PWM dimming signal. The PWM dimming signal that satisfies the following conditions is input to the PWM pin: On threshold voltage, VPWM(ON) = 2 V, and Off threshold voltage, VPWM(OFF) = 1.1 V. The voltage of 2.7 V to 3.3 V is recommended for the peak voltage of the PWM signal. The pull -down resistor is connected between the PWM and the GND pins. When the period during which the PWM pin voltage is lower than V PWM(OFF) is 36 ms or more, the LC5910S enters standby mode. In standby mode, the current consumption of the IC is reduced. When the PWM pin voltage exceeds the V PWM(ON), the IC recovers fro m the standby mode. In order to decrease the dv/dt of the PWM dimming signal, the RC filter circuit should be connected to the PWM pin as shown in Figure 9 -5. However, care should be taken when signals with high frequency are input. GND PWM RPWM PWM signal CPWM LC5910S Figure 9-5. RC filter of PWM Pin

9.8 External Power MOSFET Gate Drive

Figure 9 -6 shows the peripheral circuit of the OUT pin. The OUT pin is for driving the external power MOSFET gate. The peak output voltage of the OUT pin depends on the VCC pin voltage. When the input voltage of the VCC pin is 17 V, the output voltage of the OUT pin is also around 17 V. Therefore, it is re quired to choose the external power MOSFET having the gate breakdown voltage of ± 20 V or ± 30 V, and choose the power MOSFET such that the threshold voltage between gate and source meets V GS(TH) < VOUT in all operating temperature range. In a general EMI noise countermeasure, the dv/dt of the gate voltage of the power MOSFET is decreased to slow the switching speed. Note that, however, the switching loss increases. The circuit resistance of the drive output incorporated in the LC5910S is as follows: 30 Ω (typ.) for source, and 15 Ω (typ.) for sink. The resistance above can not be changed. Therefore, the switching speed is adjusted by external components as shown in Figure 9-6 and Figure 9-7. The components in Figure 9-6 of the gate drive circuit of the power MOSFET should be chosen as follows: R G1 is around 100 Ω, RG2 is around 10 Ω, and DG is a Schottky diode having the same breakdown voltage as the gate breakdown voltage of the external power MOSFET. Figure 9 -7 shows a method of discharging with the transistor, Q G at the power MOSFET turn -off. At turn- on, the gate of the power MOSFET is charged via RG1, DG, and R G2 from the OUT pin. At turn -off, the bias voltage of Q G is the forward voltage, V F, of D G. Decreasing RG2 increases the switching speed at turn-off. In order to prevent malfunction of the IC due to the rapid dv/dt of the drain voltage o f the power MOSFET , R GS is added between gate and ground of the power MOSFET as needed. RGS is around 10 kΩ to 100 kΩ. GND RG2 LC5910S RCSRGS OUT DG RG1 1CS Figure 9-6. OUT Pin Peripheral Circuit (1) GND RG2 LC5910S RCSRGS OUT DGRG1 1CS QG Figure 9-7. OUT Pin Peripheral Circuit (2)

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 12 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016

9.9 Overvoltage Protection (OVP)

When the inductance is saturated or shorted, the voltage between both ends of the output current resistor, RCS, increases, resulting in the increase in the CS pin voltage. When the CS pin voltage meets either of the following conditions, the overvoltage protection (OVP) is activated.

• The CS pin voltage is the overvoltage protection threshold voltage, VCSOVP1 = 2.7 V, or higher
• PWM pin voltage is in the logic low state, and the CS pin voltage is the overvoltage protection threshold voltage, VCSOVP2 = 0.72 V, or higher When the OVP operates, the OUT pin voltage becomes logic low level, and the fault signal is output from the FAULT pin. When the CS pin voltage decreases to be V CSOVP1, or VCSOVP2, or lower, and the overvoltage protection restart time, t RESTART = 11.0 ms, is exceeded, the fault signal output is stopped and the LC5910S restarts (see Section 9.10 for the detail of the fault signal). Though t he leading edge blanking (LEB) circuit is incorporated in the CS pin, the RC filter should be added as needed as shown in Figure 9 -8 when large noise s are superimposed at the voltage signal between both ends of the RCS. The standard constant for RC filter is as follows: The RCSF is around 1 kΩ, and the CCSF is around 220 pF. If the RC time constant is large, the response delay becomes large, and the operation may be unstable. Therefore, be sure to confirm the actual operation and adjust the constant. GND RG2 LC5910S RCSRGS OUT DG RG1 1CS RCSF CCSF Figure 9-8. RC Filter of CS Pin

9.10 Fault Signal

Figure 9-9 shows the peripheral circuit of the FAULT pin. FAULT pin is connected to the source of the P - channel MOSFET in the IC. When the overvoltage state is detected (see Section 9.9), the internal P -channel MOSFET turns on, and the FAULT pin voltage increases to the voltage of the VCC pin. In normal opera tion, the FAULT pin is in the high impedance state. The fault signal can be used for the protection operation to suppress the power supply to the LED driver by sending signals via optocoupler to the primary- side off-line converter IC or the PFC IC. CS FAULT 2.7 V 0.72 V VCC PWMo CS_OVP 1 GND RCS VCC LC5910S Figure 9-9. FAULT Pin Peripheral Circuit

9.11 Thermal Shutdown (TSD)

When the junction temperature of the IC reaches the thermal shutdown operation temperature, T SD = 150 °C, or higher, the thermal shutd own (TSD) is activate d and stops the oscillation. The TSD has the temperature hysteresis, T SD(HYS) = 30 °C. When the temperature of the IC decreases to be TSD − TSD(HYS), or lower, the TSD automatically restarts the normal operation. The TSD protects the IC when the loss of th e IC increases due to the instantaneous short -circuit of the OUT pin . This does not guarantee the operation including the reliability in the short -circuit state for long period or the state where the heat generation continues.

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 13 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 10. Circuit Constant Setting

10.1 Inductor Setting

The on-time period depends on the input voltage, V IN, and inductance, and is determined by the point where the IC detects the peak value of inductor current, IL(PEAK). In the control of the IC, the o ff-time period is determined by the point where the BD pin detect s the VDS drop after the regenerative current flowing through the free wheeling diode, D S, becomes zero. Therefore, the off time, tOFF, is the value that the delay time, tONDLY, of the volta ge resonant part is added to the off time, tOFF_S (see Figure 10-1). tOFF_S is determined by cycle length and on-time. The design procedure of inductor is shown below. VDS IL t ILED VBD(L) tOFF tON tOFF_S tONDLY IL(PEAK) T Figure 10-1. LC5910S Operational Waveform (1) Duty D Duty D is calculated by the following equation. D = VLED VIN (2) (2) On-time, tON When the switching period is T, the on time, t ON, is as follows: tON = T × D (3) (3) Off-time, tOFF_S The off time, t OFF_S, until the regenerative current flowing through the freewheeling diode become s zero is calculated by the following equation. tOFF_S = T − tON (4) (4) Inductor The peak current of the inductor, I L(PEAK), is twice the LED average current, ILED. The value of the inductor, L, such that the peak current is I L(PEAK) is calculated by the following equation. L = VLED × tOFF_S IL(PEAK) (5) where: tOFF_S is calculated by Equation (19), and VLED is the voltage across the LED string. (5) Current Detection Resistor, RCS The RCS value is calculated by the following equation. RCS = VCS IL(PEAK) (6) (6) Delay Time, tONDLY The delay time, tONDLY, is calculated by the following equation. tONDLY = π× √L × Cds (7) (7) Off-time, tOFF The off time, t OFF, is the value that tONDLY, is added to tOFF_S calculated by (19). tOFF = tOFF_S + tONDLY (8) (8) Corrected Frequency, fSW’ The frequency , f SW’, considering the delay time, tONDLY, is calculated by the following equation. fSW′ = 1 (tON + tOFF) (9) fSW’ is later than f SW that is set as a target operating frequency by the delay time, tONDLY. In order to adjust fSW’ to the set value, the value of the inductor , L, should be decreased and the on -time should be short. Even though the on-time is adjusted, the peak i nductor current, IL(PEAK), is always twice the ILED if VIN, VCS, and RCS are fixed. As shown in Equation (7), however, when L is reduced, tONDLY also becomes short. Therefore, the waveform should be co nfirmed on actual operation and the constant of the BD pin should be adjusted so that a sufficient delay time can be secured (the power MOSFET turns on at the bottom of VDS).

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 14 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 Based on the calculated L value, it is necessary to select an inductor that does not magnetically saturate at the maximum current. The heat generation by the DCR of the winding should be below the manufacturer guaranteed value.

• Calculation Example The value of the inductor is calculated by the Equation (2) to (9) using the set value in Table 10-1. Table 10-1. Set Value of Inductance Calculation Parameter Set Value Input Voltage, VIN 160 V LED String Voltage, VLED 130 V LED Average Current, ILED 350 mA LED Current Detection Threshold Voltage, VCS 1000 mV Target Operating Frequency, fSW 100 kHz Capacitance between Gate and Drain, Cds 81 pF (Cds = Coss – Crss, for Coss and Crss, refer to the data sheet of the power MOSFET) (1) Duty, D D = 130 V 160 V = 0.8125 (2) On-time, tON By the equation, f SW = 100 kHz, the switching frequency, T, is 10 μs. tON = 10 µs × 0.8125 = 8.125 μs (3) Off-time, tOFF_S tOFF_S = 10 µs − 8.125 µs = 1.875 µs (4) Inductor By the equation, LED average current, ILED = 350 mA, the inductor peak current, IL(PEAK) is 700 mA. L = 130 V × 1.875 μs 700 mA = 348.2 µH From the calculated value, 330 µH should be chosen for the inductor value. (5) Current Detection Resistor, RCS RCS = 1000 mV 0.7 A = 1.428 Ω (6) Delay time, tONDLY tONDLY = π × 330 µH × 81 pF = 0.514 µs (7) Off-time, tOFF tOFF = 1.875 µs + 0.514 µs = 2.389 μs (8) Corrected Frequency, fSW’ fSW′ = 1 (8.125 µs + 2.389 µs) = 95.11 kHz fSW’ is later than the target operating frequency , 100 kHz, by tONDLY. In order to adjust f SW’ to the set value, the value of the inductor, L, should be decr eased and the on-time should be short. In the calculation example above, V CS2 = 1.0 V is assumed to be 100%. Table 10- 2. shows the design values when the SEL pin voltage is switched using the inductor of the L value (330 µH) set in the calculation example and the current detection resistor, R CS (the input voltage, V IN, is constant). In the application that the SEL pin voltage is changed, it is required to choose the inductor that does not saturate at the maximum current even in VCS3. Table 10-2. Calculation Result of Internal Reference Voltage and Design Value when SEL Pin Voltage is Switched Parameter VCS1 = 750 mV VCS2 = 1000 mV VCS3 = 1100 mV IL(PEAK) 525 mA 700 mA 770 mA ILED 262 mA 350 mA 385 mA tON 6.07 μs 8.13 μs 8.90 μs tOFF_S 1.40 μs 1.87 μs 2.05 μs tONDLY 0.514 μs 0.514 μs 0.514 μs T 7.98 μs 10.51 μs 11.46 μs fSW’ 125.25 kHz 95.11 kHz 87.23 kHz

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 15 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016

10.2 Input Smoothing Electrolytic

Capacitor, CIN, Setting Since t he IC p rovides the CRM control where the inductor current fluctuates from zero to the peak, ripple current becomes larger than in case of CCM control. The ripple current of the input capacitor, C IN, is calculated assuming the worst condition (all the power is supplied to the LED driver circuit fr om CIN for a period of time). In delay time, the negative current flows through CIN as shown in Figure 10-2. VDS ICINR t VBD(L) tOFF tON tOFF_S tONDLY IR(PEAK)1 IL IR(PEAK)2 T ICINR- ICINR+ IL(PEAK) Figure 10-2. C IN Ripple Current (worst condition) The current flowing in C IN during the period, t 1, in Figure 10-2 has a triangular waveform whose peak value is IR(PEAK)2. The effective value of this current , ICINR−, is calculated by Equation (10). ICINR− = IR(PEAK)2 × t1 (10) ICIN(RIPPLE) discharging from C IN during the period, t 2, in Figure 10- 2 has a triangular waveform wh ose peak value is I R(PEAK)1. The effective value of this current, ICINR+, is calculated by Equation (11). ICINR+ = IR(PEAK)1 × t2 (11) From Equation (10) and (11), the total ripple current of CIN, ICINR, is calculated by the following equation. ICINR = ICINR− 2 + ICINR+ 2 (12) The calculated value is the maximum r ipple current considering the worst condition. A capacitor that the ripple current calculated by Equation (12) can flow should be chosen. Care should be tak en in using an electrolytic capacitor.The ripple current eve ntually should be confirmed on actual operation. It is required to select input capacitor s having sufficient margin s to the maximum applied voltage according to reco mmended use conditions of capacitor manufacturers.

• Calculation Example The total ripple c urrent, ICINR, of CIN, is calculated by Equation (10), (11), and (12). If t1 = 0.7 μs, IR(PEAK)2 = 0.26 A, and the cycle , T = 10 μs (the frequency fSW = 100 kHz), ICINR−, is calculated as follows: ICINR− = 0.26 A × 0.7 µs 3 × 10 µs = 0.039 Arms When t2 = 6.4 μs, IR(PEAK)1 = IL(PEAK) = 0.7 A, and the cycle, T = 10 μs (frequency, fSW = 100 kHz), ICINR+ is calculated as follows: ICINR+ = 0.7 A × 6.4 µs 3 × 10 µs = 0.323 Arms From the above, the total ripple current, I CINR, is as follows: ICINR = 0.0392 + 0.3232 = 0.325 Arms From the above, a capacitor that can permit the ripple current of 0.325 Arms or higher must be selected for CIN.

10.3 Output Smoothing Capacitor, COUT,

Since t he IC p rovides the CRM control where the inductor current fluctuates from zero to the peak , it is required to connect the output smoothing capacitor , COUT, to the both ends of the LED string , and reduce the ripple voltage of the LED string. The ripple current, I COUTR, flowing in C OUT, is calculated by Equation (13). ICOUTR = IL(PEAK) 2√3 (13) When the equivalent series resistor of the smoothing capacitor, COUT, is ESR, the ripple voltage, V LED(RIPPLE),

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 16 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 of COUT , is expressed by Equation (14). VLED(RIPPLE) = IL(PEAK) × ESR (14) A capacitor that the ripple current calculated by Equation (13) can flow is required for C OUT. The voltage across COUT is the sum of V F in the LED string. It is required to select the breakdo wn voltage of C OUT having margins to the maximum applied voltage considering the ripple voltage in Equation (14) according to recommended use conditions of capacitor manufacturers. VDS ICOUTR t VBD(L) tOFF tON tOFF_S tONDLY IL(PEAK) IL T IL(PEAK) Figure 10-3. C OUT Ripple Current

• Calculation Example If IL(PEAK) = 0.7 A , the ripple current, I COUTR, flowing in COUT is calculated by Equation (13) as follows: ICOUTR = 0.7 A 2√3 = 0.202 Arms From the above, a capacitor that can permit the ripple current of 0.202 Arms or higher should be selected for COUT. When ESR = 100 mΩ and IL(PEAK) = 0.7 A , the ripple voltage, VLED(RIPPLE) , of C OUT is calculated by Equation (14) as follows: VLED(RIPPLE) = 0.7 A × 100 mΩ = 70 mVp– p It is required to select the withstand voltage of C OUT having margins to the maximum applied voltage considering the ripple voltage of 70 mVp- p according to recommended use conditions of capacitor manufacturers.

10.4 Current Detection Resistor, RCS,

For the current detection resistor, R CS, noninductive resistors such as metal plate resistors , metal film resistors, and carbon film re sistors should be used. If inductive resistors such as winding wire type are used, malfunction may occur due to the surge voltage by parasitic inductance component. Axial lead and radial lead re sistor should be mounted with the lead as short as possible.

• R CS Loss Calculation The switching current flows in R CS in the on -time period of the power MOSFET. T he current flowing in RCS, IRCS, is calculated by the following equation. IRCS = ILED × D (15) where, ILED is LED current, and D is duty The average loss of RCS, PRCS, in Equation ( 15), is calculated by the following equation. PRCS = (IRCS)2 × RCS (16) When VCS is switched by the SEL pin input voltage , the power dissipation should be calculated with the maximum value, VCS3 = 1.1 V. In an abnormal operation state such as a short between the drain and source pin s of the external MOSFET , the detection resistor, R CS, may be damaged. In order to avoid damage, the following measures should be taken. - Increase the RCS power rating. - Insert the protective fuse that melts before RCS is damaged into the power supply of the LED driver circuit. - Using the FAULT pin of the LC5910S, stop the pre- converter that generates the supply power of the LED driver circuit.

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 17 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016

• Calculation Example Table 10-3. Set Value of Current Detection Resistance Calculation Parameter Set Value Detection Resistor, RCS 1.428 Ω Reference Voltage, VCS 1.0 V (VSEL = 0.75 V to 1.25 V) LED Current, ILED 0.35 A (IL(PEAK) = 0.7 A) LED String Voltage, VLED 130 V Input Voltage, VIN 160 V Duty, D 0.8125 Using the set value in Table 10 -3., t he loss of resistance is calculated by Equation (15) and Equation (16). IRCS = 0.35 A × 0.8125 = 0.284 A PRCS = (0.284 A)2 × 1.428 Ω = 0.115 W 11. Design Notes

11.1 External Components

Components fit for the use condition should be used.

11.1.1 Inductor

It is required to set design margin properly for temperature rise due to copper loss and iron loss , and for magnetic saturation.

11.1.2 Input and Output Smoothing

It is required to set a design margin properly for the ripple current, voltage, and lifetime. The electrolytic capacitor used must have high allowable ripple current for switching power supplies and low impedance.

11.1.3 Current Detection Resistor

Because high -frequency switching current flows through the current detection resistor , a resistor with low internal inductance must be chosen. In addition, the resistor is required to be selected considering the allowable loss.

11.1.4 Freewheeling Diode

The freewheeling diode, D S, that regenerates energy during the off -time period of the external power MOSFET (see Figure 6 -1) is selected from fast recovery diodes with short recovery time, t rr, or from Schottky diodes. The peak of the forward current , I F, of the freewheeling diode is twice the ILED, and flows in the tOFF period of the power MOSFET. IF flows repeatedly at the same switching frequency as the power MOSFET . Therefore, a diode should be selec ted considering the allowable loss. Do not use a general rectification diode used for commercial power supply rectification because its t rr is long and a large short -circuit cur rent f lows during the recovery period. This short-circuit current may cause not only the heat generation of the diode but also malfunction of the main circuit, resulting in damage.

11.1.5 External Power MOSFET

• Breakdown Voltage between Drain and Source The input voltage , V IN－VF (the forward voltage of the freewheeling diode) is applied between the drain and source of the power MOSFET when it is off. Considering surges at turn-off, the power MOSFET whose drain- source breakdown voltage is more than twice VIN should be selected.
• Capacitance between Drain and Source (Cds) The capacitance between drain and source (Cds) affects the delay time, t ONDLY. tONDLY becomes longer as the capacitance becomes larger (see Section 10.1). Although Cds is not described in the data sheet of the power MOSFET, it can be calculated from the output capacitance, Coss, and the feedback capacitance, Crss, as follows: Coss = Cds −Cgd (17) Crss = Cgd (18) where, Cgd is the capacitance between gate and drain, From Equation (17) and (18), Cds is calculated by the following equation. Cds = Coss −Crss (19)

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 18 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016

• Breakdown Voltage between Gate and Source The gate drive voltage of the LC5910S varies in proportion to the VCC pin voltage. Therefore, in the specification that VCC voltage rises to the upp er limit of

17 V of the recommended operating range, a power

MOSFET with gate -source breakdown voltage of 20 V to 30 V should be selected. When a stabilized voltage of 12 V is input to the VCC pin, the crest value in the output pulse of the OUT pin is about 12 V.

• Others In general, the larger the package and the lower the on-resistance of a power MOSFET, the larger the capacitance (e.g. Ciss) between junctions. Therefore, the larger drive current is required. Considering the drive capability of the LC5910S, a power MOSFET with package size smaller than TO 220 is recommend.

11.2 PCB Trace Layout

The switching power supply circuit includes high frequency and high voltage current paths that affect the IC operation, noise interference, and power dissipation. Therefore, PCB trace layouts and component placements play an important role in circuit designing. High frequency and high voltage current loops must be as small as possible with wide trace, in order to maintain a low-impedance state (see Figure 11-1). In addition, ground traces should be as wide and short as possible so that radiated EMI levels can be reduced. When the input/output conditions exceed the Safety Extra Low Voltage (SELV), component layout and pattern distance considering safety standards are required. Figure 11-1. High Frequency Loop (shaded area ) (1) Main Circuit Trace Layout The high frequency loop as shown in Figure 11-1 and the loop including the VCC pin and the GND pin are the main trace flowing switching currents . This traces must be as wide layout and small loop as possible. (2) Around the GND Pin In order to prevent the switching current of the main circuit from inf luencing the control circuit, the control ground must be separated form the main trace and be connected near the GND pin. (3) R CS Trace Layout RCS should be placed as close as possible to the CS pin. In order to reduce noise at current detection, the detection trace must be separated, and be connected to near RCS from the CS pin. (4) Peripheral Components of the IC The components connected to each pin must be placed as close as possible to the IC, and must be connected as short as possible. The ground of components for detection of the IC must be connected near the GND pin. Extra attention should be paid so that the IC ground does not become a common impedance with the main trace. Do not place the IC or small signal pattern just under the inductor, L. The IC may mal function due to the leak magnetic flux. (5) Output Smoothing Capacitor, C OUT The output smoothing capacitor, COUT is placed close to the LED string with a minimum length of traces. (6) Thermal Considerations Because the power MOSFET has a positive thermal coefficient of RDS(ON), care should be taken for thermal design. Particularly in the case of surface mount components, the drain pin pattern functions as a heat sink. Therefore, the pattern must be designed as wide as possible. CIN DS COUT

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 20 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 13. Reference Design of Power Supply As an example, the following s show the power supply specification and the bill of materials. For the circuit diagram of the bill of materials, see Figure 12-2.

• Power Supply Specification Input Voltage DC250 V VCC Pin Voltage 13 V Output Voltage DC200 V LED Current 0.478 A (VSEL = 1.0V)
• Bill of Material Symbol Part Type Rating Notes L01* Inductor 0.47 mH C1 Ceramic 1 nF C2 Ceramic 100 pF C3 Ceramic Open C4 Ceramic 15 pF, 1 kV C5 Ceramic 0.47 μF C6* Ceramic 100 pF, 50 V C7 Ceramic Open C8 Ceramic 0.47 μF, 400 V C9 Electrolytic 22 μF, 25 V C10 Electrolytic 10 μF, 250 V D01 Fast Recovery Diode 600 V, 2 A D02 Diode 60 V, 0.5 A Q1 N-channel Power MOSFET 250 V, 7.5A, 0.4Ω Q2 PNP Transistor −60 V, −0.6 A Q3 NPN Transistor 40 V, 0.6 A R0 General 1.8 kΩ R1 General Open R2 General 2.2 kΩ R3 General 33 kΩ R4 General 30 kΩ R5 General Open R6 General 470 kΩ R7 General 470 kΩ R8 General 1 kΩ R9 General 510 kΩ R10 General 10 Ω R11* General 7.5 Ω R12* General 47 Ω R13* General 10 kΩ R14* General 100 Ω R15 General 1 Ω, 2 W R16 General 2.3 Ω, 2 W R17 General Open R18 General Open R19 General 4.7 kΩ LED — — LED for Fault detection * Adjustment is required on actual operation.

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 21 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 14. Design Flow Chart The flow chart below is specialized for selection of constants. Measures against noise and heat dissipation on actual operation are not included. Therefore, another adjustment is required. START VLED = VF × n VIN ILED Condition Setting Target Frequency Setting Calculation of Duty, D tON、tOFF_S Calculation tON ≤ 20μs Inductance Calculation Power MOSFET Selection Delay Time, tONDLY Calculation tOFF Calculation Corrected Frequency Calculation VSEL Setting RCS Setting CIN Setting COUT Setting Peripheral Components Selection such as Free Wheeling Diode Actual Operation Confirmation RBD, CBD Adjustment Check the bottom-on of the BD pin wave -form END

LC5910S -DSE Rev.2.2 SANKEN ELECTRIC CO., LTD 22 Mar.01, 2017 http://www.sanken-ele.co.jp © SANKEN ELECTRIC CO., LTD. 2016 Important Notes

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