L163B

LM2734Z NSC | Alldatasheet

Document overview

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Technical content

Features

n Thin SOT23-6 package, or 6 lead LLP package n 3.0V to 20V input voltage range n 0.8V to 18V output voltage range n 1A output current n 3MHz switching frequency n 300mΩNMOS switch n 30nA shutdown current n 0.8V, 2% internal voltage reference n Internal soft-start n Current-Mode, PWM operation n Thermal shutdown

Applications

n Local Point of Load Regulation n Battery Powered Devices n USB Powered Devices Typical Application Circuit Efficiency vs Load Current VIN = 5V, VOUT = 3.3V 20130301 20130345 WEBENCH™is a trademark of Transim. March 2005 LM2734Z Thin SOT23 1A Load Step-Down DC-DC Regulator © 2005 National Semiconductor Corporation DS201303 www.national.com

6-Lead LLP (3mm x 3mm)

Ordering Information

1000 Units on Tape and Reel

3000 Units on Tape and Reel

4500 Units on Tape and Reel

• Contact the local sales office for the lead-free package. Pin Description Pin Name Function BOOST Boost voltage that drives the internal NMOS control switch. A bootstrap capacitor is connected between the BOOST and SW pins. GND Signal and Power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin for accurate regulation. FB Feedback pin. Connect FB to the external resistor divider to set output voltage. EN Enable control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V. VIN Input supply voltage. Connect a bypass capacitor to this pin. SW Output switch. Connects to the inductor, catch diode, and bootstrap capacitor. DAP GND The Die Attach Pad is internally connected to GND LM2734Z www.national.com

Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN -0.5V to 24V SW Voltage -0.5V to 24V Boost Voltage -0.5V to 30V Boost to SW Voltage -0.5V to 6.0V FB Voltage -0.5V to 3.0V EN Voltage -0.5V to (VIN + 0.3V) Junction Temperature 150˚C ESD Susceptibility (Note 2) 2kV Storage Temp. Range -65˚C to 150˚C Soldering Information Infrared/Convection Reflow (15sec) 220˚C Wave Soldering Lead Temp. (10sec) 260˚C Operating Ratings (Note 1) VIN 3V to 20V SW Voltage -0.5V to 20V Boost Voltage -0.5V to 25V Boost to SW Voltage 1.6V to 5.5V Junction Temperature Range −40˚C to +125˚C Thermal Resistance θJA (Note 3) TSOT23–6 118˚C/W

Electrical Characteristics

Specifications with standard typeface are for TJ = 25˚C, and those in boldface type apply over the full Operating Tempera- ture Range (TJ = -40˚C to 125˚C). VIN = 5V, VBOOST - VSW = 5V unless otherwise specified. Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. Symbol Parameter Conditions Min (Note 4) Typ (Note 5) Max (Note 4) Units VFB Feedback Voltage 0.784 0.800 0.816 V ∆VFB/∆VIN Feedback Voltage Line Regulation VIN = 3V to 20V 0.01 % / V IFB Feedback Input Bias Current Sink/Source 250 nA UVLO Undervoltage Lockout VIN Rising 2.74 2.90 V Undervoltage Lockout VIN Falling 2.0 2.3 UVLO Hysteresis 0.30 0.44 0.62 FSW Switching Frequency 2.2 3.0 3.6 MHz DMAX Maximum Duty Cycle DMIN Minimum Duty Cycle RDS(ON) Switch ON Resistance VBOOST - VSW = 3V (TSOT Package) 300 600 mΩ VBOOST - VSW = 3V (LLP Package) 340 650 mΩ ICL Switch Current Limit VBOOST - VSW = 3V 1.2 1.7 2.5 A IQ Quiescent Current Switching 1.5 2.5 mA Quiescent Current (shutdown) VEN = 0V nA IBOOST Boost Pin Current (Switching) 4.25 mA VEN_TH Shutdown Threshold Voltage VEN Falling 0.4 V Enable Threshold Voltage VEN Rising 1.8 IEN Enable Pin Current Sink/Source nA ISW Switch Leakage nA Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics. Note 2: Human body model, 1.5kΩin series with 100pF. Note 3: Thermal shutdown will occur if the junction temperature exceeds 165˚C. The maximum power dissipation is a function of TJ(MAX) , θJA and TA . The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) – TA)/θJA . All numbers apply for packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still air, θJA = 204˚C/W. Note 4: Guaranteed to National’s Average Outgoing Quality Level (AOQL). Note 5: Typicals represent the most likely parametric norm. LM2734Z www.national.com

Typical Performance Characteristics All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 2.2 µH and TA = 25˚C, unless specified otherwise. Efficiency vs Load Current VOUT = 5V Efficiency vs Load Current VOUT = 3.3V 20130336 20130351 Efficiency vs Load Current VOUT = 1.5V Oscillator Frequency vs Temperature 20130337 20130327 Line Regulation VOUT = 1.5V, IOUT = 500mA Line Regulation VOUT = 3.3V, IOUT = 500mA 20130354 20130355 LM2734Z www.national.com

Application Information

(Continued) ENABLE PIN / SHUTDOWN MODE The LM2734Z has a shutdown mode that is controlled by the enable pin (EN). When a logic low voltage is applied to EN, the part is in shutdown mode and its quiescent current drops to typically 30nA. Switch leakage adds another 40nA from the input supply. The voltage at this pin should never exceed VIN + 0.3V. SOFT-START This function forces VOUT to increase at a controlled rate during start up. During soft-start, the error amplifier’s refer- ence voltage ramps from 0V to its nominal value of 0.8V in approximately 200µs. This forces the regulator output to ramp up in a more linear and controlled fashion, which helps reduce inrush current. OUTPUT OVERVOLTAGE PROTECTION The overvoltage comparator compares the FB pin voltage to a voltage that is 10% higher than the internal reference Vref. Once the FB pin voltage goes 10% above the internal refer- ence, the internal NMOS control switch is turned off, which allows the output voltage to decrease toward regulation. UNDERVOLTAGE LOCKOUT Undervoltage lockout (UVLO) prevents the LM2734Z from operating until the input voltage exceeds 2.74V(typ). The UVLO threshold has approximately 440mV of hyster- esis, so the part will operate until VIN drops below 2.3V(typ). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic. CURRENT LIMIT The LM2734Z uses cycle-by-cycle current limiting to protect the output switch. During each switching cycle, a current limit comparator detects if the output switch current exceeds 1.7A (typ), and turns off the switch until the next switching cycle begins. THERMAL SHUTDOWN Thermal shutdown limits total power dissipation by turning off the output switch when the IC junction temperature ex- ceeds 165˚C. After thermal shutdown occurs, the output switch doesn’t turn on until the junction temperature drops to approximately 150˚C. Design Guide INDUCTOR SELECTION The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VO) to input voltage (VIN): The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to calculate a more accurate duty cycle. Calculate D by using the follow- ing formula: VSW can be approximated by: VSW = IO x RDS(ON) The diode forward drop (VD) can range from 0.3V to 0.7V depending on the quality of the diode. The lower VD is, the higher the operating efficiency of the converter. The inductor value determines the output ripple current. Lower inductor values decrease the size of the inductor, but increase the output ripple current. An increase in the inductor value will decrease the output ripple current. The ratio of ripple current (∆iL) to output current (IO) is optimized when it is set between 0.3 and 0.4 at 1A. The ratio r is defined as: One must also ensure that the minimum current limit (1.2A) is not exceeded, so the peak current in the inductor must be calculated. The peak current (ILPK) in the inductor is calcu- lated by: ILPK = IO + ∆IL/2 If r = 0.5 at an output of 1A, the peak current in the inductor will be 1.25A. The minimum guaranteed current limit over all operating conditions is 1.2A. One can either reduce r to 0.4 resulting in a 1.2A peak current, or make the engineering judgement that 50mA over will be safe enough with a 1.7A typical current limit and 6 sigma limits. When the designed maximum output current is reduced, the ratio r can be in- creased. At a current of 0.1A, r can be made as high as 0.9. The ripple ratio can be increased at lighter loads because the net ripple is actually quite low, and if r remains constant the inductor value can be made quite large. An equation empirically developed for the maximum ripple ratio at any current below 2A is: r = 0.387 x IOUT -0.3667 Note that this is just a guideline. 20130348 FIGURE 5. Boost Voltage Supplied from the Shunt

(Continued) The LM2734Z operates at frequencies allowing the use of ceramic output capacitors without compromising transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple. See the output capacitor section for more details on calculating output volt- age ripple. Now that the ripple current or ripple ratio is determined, the inductance is calculated by: where fs is the switching frequency and IO is the output current. When selecting an inductor, make sure that it is capable of supporting the peak output current without satu- rating. Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating cor- rectly. Because of the speed of the internal current limit, the peak current of the inductor need only be specified for the required maximum output current. For example, if the de- signed maximum output current is 0.5A and the peak current is 0.7A, then the inductor should be specified with a satura- tion current limit of >0.7A. There is no need to specify the saturation or peak current of the inductor at the 1.7A typical switch current limit. The difference in inductor size is a factor of 5. Because of the operating frequency of the LM2734Z, ferrite based inductors are preferred to minimize core losses. This presents little restriction since the variety of ferrite based inductors is huge. Lastly, inductors with lower series resistance (DCR) will provide better operating efficiency. For recommended inductors see Example Circuits. INPUT CAPACITOR An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The primary specifications of the input capacitor are capacitance, volt- age, RMS current rating, and ESL (Equivalent Series Induc- tance). The recommended input capacitance is 10µF, al- though 4.7µF works well for input voltages below 6V. The input voltage rating is specifically stated by the capacitor manufacturer. Make sure to check any recommended derat- ings and also verify if there is any significant change in capacitance at the operating input voltage and the operating temperature. The input capacitor maximum RMS input cur- rent rating (IRMS-IN) must be greater than: It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always calcu- late the RMS at the point where the duty cycle, D, is closest to 0.5. The ESL of an input capacitor is usually determined by the effective cross sectional area of the current path. A large leaded capacitor will have high ESL and a 0805 ce- ramic chip capacitor will have very low ESL. At the operating frequencies of the LM2734Z, certain capacitors may have an ESL so large that the resulting impedance (2πfL) will be higher than that required to provide stable operation. As a result, surface mount capacitors are strongly recommended. Sanyo POSCAP, Tantalum or Niobium, Panasonic SP or Cornell Dubilier ESR, and multilayer ceramic capacitors (MLCC) are all good choices for both input and output ca- pacitors and have very low ESL. For MLCCs it is recom- mended to use X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating conditions. OUTPUT CAPACITOR The output capacitor is selected based upon the desired output ripple and transient response. The initial current of a load transient is provided mainly by the output capacitor. The output ripple of the converter is: When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the out- put ripple will be approximately sinusoidal and 90˚ phase shifted from the switching action. Given the availability and quality of MLCCs and the expected output voltage of designs using the LM2734Z, there is really no need to review any other capacitor technologies. Another benefit of ceramic ca- pacitors is their ability to bypass high frequency noise. A certain amount of switching edge noise will couple through parasitic capacitances in the inductor to the output. A ce- ramic capacitor will bypass this noise while a tantalum will not. Since the output capacitor is one of the two external components that control the stability of the regulator control loop, most applications will require a minimum at 10 µF of output capacitance. Capacitance can be increased signifi- cantly with little detriment to the regulator stability. Like the input capacitor, recommended multilayer ceramic capacitors are X7R or X5R. Again, verify actual capacitance at the desired operating voltage and temperature. Check the RMS current rating of the capacitor. The RMS current rating of the capacitor chosen must also meet the following condition: CATCH DIODE The catch diode (D1) conducts during the switch off-time. A Schottky diode is recommended for its fast switching times and low forward voltage drop. The catch diode should be chosen so that its current rating is greater than: ID1 = IO x (1-D) The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin. To im- prove efficiency choose a Schottky diode with a low forward voltage drop. BOOST DIODE A standard diode such as the 1N4148 type is recommended. For VBOOST circuits derived from voltages less than 3.3V, a small-signal Schottky diode is recommended for greater ef- ficiency. A good choice is the BAT54 small signal diode. BOOST CAPACITOR A ceramic 0.01µF capacitor with a voltage rating of at least 6.3V is sufficient. The X7R and X5R MLCCs provide the best performance. LM2734Z www.national.com

(Continued) OUTPUT VOLTAGE The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and R1 is connected between VO and the FB pin. A good value for R2 is 10kΩ. PCB Layout Considerations When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The most important consideration when completing the layout is the close coupling of the GND connections of the CIN capacitor and the catch diode D1. These ground ends should be close to one another and be connected to the GND plane with at least two through-holes. Place these components as close to the IC as possible. Next in importance is the location of the GND connection of the COUT capacitor, which should be near the GND connections of CIN and D1. There should be a continuous ground plane on the bottom layer of a two-layer board except under the switching node island. The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup and inaccurate regulation. The feedback resistors should be placed as close as possible to the IC, with the GND of R2 placed as close as possible to the GND of the IC. The VOUT trace to R1 should be routed away from the inductor and any other traces that are switching. High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible. However, making the traces wide increases radiated noise, so the designer must make this trade-off. Radiated noise can be decreased by choosing a shielded inductor. The remaining components should also be placed as close as possible to the IC. Please see Application Note AN-1229 for further considerations and the LM2734Z demo board as an example of a four-layer layout. Calculating Efficiency, and Junction Temperature: The complete LM2734Z DC/DC converter efficiency can be calculated in the following manner. Or Calculations for determining the most significant power losses are shown below. Other losses totaling less than 2% are not discussed. Power loss (PLOSS) is the sum of two basic types of losses in the converter, switching and conduction. Conduction losses usually dominate at higher output loads, where as switching losses remain relatively fixed and dominate at lower output loads. The first step in determining the losses is to calculate the duty cycle (D). VSW is the voltage drop across the internal NFET when it is on, and is equal to: VSW = IOUT x RDSON VD is the forward voltage drop across the Schottky diode. It can be obtained from the Electrical Characteristics section. If the voltage drop across the inductor (VDCR) is accounted for, the equation becomes: This usually gives only a minor duty cycle change, and has been omitted in the examples for simplicity. The conduction losses in the free-wheeling Schottky diode are calculated as follows: PDIODE = VD x IOUT(1-D) Often this is the single most significant power loss in the circuit. Care should be taken to choose a Schottky diode that has a low forward voltage drop. Another significant external power loss is the conduction loss in the output inductor. The equation can be simplified to: PIND = IOUT 2 x RDCR The LM2734Z conduction loss is mainly associated with the internal NFET: PCOND = IOUT 2 x RDSON x D Switching losses are also associated with the internal NFET. They occur during the switch on and off transition periods, where voltages and currents overlap resulting in power loss. The simplest means to determine this loss is to empirically measuring the rise and fall times (10% to 90%) of the switch at the switch node: PSWF = 1/2(VIN x IOUT x freq x TFALL) PSWR = 1/2(VIN x IOUT x freq x TRISE) PSW = PSWF + PSWR Typical Rise and Fall Times vs Input Voltage VIN TRISE TFALL 8ns 4ns 10V 9ns 6ns 15V 10ns 7ns Another loss is the power required for operation of the inter- nal circuitry: PQ = IQ x VIN IQ is the quiescent operating current, and is typically around 1.5mA. The other operating power that needs to be calcu- lated is that required to drive the internal NFET: PBOOST = IBOOST x VBOOST LM2734Z www.national.com

removed through conduction and/or convection. ductivity properties (insulator vs conductor). silicon→package→lead frame→PCB. also make a large difference in the thermal impedance. duct heat from the surface of the PCB to the ground plane. vias close to the ground pin of the device. ence in thermal impedance for a four-layer board with 2oz. 235˚C/W for 1oz. copper traces and GND plane. FIGURE 6. Cross-Sectional View of Integrated Circuit Mounted on a Printed Circuit Board.

approximately 150˚C, the device will start to switch again. ture and the ambient temperature, RθJA can be determined. allowed for a desired junction temperature can be found. 2.5cm x 3cm. It was placed in an oven with no forced airflow. temperature, the device went into thermal shutdown. the ambient temperature cannot go above 54.2˚C. same manner as described in method #2 (see example 3). posed ground paddle on the bottom of the package. configuration (area, copper weight, thermal vias). FIGURE 7. Dog Bone

(Continued) For certain high power applications, the PCB land may be modified to a "dog bone" shape (see Figure 7). By increasing the size of ground plane, and adding thermal vias, the RθJA for the application can be reduced. Design Example 4: Operating Conditions Package LLP-6 VIN 12.0V POUT 2.475W VOUT 3.3V PDIODE 523mW IOUT 750mA PIND 56.25mW VD 0.35V PSWF 108mW Freq 3MHz PSWR 108mW IQ 1.5mA PCOND 68.2mW IBOOST 4mA PQ 18mW VBOOST PBOOST 20mW TRISE 8ns PLOSS 902mW TFALL 8ns RDSON 400mΩ INDDCR 75mΩ D 30.3% This example follows example 2, but uses the LLP package. Using a standard National Semiconductor LLP-6 demonstra- tion board, use Method 2 to determine RθJA of the board. The four layer PCB is constructed using FR4 with 1/2oz copper traces. The copper ground plane is on the bottom layer. The ground plane is accessed by four vias. The board measures 2.5cm x 3cm. It was placed in an oven with no forced airflow. The ambient temperature was raised to 113˚C, and at that temperature, the device went into thermal shutdown. If the junction temperature is to be kept below 125˚C, then the ambient temperature cannot go above 73.2˚C. TJ - (RθJA x PLOSS) = TA Package Selection To determine which package you should use for your specific application, variables need to be known before you can determine the appropriate package to use. Maximum ambient system temperature Internal LM2734Z power losses Maximum junction temperature desired RθJA of the specific application, or RθJC (LLP or Thin SOT23-6) The junction temperature must be less than 125˚C for the worst-case scenario. LM2734Z www.national.com

FIGURE 8. VBOOST Derived from VIN

FIGURE 9. VBOOST Derived from VOUT

FIGURE 10. VBOOST Derived from VSHUNT

FIGURE 11. VBOOST Derived from Series Zener Diode (VIN)

FIGURE 12. VBOOST Derived from Series Zener Diode (VOUT)

inches (millimeters) unless otherwise noted 6-Lead SOT23 Package 6-Lead LLP Package LM2734Z www.national.com

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. For the most current product information visit us at www.national.com. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. BANNED SUBSTANCE COMPLIANCE National Semiconductor manufactures products and uses packing materials that meet the provisions of the Customer Products Stewardship Specification (CSP-9-111C2) and the Banned Substances and Materials of Interest Specification (CSP-9-111S2) and contain no ‘‘Banned Substances’’ as defined in CSP-9-111S2. National Semiconductor Americas Customer Support Center Email: new.feedback@nsc.com Tel: 1-800-272-9959 National Semiconductor Europe Customer Support Center Fax: +49 (0) 180-530 85 86 Email: europe.support@nsc.com Deutsch Tel: +49 (0) 69 9508 6208 English Tel: +44 (0) 870 24 0 2171 Français Tel: +33 (0) 1 41 91 8790 National Semiconductor Asia Pacific Customer Support Center Email: ap.support@nsc.com National Semiconductor Japan Customer Support Center Fax: 81-3-5639-7507 Email: jpn.feedback@nsc.com Tel: 81-3-5639-7560 www.national.com LM2734Z Thin SOT23 1A Load Step-Down DC-DC Regulator