FAN2356AMPX

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© 2015 Fairchild Semiconductor Corporation www.fairchildsemi.com FAN2356A • Rev. 1.0 FAN2356AMPX —6 A Synchronous Buck Regulator FAN2356AMPX

6 A Synchronous Buck Regulator

Features

 VIN Range: 4.5 V to 24 V  High Efficiency: Over to 96% Peak  Continuous Output Current: 6 A  PFM Mode for Light-Load Efficiency  Excellent Line and Load Transient Response  Precision Reference: ±1% Over Temperature  Output Voltage Range: 0.6 to 5.5 V  Programmable Frequency: 200 kHz to 1.5 MHz  Programmable Soft-Start  Low Shutdown Current  Adjustable Sourcing Current Limit  Internal Boot Diode  Thermal Shutdown  Halogen and Lead Free, RoHS Compliant

Applications

 Mainstream Notebooks  Servers and Desktop Computers  Game Consoles  Telecommunications  Storage  Base Stations

Description

The FAN2356A is a highly efficient synchronous buck regulator. The regulator is capable of operating with an input range from 4.5 V to 24 V and supporting up to 6 A continuous load currents. The FAN2356 utilizes Fairchild’s constant on -time control architecture to provide excellent transient response and to maintain a relatively constant switching frequency. This device utilizes Pulse Frequency Modulation (PFM) mode to maximize light -load efficiency by reducing switching frequency when the inductor is operating in discontinuous conduction mode at light loads, while clamping the minimum frequency above the audible range with ultrasonic mode. Switching frequency and over -current protection can be programmed to provide a flexible solution for various applications. Output over -voltage, under - voltage, over-current, and thermal shutdown protections help prevent damage to the device during fau lt conditions. After thermal shutdown is activated, a hysteresis feature restarts the device when normal operating temperature is reached.

Ordering Information

Part Number Configuration Operating Temperature Range Output Current (A) Package FAN2356AMPX PFM with Ultrasonic Mode -40 to 125°C 6 34-Lead, PQFN, 5.5 mm x 5.0 mm

Figure 3. Pin Assignments(Bottom View) Figure 4. Pin Assignments(Top View) PVIN P2, 5-11 Power input for the power stage. VIN 1 Input to the modulator for input voltage feed-forward. PVCC 25 Power input for the low-side gate driver and boot diode. VCC 26 Power supply input for the controller. PGND 18-21 Power ground for the low-side power MOSFET and for the low-side gate driver. AGND P1, 4, 23 Analog ground for the analog portions of the IC and for substrate. SW P3, 2, 12-17, 22 Switching node; junction between high-and low-side MOSFETs. ILIM 24 Current limit. A resistor between ILIM and SW sets the current limit threshold. FB 27 Output voltage feedback to the modulator. EN 29 Enable input to the IC. Pin must be driven logic high to enable, or logic low to disable. SS 31 Soft-start input to the modulator. AGND to program on-time and switching frequency. PGOOD 30 Power good; open-drain output indicating VOUT is within set limits. NC 28, 33-34 Leave pin open or connect to AGND.

© 2015 Fairchild Semiconductor Corporation www.fairchildsemi.com FAN2356A • Rev. 1.0 4 FAN2356AMPX — 6 A Synchronous Buck Regulator Absolute Maximum Ratings Stresses exceeding the absolute maximum ratings may damage the device. The device may not function or be operable above the recommended operating conditions and stressing the parts to these levels is not recommended. In addition, extended exposure to stresses above the recommended operating conditions may affect device reliability. The absolute maximum ratings are stress ratings only. Symbol Parameter Conditions Min. Max. Unit VPVIN Power Input Referenced to PGND -0.3 30.0 V VIN Modulator Input Referenced to AGND -0.3 30.0 V VBOOT Boot Voltage Referenced to PVCC -0.3 30.0 V Referenced to PVCC, <20 ns -0.3 33.0 V VSW SW Voltage to GND Referenced to PGND, AGND -1 30 V Referenced to PGND, AGND < 20 ns -5 30 V VBOOT Boot to SW Voltage Referenced to SW -0.3 6.0 V Boot to PGND Referenced to PGND -0.3 30 V VPVCC Gate Drive Supply Input Referenced to PGND, AGND -0.3 6.0 V VVCC Controller Supply Input Referenced to PGND, AGND -0.3 6.0 V VILIM Current Limit Input Referenced to AGND -0.3 6.0 V VFB Output Voltage Feedback Referenced to AGND -0.3 6.0 V VEN Enable Input Referenced to AGND -0.3 6.0 V VSS Soft Start Input Referenced to AGND -0.3 6.0 V VFREQ Frequency Input Referenced to AGND -0.3 6.0 V VPGOOD Power Good Output Referenced to AGND -0.3 6.0 V ESD Electrostatic Discharge Human Body Model, JESD22-A114 1000 V Charged Device Model, JESD22-C101 2500 V TJ Junction Temperature +150 °C TSTG Storage Temperature -55 +150 °C Recommended Operating Conditions The Recommended Operating Conditions table defines the conditions for actual device operation. Recommended operating conditions are specified to ensure optimal performance to the datasheet specifications. Fairchild does not recommend exceeding them or designing to Absolute Maximum Ratings. Symbol Parameter Conditions Typ. Max. Unit VPVIN Power Input Referenced to PGND 4.5 24 V VIN Modulator Input Referenced to AGND 4.5 24 V TJ Junction Temperature -40 +125 °C ILOAD Load Current TA=25°C, No Airflow 9 A VPVCC Gate Drive Supply Input Referenced to PGND, AGND 4.5 5.5 V

© 2015 Fairchild Semiconductor Corporation www.fairchildsemi.com FAN2356A • Rev. 1.0 5 FAN2356AMPX — 6 A Synchronous Buck Regulator Thermal Characteristics The thermal characteristics were evaluated on a 4-layer pcb structure (1 oz/1 oz/1 oz/1 oz) measuring 7 cm x 7 cm). Symbol Parameter Typ. Unit JA Thermal Resistance, Junction-to-Ambient 35 °C/W ψJC Thermal Characterization Parameter, Junction-to-Top of Case 2.7 °C/W ψJPCB Thermal Characterization Parameter, Junction-to-PCB 2.3 °C/W

Electrical Characteristics

Unless otherwise noted; VIN=12 V, VOUT=1.2 V, and TA=TJ = -40 to +125°C. (1) Symbol Parameter Condition Min. Typ. Max. Unit Supply Current IVCC,SD Shutdown Current EN=0 V 10 µA IVCC,Q Quiescent Current EN=5 V, Not Switching 1.8 mA IVCC,GateCharge Gate Charge Current EN=5 V, fSW=500 kHz 10 mA Reference, Feedback Comparator VFB FB Voltage Trip Point 590 596 602 mV IFB FB Pin Bias Current -100 0 100 nA Modulator tON On-Time Accuracy RFREQ=56.2 k, VIN=10 V, tON=250 ns, No Load -20 20 % tON,PFM PFM On-Time Multiplier 150 % tOFF,MIN Minimum SW Off-Time 320 374 ns tON,MIN Minimum SW On-Time 45 ns DMIN Minimum Duty Cycle FB=1 V 0 % fMINF Minimum Frequency Clamp 18.2 25.4 32.7 kHz Soft-Start ISS Soft-Start Current SS=0.5 V 7 10 13 µA tON,SSMOD SS On-Time Modulation SS<0.6 V 25 100 % VSSCLAMP,NOM Nominal Soft-Start Voltage Clamp VFB=0.6 V 400 mV VSSCLAMP,OVL Soft-Start Voltage Clamp in Overload Condition VFB=0.3 V, OC Condition 40 mV PFM Zero-Crossing Detection Comparator VOFF ZCD Offset Voltage TA=TJ=25°C -6 0 mV Current Limit ILIM Valley Current Limit Accuracy TA=TJ=25°C, IVALLEY=7 A -10 10 % KILIM ILIM Set-Point Scale Factor 258 ILIMTC Temperature Coefficient 4000 ppm/°C Continued on the following page…

© 2015 Fairchild Semiconductor Corporation www.fairchildsemi.com FAN2356A • Rev. 1.0 6 FAN2356AMPX — 6 A Synchronous Buck Regulator Electrical Characteristics (Continued) Unless otherwise noted; VIN=12 V, VOUT=1.2 V, and TA=TJ = -40 to +125°C. (1) Symbol Parameter Condition Min. Typ. Max. Unit Enable VTH+ Rising Threshold 2.0 V VTH- Falling Threshold 0.8 V IENLK Enable Pin Leakage EN=1.2 V 100 nA IENLK Enable Pin Leakage VEN=5 V 76 µA UVLO VON VCC Good Threshold Rising 4.4 V VHYS Hysteresis Voltage 160 mV Fault Protection VUVP PGOOD UV Trip Point On FB Falling 86 89 92 % VVOP1 PGOOD OV Trip Point On FB Rising 108 111 115 % VOVP2 Second OV Trip Point On FB Rising; LS=On 118 122 125 % RPGOOD PGOOD Pull-Down Resistance IPGOOD=2 mA 125 Ω tPG,SSDELAY PGOOD Soft-Start Delay 0.82 1.42 2.03 ms IPG,LEAK PGOOD Leakage Current 1 µA Thermal Shutdown TOFF Thermal Shutdown Trip Point(2) 155 °C THYS Hysteresis(2) 15 °C Internal Bootstrap Diode VFBOOT Forward Voltage IF=10 mA 0.6 V IR Reverse Leakage VR=24 V 1000 µA Notes: 1. Device is 100% production tested at TA=25°C. Limits over that temperature are guaranteed by design. 2. Guaranteed by design; not production tested.

no airflow; unless otherwise specified. Figure 5. Efficiency vs. Load Current with VIN=19 V Figure 6. Efficiency vs. Load Current with VIN=12 V Figure 7. Efficiency vs. Load Current with VIN=19 V Figure 8. Efficiency vs. Load Current with VIN=12 V Figure 9. Case Temperature Rise vs. Load Current on

4 Layer PCB, 1 oz Copper, 7 cm x 7 cm

Figure 10. Case Temperature Rise vs. Load Current

© 2015 Fairchild Semiconductor Corporation www.fairchildsemi.com FAN2356A • Rev. 1.0 11 FAN2356AMPX — 6 A Synchronous Buck Regulator Circuit Operation The FAN2356A uses a constant on -time modulation architecture with a V IN feed-forward input to accommodate a wide V IN range. This method provides fixed switching frequency (f SW) operation when the inductor operates in Continuous Conduction Mode (CCM) and variable frequency when operating in Pulse Frequency Mode (PFM) at light loads. Additional benefits include excellent line and load transient response, cycle -by-cycle current limiting, and no loop compensation is required. At the beginning of each cycle, FAN2356A turns on the high-side MOSFET (HS) for a fixed duration (t ON). At the end of t ON, HS turns off for a duration (tOFF) determined by the operating conditions. Once the FB voltage (V FB) falls below the reference voltage (VREF), a new switching cycle begins. The modulator provides a minimum off -time (tOFF-MIN) of 320 ns to provide a guaranteed interval for low -side MOSFET (LS) current sensing and PFM operation. t OFF- MIN is also used to provide stability against multiple pulsing and limits maximum switching frequency during transient events. Enable The enable pin is TTL compatible, which supports low - shutdown-current applications, such as notebooks. VCC should be applied after V IN / PV IN is applied to the circuit. The EN pin can be directly driven by logic voltages of 5 V, 3.3 V, 2.5 V, etc. If the EN pin is driven by 5 V logic, a small current flows into the pin when the EN pin voltage exceeds the internal clamp voltage of 4.3 V. To eliminate clamp current flowing into the EN pin use a voltage divider to limit the EN pin voltage to < 4 V. Constant On-time Modulation The FAN2356A uses a constant on -time modulation technique, in which the HS MOSFET is turned on for a fixed time, set by the modulator, in response to the input voltage and the frequency setting resistor. This on -time is proportional to the desired output voltage, divided by the input voltage. With this proportionality, the frequency is essentially constant over the load range where inductor current is continuous. For buck converter in Continuous -Conduction Mode (CCM), the switching frequency fSW is expressed as: (1) The on -time generator sets the on -time (t ON) for the high-side MOSFET, which results in the switching frequency of the regulator during steady-state operation. To maintain a relatively constant switching frequency over a wide range of input conditions, the input voltage information is fed into the on-time generator. tON is determined by: (2) where ItON is: (3) where R FREQ is the frequency -setting resistor described in the Setting Switching Frequency section; CtON is the internal 2.3 pF capacitor; and ItON is the VIN feed-forward current that generates the on-time. The FAN2356A implements open -circuit detection on the FREQ pin to protect the output from an infinitely long on-time. In the event the FREQ pin is left floating, switching of the regulator is disabled. The FAN2356A is designed for VIN input range 4.5 to 25 V, fSW 200 kHz to 1.5 MHz, resulting in an ItON ratio exceeding 1 to 25. As the ratio of VOUT to VIN increases, tOFF,min introduces a limit on the maximum switching frequency as calculated in the following equation, where the factor 1.2 is included in the denominator to add some headroom for transient operation: (4) Soft-Start (SS) A conventional soft-start ramp is implemented to provide a controlled startup sequence of the output voltage. A current is generated on the SS pin to charge an external capacitor. The lesser of the voltage on the SS pin and the reference voltage is used for output regulation. To reduce V OUT ripple and achieve a smoother ramp of the output voltage, tON is modulated during soft-start. tON starts at 50% of the steady -state on-time (PWM Mode) and ramps up to 100% gradually. During normal operation, the SS voltage is clamped to 400 mV above the FB voltage. The clamp voltage drops to 40 mV during an overload condition to allow the converter to recover using the soft -start ramp once the overload condition is removed. O n-time modulation during SS is disabled when an overload condition exists. To maintain a monotonic soft-start ramp, the regulator is forced into PFM Mode during soft -start. The minimum frequency clamp is disabled during soft-start. The nominal startup tim e is programmable through an internal current source charging the external soft -start capacitor CSS: (5) where: CSS = External soft-start programming capacitor; ISS = Internal soft -start charging current source, 10 A; tSS = Soft-start time; and VREF = 600 mV For example; for 1 ms startup time, CSS=15 nF. The soft-start option can be used for ratiometric tracking. When EN is LOW, the soft-start capacitor is discharged.

© 2015 Fairchild Semiconductor Corporation FAN2356A • Rev. 1.0 12 www.fairchildsemi.com FAN2356A -— 6 A Synchronous Buck Regulator Startup on Pre-Bias FAN2356A allows the regulator to start on a pre -bias output, VOUT, and ensures VOUT is not discharged during the soft-start operation. To guarantee no glitches on VOUT at the beginning of the soft-start ramp, the LS is disabled until the first positive - going edge of the PWM signal. The regulator is also forced into PFM Mode during soft -start to ensure the inductor current remains positive, reducing the possibility of discharging the output voltage. PVCC The FAN2356A requires an external source connected to PVCC to supply power to the internal gate drivers. The PVCC pin should be bypassed with a 2.2 µF ceramic capacitor. VCC Bias Supply and UVLO The V CC rail supplies po wer to the controller. It is generally connected to the PVCC rail through a low - pass filter of a 10  resistor and 0.1 µF capacitor to minimize any noise sources from the driver supply. An Under -Voltage Lockout (UVLO) circuit monitors the VCC voltage to ensure proper operation. Once the V CC voltage is above the UVLO threshold, the part begins operation after an initialization routine of 50 µs. There is no UVLO circuitry on either the PVCC or VIN rails. Pulse Frequency Modulation (PFM) One of t he key benefits of using a constant on -time modulation scheme is the seamless transitions in and out of Pulse Frequency Modulation (PFM) Mode. The PWM signal is not slave to a fixed oscillator and, therefore, can operate at any frequency below the target steady-state frequency. By reducing the frequency during light -load conditions, the efficiency can be significantly improved. The FAN2356A provides a Zero -Crossing Detector (ZCD) circuit to identify when the current in the inductor reverses direction. To im prove efficiency at light load, the LS MOSFET is turned off around the zero crossing to eliminate negative current in the inductor. For predictable operation entering PFM mode the controller waits for nine consecutive zero crossings before allowing the LS MOSFET to turn off. In PFM Mode, f SW varies or modulates proportionally to the load; as load decreases, f SW also decreases. The switching frequency, while the regulator is operating in PFM, can be expressed as: (6) where L is inductance and IOUT is output load current. Minimum Frequency Clamp To maintain a switching frequency above the audible range, the FAN2356A clamps the switching frequency to a minimum value of 18 kHz. The LS MOSFET is turned on to discharge the output and trigger a new PWM cycle. The minimum frequency clamp is disabled during soft-start. Protection Features The converter output is monitored and protected against over-current, over -voltage, under -voltage, and high - temperature conditions. Over-Current Protection (OCP) The FAN2356A uses current information through the LS to implement valley -current limiting. While an OC event is detected, the HS is prevented from turning on and the LS is kept on until the cu rrent falls below the user - defined set point. Once the current is below the set point, the HS is allowed to turn on. During an OC event, the output voltage may droop if the load current is greater than the current the converter is providing. If the output voltage drops below the UV threshold, an overload condition is triggered. During an overload condition, the SS clamp voltage is reduced to 40 mV and the on -time is fixed at the steady -state duration. By nature of the control method; as VOUT drops, the switching frequency is lower due to the reduced rate of inductor current decay during the off-time. The ILIM pin has an open -detection circuit to provide protection against operation without a current limit. Under-Voltage Protection (UVP) If VFB is below the under-voltage threshold of -11% VREF (534 mV), the part enters UVP and PGOOD pulls LOW. Over-Voltage Protection (OVP) There are two levels of OV protection: +11% and + 22%. During an over-voltage event, PGOOD pulls LOW. When VFB is > +11% of V REF (666 mV), both HS and LS turn off. By turning off the LS during an OV event, V OUT overshoot can be reduced when there is positive inductor current by increasing the rate of discharge. Once the V FB voltage falls below V REF, the latched OV signal is cleared and operation returns to normal. A second over -voltage detection is implemented to protect the load from more serious failure. When V FB rises +22% above the VREF (732 mV), the HS latches off until a power cycle on VCC and the LS is forced on until 530 mV of VFB. Over-Temperature Protection (OTP) The FAN2356A incorporates an over -temperature protection circuit that disables the converter when the die temperature reaches 155°C. The IC restarts when the die temperature falls below 140°C. Power Good (PGOOD) The PGOOD pin serves as an indication to the system that the output voltage of the regulator is stable and within regulation. Whenever V OUT is outside the regulation window or the regulator is at over - temperature (UV, OV, and OT), the PGOOD pin is pulled LOW. PGOOD is an open-drain output that asserts LOW when VOUT is out of regulation or when OT is detected.

© 2015 Fairchild Semiconductor Corporation FAN2356A • Rev. 1.0 13 www.fairchildsemi.com FAN2356A -— 6 A Synchronous Buck Regulator

Application Information

Constant on -time stability consists of two parameters: stability criterion and sufficient signal at VFB. Stability criterion is given by: (7) Sufficient signal requirement is given by: (8) where IIND is the inductor current ripple and VFB is the ripple voltage on VFB, which should be ≥12 mV. In certain applications, especially designs utilizing only ceramic output capacitors, there may not be sufficient ripple magnitude available on the feedback pin for stable operation. In this case, an external circuit consisting of 2 resistors (R2 and R6) and 2 capacitors (C4 and C5) can be added to inject ripple voltage into the FB pin (see Figure 1). There are some specific considerations when sel ecting the RCC ripple injector circuit. For typical applications, use 4.99 kΩ for R6; the value of C4 can be selected as 0.1 µF and approximate values for R2 and C5 can be determined using the following equations. R2 must be small enough to develop 12 mV of ripple: (9) R2 must be selected such that the R2C4 time constant enables stable operation: (10) The minimum value of C5 can be selected to minimize the capacitive component of ripple appearing on the feedback pin: (11) Using the minimum value of C5 generally offers the best transient response, and 100 pF is a good initial value in many applications. However, under some operating conditions excessive pulse jitter may be observed. To reduce jitter and improve stability, the value of C5 can be increased: (12)

5 V PVCC

The PVCC is supplied from an external source to provide power to the drivers and VCC. It is crucial to keep this pin decoupled to PGND with a ≥1 µF X5R or X7R ceramic capacitor. Because VCC powers internal analog circuit, it is filtered from PVCC with a 10 Ω resistor and 0.1 µF X7R decoupling ceramic capacitor to AGND. Setting the Output Voltage (VOUT) The output voltage VOUT is regulated by initiating a high - side MOSF ET on -time interval when the valley of the divided output voltage appearing at the FB pin reaches VREF. Since this method regulates at the valley of the output ripple voltage, the actual DC output voltage on VOUT is offset from the programmed output voltag e by the average value of the output ripple voltage. The initial VOUT setting of the regulator can be programmed from 0.6 V to 5.5 V by an external resistor divider ( R3 and R4): (13) where VREF is 600 mV. For example; for 1.2 V VOUT and 10 k R3, then R4 is 10 k. For 600 mV V OUT, R4 is left open. VFB is trimmed to a value of 596 mV when V REF=600 mV, so the final output voltage, including the effect of the output ripple voltage, can be approximated by the equation: (14) Setting the Switching Frequency (fSW) fSW is programmed through external RFREQ as follows: (15) where C tON=2.2 pF internal capacitor that generates tON. For example; for f SW=500 kHz and V OUT=1.2 V, select a standard resistor value for RFREQ=54.9 k. Inductor Selection The inductor is typically selected based on the ripple current (IL), which is approximately 25% to 45% of the maximum DC load. The inductor current rating should be selected such that the saturation and heating current ratings exceed the intended currents encountered in the application over the expected temperature range of operation. Regulators that require fast transient response use smaller inductance and higher current ripple; while regulators that require higher efficiency keep ripple current on the low side. The inductor value is given by: (16) For example: for 19 V V IN, 1.2 V V OUT, 6 A load, 30% IL, and 500 kHz fSW; L is 1.2 µH.

© 2015 Fairchild Semiconductor Corporation www.fairchildsemi.com FAN2356A • Rev. 1.0 14 FAN2356AMPX — 6 A Synchronous Buck Regulator Input Capacitor Selection Input capacitor C IN is selected based on voltage rating, RMS current I CIN(RMS) rating, and capacitance. For capacitors having DC voltage bias derating, such as ceramic capacitors, higher rating is strongly recommended. RMS current rating is given by: (17) where ILOAD-MAX is the maximum load current and D is the duty cycle VOUT/VIN. The maximum ICIN(RMS) occurs at 50% duty cycle. The capacitance is given by: (18) where VIN is the input voltage ripple, normally 1% of VIN. For example; for V IN=19 V, VIN=120 mV, V OUT=1.2 V, 6 A load, and f SW=500 kHz; CIN is 5.9 µF and is I CIN(RMS) is 1. 45 ARMS. Select two 10 µF 25 V-rated ceramic capacitors with X7R or similar dielectric, rec ognizing that the capacitor DC bias characteristic indicates that the capacitance value falls approximately 60% at VIN=19 V. Also, the 10 µF X7R capacitor ca carry over

3 ARMS in the frequency range from 100 kHz to 1 MHz,

exceeding the input capacitor curr ent rating requirements. An additional 0.1 µF capacitor may be needed to suppress noise generated by high frequency switching transitions. Output Capacitor Selection Output capacitor COUT is also selected based on voltage rating, RMS current I COUT(RMS) rating, and capacitance. For capacitors having DC voltage bias derating, such as ceramic capacitors, higher rating is highly recommended. When calculating C OUT, usually the dominant requirement is the current load step transient. If the unloading transien t requirement (I OUT transitioning from HIGH to LOW), is satisfied, then the load transient (I OUT transitioning LOW to HIGH), is also usually satisfied. The unloading C OUT calculation, assuming C OUT has negligible parasitic resistance and inductance in the circuit path, is given by: (19) where IMAX and IMIN are maximum and minimum load steps, respectively and VOUT is the voltage overshoot, usually specified at 3 to 5%. For example: for VI=19 V, VOUT=1.2 V, 4 A IMAX, 2 A IMIN, fSW=500 kHz, L OUT=1.2 µH, and 3% VOUT ripple of 36 mV; the C OUT value is calculated to be 164 µF. This capacitor requirement can be satisfied using four 47 µF, 6.3 V-rated X5R ceramic capacitors. This calculation applies for load current slew rates that are faster than the inductor current slew rate, which can be defined as VOUT/L during the load current removal. For reduced - load-current slew rates and/or reduced transient requirements, the output capac itor value may be reduced and comprised of low-cost 22 µF capacitors. Setting the Current Limit Current limit is implemented by sensing the inductor valley current across the LS MOSFET VDS during the LS on-time. The current limit comparator prevents a new on-time from being started until the valley current is less than the current limit. The set point is configured by connecting a resistor from the ILIM pin to the SW pin. A trimmed current is output onto the ILIM pin, which creates a voltage across the resistor. When the voltage on ILIM goes negative, an over-current condition is detected. RILIM is calculated by: (20) where KILIM is the current source scale factor, and IVALLEY is the inductor valley current when the current limit threshold is reached. The factor 1.02 accounts for the temperature offset of the LS MOSFET compared to the control circuit. With the constant on -time architecture, HS is always turned on for a fixed on -time; this determines the peak - to-peak inductor current. Current ripple I is given by: (21) From the equation above, the worst -case ripple occurs during an output short circuit (where V OUT is 0 V). This should be taken into account when selecting the current limit set point. The FAN2356A uses valley-current sensing, the current limit (IILIM) set point is the valley (IVALLEY). The valley current level for calculating RILIM is given by: (22) where ILOAD (CL) is the DC load current when the current limit threshold is reached. For example: In a converter designed for 6 A steady - state operation and 1.8 A current ripple, the current-limit threshold could be selected at 120% of I LOAD,(MAX) to accommodate transient operation and inductor value decrease under loading. As a result, I LOAD,(MAX) is 7.2 A, IVALLEY=6.3 A, and R ILIM is selected as the standard value of 1.65 k

© 2015 Fairchild Semiconductor Corporation www.fairchildsemi.com FAN2356A • Rev. 1.0 15 FAN2356AMPX — 6 A Synchronous Buck Regulator Boot Resistor In some applications, especially with higher input voltage, the VSW ring voltage may exceed derating guidelines of 80% to 90% of absolute rating for VSW. In this situation a resistor can be connected in series with boot c apacitor (C3 in Figure 1) to reduce the turn -on speed of the high side MOSFET to reduce the amplitude of the V SW ring voltage. If necessary, a resistor and capacito r snubber can be added from VSW to PGND to reduce the magnitude of the ringing voltage. Please contact Fairchild Customer Support for assistance selecting a boot resistor or snubber circuit in appl ications that operate above a 21 V typical input voltage. Printed Circuit Board (PCB) Layout Guidelines The following points should be considered before beginning a PCB layout using the FAN2356A. A sample PCB layout from the evaluation board is shown in Figure 25-Figure 28 following the layout guidelines. Power components consisting of the input capacitors, output capacitors, inductor, and device should be placed on a common side of the pcb in close proximity to each other and connected using surface copper. Sensitive analog components includin g SS, FB, ILIM, FREQ, and EN should be placed away from the high - voltage switching circuits such as SW and BOOT, and connected to their respective pins with short traces. The inner PCB layer closest to the device should have Power Ground (PGND) under the p ower processing portion of the device (PVIN, SW, and PGND). This inner PCB layer should have a separate Analog Ground (AGND) under the P1 pad and the associated analog components. AGND and PGND should be connected together near the IC between PGND pins 18 -21 and AGND pin 23 which connects to P1 thermal pad. The AGND thermal pad (P1) should be connected to AGND plane on inner layer using four 0.25 mm vias spread under the pad. No vias are included under PVIN (P2) and SW (P3) to maintain the PGND plane under the power circuitry intact. Power circuit loops that carry high currents should be arranged to minimize the loop area. Primary focus should be directed to minimize the loop for current flow from the input capacitor to PVIN, through the internal MOSFETs, an d returning to the input capacitor. The input capacitor should be placed as close to the PVIN terminals as possible. The current return path from PGND at the low -side MOSFET source to the negative terminal of the input capacitor can be routed under the in ductor and also through vias that connect the input capacitor and low - side MOSFET source to the PGND region under the power portion of the IC. The SW node trace which connects the source of the high-side MOSFET and the drain of the low -side MOSFET to the inductor should be short and wide. To control the voltage across the output capacitor, the output voltage divider should be located close to the FB pin, with the upper FB voltage divider resistor connected to the positive side of the output capacitor, and the bottom resistor should be connected to the AGND portion of the device. When using ceramic capacitor solutions with external ramp injection circuitry (R2, C4, C5 in Figure 1), R2 and C4 should be connected near the inductor, and coupling capacitor C5 should be placed near FB pin to minimize FB pin trace length. Decoupling capacitors for PVCC and VCC should be located close to their respective device pins. SW node connections to BOOT, ILIM, and ripple injection resistor R2 should be made through separate traces.

NOTES: UNLESS OTHERWISE SPECIFIED A) NO INDUSTRY REGISTRATION APPLIES. B) ALL DIMENSIONS ARE IN MILLIMETERS. C) DIMENSIONS DO NOT INCLUDE BURRS OR MOLD FLASH. MOLD FLASH OR BURRS DOES NOT EXCEED 0.10MM. D) DIMENSIONING AND TOLERANCING PER ASME Y14.5M-2009. E) DRAWING FILE NAME: MKT-PQFN34AREV2 F) FAIRCHILD SEMICONDUCTOR SEE DETAIL 'A' SCALE: 2:1 SEATING PLANE 0.25±0.05 0.025±0.025 1.05±0.10 5.50±0.10 5.00±0.10 (30X) PIN#1 INDICATOR 18 26 18 26 0.25±0.05 (0.43) (0.35) 0.50±0.01 0.40±0.01 (0.35) (1.75) 3.50±0.01 (0.35) (0.75) (0.35) (0.24) (0.28) 1.75±0.01 (0.25) 0.43±0.01 (0.33) 2.58±0.01 (0.35) 0.68±0.01 (0.25) (30X) (3X)

1.80 (0.35) 2.10 2.18 1.58 5.70 0.55 (30X) (1.75) (1.85) 3.50 4.10 (0.08) 0.43 (0.35) (0.30) 0.30 (30X) 0.20 2.58 0.75 0.68 3.60 5.20 0.55 0.50±0.05 4.10

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