CMMSH1-100

LM3402_11 TI1 | Alldatasheet

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LM3402,LM3402HV LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs Literature Number: SNVS450D

February 5, 2010 0.5A Constant Current Buck Regulator for Driving High Power LEDs General Description The LM3402/02HV are monolithic switching regulators de- signed to deliver constant currents to high power LEDs. Ideal for automotive, industrial, and general lighting applications, they contain a high-side N-channel MOSFET switch with a current limit of 735 mA (typical) for step-down (Buck) regula- tors. Hysteretic control with controlled on-time coupled with an external resistor allow the converter output voltage to ad- just as needed to deliver a constant current to series and series - parallel connected arrays of LEDs of varying number and type, LED dimming by pulse width modulation (PWM), broken/open LED protection, low-power shutdown and ther- mal shutdown complete the feature set.

Features

■ Integrated 0.5A N-channel MOSFET ■ VIN Range from 6V to 42V (LM3402) ■ VIN Range from 6V to 75V (LM3402HV) ■ 500 mA Output Current Over Temperature ■ Cycle-by-Cycle Current Limit ■ No Control Loop Compensation Required ■ Separate PWM Dimming and Low Power Shutdown ■ Supports all-ceramic output capacitors and capacitor-less outputs ■ Thermal shutdown protection ■ MSOP-8, PSOP-8 Packages

Applications

■ LED Driver ■ Constant Current Source ■ Automotive Lighting ■ General Illumination ■ Industrial Lighting Typical Application 20192101 © 2010 National Semiconductor Corporation 201921 www.national.com LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs

8-Lead Plastic MSOP-8 Package 20192145 8-Lead Plastic PSOP-8 Package

Ordering Information

Order Number Package Type NSC Package Drawing Supplied As LM3402MM MSOP-8 MUA08A 1000 units on tape and reel LM3402MMX 3500 units on tape and reel LM3402HVMM 1000 units on tape and reel LM3402HVMMX 3500 units on tape and reel LM3402MR PSOP-8 MRA08B 95 units in anti-static rails LM3402MRX 2500 units on tape and reel LM3402HVMR 95 units in anti-static rails LM3402HVMRX 2500 units on tape and reel Pin Descriptions Pin(s) Name Description Application Information 1 SW Switch pin Connect this pin to the output inductor and Schottky diode. 2 BOOT MOSFET drive bootstrap pin Connect a 10 nF ceramic capacitor from this pin to SW.

3 DIM Input for PWM dimming Connect a logic-level PWM signal to this pin to enable/disable the

power FET and reduce the average light output of the LED array. 4 GND Ground pin Connect this pin to system ground.

5 CS Current sense feedback pin Set the current through the LED array by connecting a resistor from

this pin to ground.

6 RON On-time control pin A resistor connected from this pin to VIN sets the regulator controlled

on-time.

7 VCC Output of the internal 7V linear

Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor with X5R or X7R dielectric.

8 VIN Input voltage pin Nominal operating input range is 6V to 42V (LM3402) or 6V to 75V

(LM3402HV). DAP GND Thermal Pad Connect to ground. Place 4 to 6 vias from DAP to bottom layer ground plane. www.national.com 2 LM3402/LM3402HV

(LM3402) (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND -0.3V to 45V BOOT to GND -0.3V to 59V SW to GND -1.5V BOOT to VCC -0.3V to 45V BOOT to SW -0.3V to 14V VCC to GND -0.3V to 14V DIM to GND -0.3V to 7V CS to GND -0.3V to 7V RON to GND -0.3V to 7V Junction Temperature 150°C Storage Temp. Range -65°C to 125°C ESD Rating (Note 2) 2kV Soldering Information Lead Temperature (Soldering, 10sec) 260°C Infrared/Convection Reflow (15sec) 235°C Operating Ratings (LM3402) (Note 1) VIN 6V to 42V Junction Temperature Range −40°C to +125°C Thermal Resistance θJA (MSOP-8 Package) (Note 3) 200°C/W Thermal Resistance θJA (PSOP-8 Package) (Note 5) 50°C/W 3 www.national.com LM3402/LM3402HV

(LM3402HV) (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND -0.3V to 76V BOOT to GND -0.3V to 90V SW to GND -1.5V BOOT to VCC -0.3V to 76V BOOT to SW -0.3V to 14V VCC to GND -0.3V to 14V DIM to GND -0.3V to 7V CS to GND -0.3V to 7V RON to GND -0.3V to 7V Junction Temperature 150°C Storage Temp. Range -65°C to 125°C ESD Rating (Note 2) 2kV Soldering Information Lead Temperature (Soldering, 10sec) 260°C Infrared/Convection Reflow (15sec) 235°C Operating Ratings (LM3402HV) (Note 1) VIN 6V to 75V Junction Temperature Range −40°C to +125°C Thermal Resistance θJA (MSOP-8 Package) (Note 3) 200°C/W Thermal Resistance θJA (PSOP-8 Package) (Note 5) 50°C/W www.national.com 4 LM3402/LM3402HV

Electrical Characteristics VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA = TJ = +25°C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. LM3402 Symbol Parameter Conditions Min Typ Max Units SYSTEM PARAMETERS tON-1 On-time 1 VIN = 10V, RON = 200 kΩ 2.1 2.75 3.4 µs tON-2 On-time 2 VIN = 40V, RON = 200 kΩ 490 650 810 ns LM3402HV Symbol Parameter Conditions Min Typ Max Units SYSTEM PARAMETERS tON-1 On-time 1 VIN = 10V, RON = 200 kΩ 2.1 2.75 3.4 µs tON-2 On-time 2 VIN = 70V, RON = 200 kΩ 290 380 470 ns LM3402/LM3402HV Symbol Parameter Conditions Min Typ Max Units REGULATION AND OVER-VOLTAGE COMPARATORS VREF-REG CS Regulation Threshold CS Decreasing, SW turns on 194 200 206 mV VREF-0V CS Over-voltage Threshold CS Increasing, SW turns off 300 mV ICS CS Bias Current CS = 0V 0.1 µA SHUTDOWN VSD-TH Shutdown Threshold RON / SD Increasing 0.3 0.7 1.05 V VSD-HYS Shutdown Hysteresis RON / SD Decreasing 40 mV OFF TIMER tOFF-MIN Minimum Off-time CS = 0V 300 ns INTERNAL REGULATOR VCC-REG VCC Regulated Output 6.6 7 7.4 V VIN-DO VIN - VCC Dropout ICC = 5 mA, 6.0V < VIN < 8.0V 300 mV VCC-BP-TH VCC Bypass Threshold VIN Increasing 8.8 V VCC-BP-HYS VCC Bypass Hysteresis VIN Decreasing 225 mV VCC-Z-6 VCC Output Impedance (0 mA < ICC < 5 mA) VIN = 6V 55 Ω VCC-Z-8 VIN = 8V 50 VCC-Z-24 VIN = 24V 0.4 VCC-LIM VCC Current Limit (Note 3) VIN = 24V, VCC = 0V 16 mA VCC-UV-TH VCC Under-voltage Lock-out Threshold VCC Increasing 5.25 V VCC-UV-HYS VCC Under-voltage Lock-out Hysteresis VCC Decreasing 150 mV VCC-UV-DLY VCC Under-voltage Lock-out Filter Delay 100 mV Overdrive 3 µs IIN-OP IIN Operating Current Non-switching, CS = 0V 600 900 µA IIN-SD IIN Shutdown Current RON / SD = 0V 90 180 µA CURRENT LIMIT ILIM Current Limit Threshold 530 735 940 mA 5 www.national.com LM3402/LM3402HV

Symbol Parameter Conditions Min Typ Max Units DIM COMPARATOR VIH Logic High DIM Increasing 2.2 V VIL Logic Low DIM Decreasing 0.8 V IDIM-PU DIM Pull-up Current DIM = 1.5V 75 µA N-MOSFET AND DRIVER RDS-ON Buck Switch On Resistance ISW = 200mA, BOOT-SW = 6.3V 0.7 1.5 Ω VDR-UVLO BOOT Under-voltage Lock-out Threshold BOOT–SW Increasing 1.7 3 4 V VDR-HYS BOOT Under-voltage Lock-out Hysteresis BOOT–SW Decreasing 400 mV THERMAL SHUTDOWN TSD Thermal Shutdown Threshold 165 °C TSD-HYS Thermal Shutdown Hysteresis 25 °C THERMAL RESISTANCE θJA Junction to Ambient MSOP-8 Package 200 °C/W Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading. Note 4: Typical specifications represent the most likely parametric norm at 25°C operation. Note 5: θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1 oz. copper on the top or bottom PCB layer. www.national.com 6 LM3402/LM3402HV

Typical Performance Characteristics VREF vs Temperature (VIN = 24V) 20192129 VREF vs VIN, LM3402 (TA = 25°C) 20192130 VREF vs VIN, LM3402HV (TA = 25°C) 20192131 Current Limit vs Temperature (VIN = 24V) 20192132 Current Limit vs VIN, LM3402 (TA = 25°C) 20192133 Current Limit vs VIN, LM3402HV (TA = 25°C) 20192134 7 www.national.com LM3402/LM3402HV

TON vs VIN, RON = 100 kΩ (TA = 25°C) 20192135 TON vs VIN, (TA = 25°C) 20192136 TON vs VIN, (TA = 25°C) 20192137 TON vs RON, LM3402 (TA = 25°C) 20192144 TON vs RON, LM3402HV (TA = 25°C) 20192138 VCC vs VIN (TA = 25°C) 20192139 www.national.com 8 LM3402/LM3402HV

VO-MAX vs fSW, LM3402 (TA = 25°C) 20192140 VO-MIN vs fSW, LM3402 (TA = 25°C) 20192141 VO-MAX vs fSW, LM3402HV (TA = 25°C) 20192142 VO-MIN vs fSW, LM3402HV (TA = 25°C) 20192143 9 www.national.com LM3402/LM3402HV

Application Information

The LM3402 and LM3402HV are buck regulators with a wide input voltage range, low voltage reference, and a fast output enable/disable function. These features combine to make them ideal for use as a constant current source for LEDs with forward currents as high as 500 mA. The controlled on-time (COT) architecture is a combination of hysteretic mode con- trol and a one-shot on-timer that varies inversely with input voltage. Hysteretic operation eliminates the need for small- signal control loop compensation. When the converter runs in continuous conduction mode (CCM) the controlled on-time maintains a constant switching frequency over the range of input voltage. Fast transient response, PWM dimming, a low power shutdown mode, and simple output overvoltage pro- tection round out the functions of the LM3402/02HV. CONTROLLED ON-TIME OVERVIEW Figure 1 shows the feedback system used to control the cur- rent through an array of LEDs. A voltage signal, V SNS, is created as the LED current flows through the current setting resistor, R SNS, to ground. V SNS is fed back to the CS pin, where it is compared against a 200 mV reference, VREF. The on-comparator turns on the power MOSFET when VSNS falls below VREF. The power MOSFET conducts for a controlled on-time, tON, set by an external resistor, RON, and by the input voltage, VIN. On-time is governed by the following equation: At the conclusion of t ON the power MOSFET turns off for a minimum off-time, tOFF-MIN, of 300 ns. Once t OFF-MIN is com- plete the CS comparator compares V SNS and V REF again, waiting to begin the next cycle. www.national.com 10 LM3402/LM3402HV

FIGURE 1. Comparator and One-Shot Design Considerations section.

a 7V output, connected between the VIN and the VCC pins. when its source voltage is equal to VIN. steady state switching frequency in order to prevent aliasing. current while the MOSFET is on) exceeds 735 mA (typical). 300 mV / RSNS by this comparator during transients. maximum recommended value for RZ is 1 kΩ. the inductor will oscillate and can go above VIN or below 0V. to rise in the case of an open-circuit LED failure. FIGURE 2. Output Open Circuit Protection

current does not exceed 300 mV / RSNS. FIGURE 3. Low Power Shutdown down the MOSFET and driver are disabled.

due to low forward drop and near-zero reverse recovery time. FIGURE 5. LED Current From SW Pin

pendent. It will not affect regular PWM dimming operation. the SW pin to ground according to the chart below. FIGURE 6. CS Pin, Transient Path

FIGURE 10. VIN Pin with Typical Input Protection

To select RON the expression relating tON to input voltage from the Controlled On-time Overview section can be re-written as: Minimum on-time occurs at the maximum VIN, which is 24V x 110% = 26.4V. RON is therefore calculated as: RON = (300 x 10-9 x 26.4) / 1.34 x 10-10 = 59105 Ω The closest 1% tolerance resistor is 59.0 k Ω. The switching frequency of the circuit can then be found using the equation relating RON to fSW: fSW = 3.7 / (59000 x 1.34 x 10-10) = 468 kHz USING AN OUTPUT CAPACITOR The inductor will be the largest component used in this design. Because the application does not require any PWM dimming, an output capacitor can be used to greatly reduce the induc- tance needed without worry of slowing the potential PWM dimming frequency. The total solution size will be reduced by using an output capacitor and small inductor as opposed to one large inductor. OUTPUT INDUCTOR Knowing that an output capacitor will be used, the inductor can be selected for a larger current ripple. The desired max- imum value for ΔiL is ±30%, or 0.6 x 350 mA = 210 mA P-P. Minimum inductance is selected at the maximum input volt- age. Re-arranging the equation for current ripple selection yields the following: The closest standard inductor value is 33 µH. Off-the-shelf inductors rated at 33 µH are available from many magnetics manufacturers. Inductor datasheets should contain three specifications which are used to select the inductor. The first of these is the aver- age current rating, which for a buck regulator is equal to the average load current, or IF. The average current rating is given by a specified temperature rise in the inductor, normally 40° C. For this example, the average current rating should be greater than 350 mA to ensure that heat from the inductor does not reduce the lifetime of the LED or cause the LM3402 to enter thermal shutdown. The second specification is the tolerance of the inductance itself, typically ±10% to ±30% of the rated inductance. In this example an inductor with a tolerance of ±20% will be used. With this tolerance the typical, minimum, and maximum in- ductor current ripples can be calculated: = 206 mAP-P = 172 mAP-P = 258 mAP-P The third specification for an inductor is the peak current rat- ing, normally given as the point at which the inductance drops off by a given percentage due to saturation of the core. The worst-case peak current occurs at maximum input voltage and at minimum inductance, and can be determined with the equation from the Design Considerations section: IL(PEAK) = 0.35 + 0.258 / 2 = 479 mA For this example the peak current rating of the inductor should be greater than 479 mA. In the case of a short circuit across the LED array, the LM3402 will continue to deliver rated cur- rent through the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition is equal to: = 298 mAP-P IL(PEAK) = 0.35 + 0.149 = 499 mA In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 735 mA. In order to prevent in- ductor saturation during these short circuits the inductor’s peak current rating must be above 735 mA. The device se- lected is an off-the-shelf inductor rated 33 µH ±20% with a DCR of 96 mΩ and a peak current rating of 0.82A. The phys- ical dimensions of this inductor are 7.0 x 7.0 x 4.5 mm. RSNS The current sensing resistor value can be determined by re- arranging the expression for average LED current from the LED Current Accuracy section: RSNS = 0.74Ω, tSNS = 220 ns Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.75Ω resistor will give the best accuracy of the aver- age LED current. To determine the resistor size the power dissipation can be calculated as: PSNS = (IF)2 x RSNS PSNS = 0.352 x 0.75 = 92 mW Standard 0805 size resistors are rated to 125 mW and will be suitable for this application. 21 www.national.com LM3402/LM3402HV

To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to yield the following: The target tolerance for LED ripple current is ±5% or 10% P- P = 35 mAP-P, and the LED datasheet gives a typical value for rD of 1.0Ω at 350 mA. The required capacitor impedance to reduce the worst-case inductor ripple current of 258 mAP-P is therefore: A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 468 kHz: CO = 1/(2 x π x 0.157 x 4.68 x 105) = 2.18 µF This calculation assumes that impedance due to the equiva- lent series resistance (ESR) and equivalent series inductance (ESL) of CO is negligible. The closest 10% tolerance capacitor value is 2.2 µF. The capacitor used should be rated to 10V or more and have an X7R dielectric. Several manufacturers pro- duce ceramic capacitors with these specifications in the 0805 case size. A typical value for ESR of 1 mΩ can be read from the curve of impedance vs. frequency in the product datasheet. INPUT CAPACITOR Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 1%P-P = 240 mV. The minimum required ca- pacitance is: CIN(MIN) = (0.35 x 300 x 10-9) / 0.24 = 438 nF In expectation that more capacitance will be needed to pre- vent power supply interaction a 1.0 µF ceramic capacitor rated to 50V with X7R dielectric in a 1206 case size will be used. From the Design Considerations section, input rms cur- rent is: IIN-RMS = 0.35 x Sqrt(0.154 x 0.846) = 126 mA Ripple current ratings for 1206 size ceramic capacitors are typically higher than 1A, more than enough for this design. RECIRCULATING DIODE The first parameter for D1 which must be determined is the reverse voltage rating. Schottky diodes are available at re- verse ratings of 30V and 40V, often in the same package, with the same forward current rating. To account for ringing a 40V Schottky will be used. The next parameters to be determined are the forward current rating and case size. In this example the low duty cycle (D = 3.7 / 24 = 15%) requires the recirculating diode D1 to carry the load current much longer than the internal power MOS- FET of the LM3402. The estimated average diode current is: ID = 0.35 x 0.85 = 298 mA Schottky diodes are available at forward current ratings of 0.5A, however the current rating often assumes a 25°C am- bient temperature and does not take into account the appli- cation restrictions on temperature rise. A diode rated for higher current may be needed to keep the temperature rise below 40°C.To determine the proper case size, the dissipa- tion and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD for a small case size such as SOD-123 in a 40V, 0.5A Schottky diode at 350 mA is approximately 0.4V and the θJA is 206°C/W. Power dissipa- tion and temperature rise can be calculated as: PD = 0.298 x 0.4 = 119 mW TRISE = 0.119 x 206 = 24.5°C According to these calculations the SOD-123 diode will meet the requirements. Heating and dissipation are among the fac- tors most difficult to predict in converter design. If possible, a footprint should be used that is capable of accepting both SOD-123 and a larger case size, such as SMA. A larger diode with a higher forward current rating will generally have a lower forward voltage, reducing dissipation, as well as having a lower θJA, reducing temperature rise. CB and CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. This term should not be confused with the op- tical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: PO = IF x VO = 0.35 x 3.7 = 1.295W Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.352 x 1.5) x 0.154 = 28 mW Gate charging and VCC loss, PG, in the gate drive and linear regulator: PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 468000 x 3 x 10-9) x 24 = 48 mW Switching loss, PS, in the internal MOSFET: PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 24 x 0.35 x (40 x 10-9) x 468000 = 78 mW AC rms current loss, PCIN, in the input capacitor: PCIN = IIN(rms)2 x ESR = (0.126)2 x 0.006 = 0.1 mW (negligible) www.national.com 22 LM3402/LM3402HV

DCR loss, PL, in the inductor PL = IF2 x DCR = 0.352 x 0.096 = 11.8 mW Recirculating diode loss, PD = 119 mW Current Sense Resistor Loss, PSNS = 92 mW Electrical efficiency, η = PO / (PO + Sum of all loss terms) = DIE TEMPERATURE TLM3402 = (PC + PG + PS) x θJA TLM3402 = (0.028 + 0.05 + 0.078) x 200 = 31°C Design Example 2: LM3402HV The second example application is an RGB backlight for a flat screen monitor. A separate boost regulator provides a 60V ±5% DC input rail that feeds three LM3402HV current regu- lators to drive one series array each of red, green, and blue 1W LEDs. The target for average LED current is 350 mA ±5% in each string. The monitor will adjust the color temperature dynamically, requiring fast PWM dimming of each string with external, parallel MOSFETs. 1W green and blue InGaN LEDs have a typical forward voltage of 3.5V, however red LEDs use AlInGaP technology with a typical forward voltage of 2.9V. In order to match color properly the design requires 14 green LEDs, twice as many as needed for the red and blue LEDs. This example will follow the design for the green LED array, providing the necessary information to repeat the exercise for the blue and red LED arrays. The circuit schematic for Design Example 2 is the same as the Typical Application on the front page. The bill of materials (green array only) can be found in Table 2 at the end of this datasheet. OUTPUT VOLTAGE Green Array: VO(G) = 14 x 3.5 + 0.2 = 49.2V Blue Array: VO(B) = 7 x 3.5 + 0.2 = 24.7V Red Array: VO(R) = 7 x 2.9 + 0.2 = 20.5V RON and tON A compromise in switching frequency is needed in this appli- cation to balance the requirements of magnetics size and efficiency. The high duty cycle translates into large conduc- tion losses and high temperature rise in the IC. For best response to a PWM dimming signal this circuit will not use an output capacitor; hence a moderate switching frequency of 300 kHz will keep the inductance from becoming so large that a custom-wound inductor is needed. This design will use only surface mount components, and the selection of off-the-shelf SMT inductors for switching regulators is poor at 1000 µH and above. RON is selected from the equation for switching fre- quency as follows: RON = 49.2 / (1.34 x 10-10 x 3 x 105) = 1224 kΩ The closest 1% tolerance resistor is 1.21 MΩ. The switching frequency and on-time of the circuit can then be found using the equations relating RON and tON to fSW: fSW = 49.2 / (1210000 x 1.34 x 10-10) = 303 kHz tON = (1.34 x 10-10 x 1210000) / 60 = 2.7 µs USING AN OUTPUT CAPACITOR This application is dominated by the need for fast PWM dim- ming, requiring a circuit without any output capacitance. OUTPUT INDUCTOR In this example the ripple current through the LED array and the inductor are equal. Inductance is selected to give the smallest ripple current possible while still providing enough ΔvSNS signal for the CS comparator to operate correctly. De- signing to a desired ΔvSNS of 25 mV and assuming that the average inductor current will equal the desired average LED current of 350 mA yields the target current ripple in the in- ductor and LEDs: ΔiF = ΔiL = ΔvSNS / RSNS, RSNS = VSNS / IF ΔiF = 0.025 / 0.57 = 43.8 mA With the target ripple current determined the inductance can be chosen: The closest standard inductor value is 680 µH. As with the previous example, the average current rating should be greater than 350 mA. Separation between the LM3402HV drivers and the LED arrays mean that heat from the inductor will not threaten the lifetime of the LEDs, but an overheated inductor could still cause the LM3402HV to enter thermal shutdown. The inductance itself of the standard part chosen is ±20%. With this tolerance the typical, minimum, and maximum in- ductor current ripples can be calculated: = 43 mAP-P = 36 mAP-P = 54 mAP-P The peak LED/inductor current is then estimated: IL(PEAK) = IL + [ΔiL(MAX)] / 2 23 www.national.com LM3402/LM3402HV

IL(PEAK) = 0.35 + 0.027 = 377 mA In the case of a short circuit across the LED array, the LM3402HV will continue to deliver rated current through the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition would be equal to: ΔiF(LED-SHORT) = [(63 – 0.2) x 2.7 x 10-6] / 544 x 10-6 = 314 mAP-P IF(PEAK) = 0.35 + 0.156 = 506 mA In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 735 mA. In order to prevent in- ductor saturation during these fault conditions the inductor’s peak current rating must be above 735 mA. A 680 µH off-the shelf inductor rated to 1.2A (peak) and 0.72A (average) with a DCR of 1.1Ω will be used for the green LED array. RSNS A preliminary value for R SNS was determined in selecting ΔiL. This value should be re-evaluated based on the calcula- tions for ΔiF: Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.56Ω device is the closest value, and a 0.125W, 0805 size device will handle the power dissipation of 69 mW. With the resistance selected, the average value of LED current is re-calculated to ensure that current is within the ±5% toler- ance requirement. From the expression for LED current ac- curacy: IF = 0.19 / 0.56 + 0.043 / 2 = 361 mA, 3% above 350 mA INPUT CAPACITOR Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 1%P-P = 600 mV. The minimum required ca- pacitance is: In expectation that more capacitance will be needed to pre- vent power supply interaction a 2.2 µF ceramic capacitor rated to 100V with X7R dielectric in an 1812 case size will be used. From the Design Considerations section, input rms cur- rent is: IIN-RMS = 0.35 x Sqrt(0.82 x 0.18) = 134 mA Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A, more than enough for this design. RECIRCULATING DIODE The input voltage of 60V ±5% requires Schottky diodes with a reverse voltage rating greater than 60V. Some manufactur- ers provide Schottky diodes with ratings of 70, 80 or 90V; however the next highest standard voltage rating is 100V. Selecting a 100V rated diode provides a large safety margin for the ringing of the switch node and also makes cross-ref- erencing of diodes from different vendors easier. The next parameters to be determined are the forward current rating and case size. In this example the high duty cycle (D = 49.2 / 60 = 82%) places less thermals stress on D1 and more on the internal power MOSFET of the LM3402. The estimated average diode current is: ID = 0.361 x 0.18 = 65 mA A Schottky with a forward current rating of 0.5A would be ad- equate, however at 100V the majority of diodes have a mini- mum forward current rating of 1A. To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD for a small case size such as SOD-123F in a 100V, 1A Schottky diode at 350 mA is approximately 0.65V and the θJA is 88°C/W. Power dissipation and temperature rise can be calculated as: PD = 0.065 x 0.65 = 42 mW TRISE = 0.042 x 88 = 4°C CB AND CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: PO = IF x VO = 0.361 x 49.2 = 17.76W Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.3612 x 1.5) x 0.82 = 160 mW Gate charging and VCC loss, PG, in the gate drive and linear regulator: PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 3 x 105 x 3 x 10-9) x 60 = 90 mW Switching loss, PS, in the internal MOSFET: PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 60 x 0.361 x 40 x 10-9 x 3 x 105 = 130 mW AC rms current loss, PCIN, in the input capacitor: PCIN = IIN(rms)2 x ESR = (0.134)2 x 0.006 = 0.1 mW (negligible) www.national.com 24 LM3402/LM3402HV

much upon the layout of the PCB as the component selection. FIGURE 13. Buck Converter Current Loops resents the high current path during the off-time.

TABLE 1. BOM for Design Example 1 TABLE 2. BOM for Design Example 2

Physical Dimensions inches (millimeters) unless otherwise noted 8-Lead MSOP Package 8-Lead PSOP Package 27 www.national.com LM3402/LM3402HV

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