AN912

AN912 MICROCHIP | Alldatasheet

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DS00912A-page 2  2004 Microchip Technology Inc. A high voltage peak detector is used to extract the basic envelope of the base station's resonant tank. The output of the peak detector will be 150 VDC with about 2V peak-to-peak of carrier ripple at 125 kHz and then about 2 mV of modulated signal. The modulation signal strength is mostly dependent on the distance between the transponder and the transmitter coil as the magnetic coupling decreases to the third power of the distance between the two devices. The next stage is a passive high-pass filter to decouple or block the high DC voltage. The DC extracted voltage is then fed into a low-pass filter, leaving the required modulating signal. The last stage is the data slicer that compares the modulating signal to some reference to extract the original signal sent by the transponder. LF Talkback receiver can be thought of as detecting and decoding an amplitude modulation (AM) signal that has a very low modulation index on a relatively large carrier. SYSTEM ASSUMPTIONS The LF Talkback system designed in this document is targeted for a LF base station that has the following characteristics and is based on the design as per AN232:  The LF Talkback signal is amplitude modulated at 200 µs multiples. This is also referred to as the basic pulse element period or T  The tank is driven by a 12V half-bridge driver.  The tank inductance is 162µH and the resonant capacitor is 10 nF with a resonant frequency of 125 kHz.  The tank Q is 25. As a result, the tank or carrier voltage is 300V peak-to-peak or 150V 0-to-peak.  Transponder induced modulation of 2 mV in magnitude needs to be detected. To get an understanding of the impedances involved, lets consider the following: using Equation 1, the equivalent parallel resistance of the tank is 3.18 kΩ . The additional parallel impedance that a transponder represents to induce a 2 mV signal on the tank is in the order of 500 MΩ . What the LF Talkback system detects is the result of a 500 MΩ resistor being switched in and out in parallel with the tank at the data rate. Therefore, it is very important that the peak detector have a high- impedance at the data rate to maintain good sensitivity. EQUATION 1: THE EFFECTIVE PARALLEL IMPEDANCE OF A RESONANT TANK THE PEAK DETECTOR There are a number of aspects to consider in designing a peak detector for this application: 1. The peak detector has to be able to operate at the high voltages of the resonant tank. 2. Maintain a good tank Q or, in other words, it should not add unnecessary loading on the main resonant tank. If it does load the tank, it will result in a lower modulation voltage induced by the transponder. 3. Reduce carrier ripple as far as possible. 4. Maintain the modulation signal. 5. Have a fast large swing dynamic response and be able to settle quickly after the field is turned on. 6. Cost of the system. Some of the peak detector requirements are conflicting and as a result, the designer has to find an acceptable compromise with the final system performance in mind. One can sacrifice a specific parameter and make up for it in a later stage where optimization of that aspect is easily accomplished. As an example to optimize requirement 3, one needs to increase the size of the capacitor C2 (Figure 2), but that will negatively affect requirements 2, 4 and 5 if a passive peak detector is used. An active peak detector could have solved the conflict, but at the 600V swing, one has little choice but to use a passive peak detector while maintaining a low-cost design. A relatively low capacitance value is chosen for C1 of 1 nF. This maintains the dynamic response requirement for settling quickly after the field is applied and does not load the tank unnecessarily. Capacitor C2 should have at least a 300 V DC peak rating and a high tolerance capacity is acceptable to save cost. An ultra fast diode is required in the peak detector with a 400V or better rating and low junction capacitance. A UF1005 diode was chosen, it has a 600V rating and 10 pF of junction capacitance. FIGURE 2: R PARALLEL = 2πLFC Q L = Tank inductance in H = 162µH Fc = Center frequency of tank = 125 kHz Q = Tank quality factor = 25 D1 HV-Env

 2004 Microchip Technology Inc. DS00912A-page 3 AN912 The envelope detector with only D1 and C2 has a greatly different response to increasing and decreasing voltage amplitudes of the resonant tank. The voltage designated by the signal HV_Env (Figure 2) rises quickly with increasing tank amplitudes because D1 has a low-impedance in forward conduction. The tank voltage decreases slower when the tank amplitude is lowered because C2 can only discharge through D1, which has a high-impedance in the reverse direction. The situation can be remedied to some extent by the introduction of R1 which helps to discharge C2, but the value of R1 should be high enough to maintain a good tank Q as per requirement 2 above. A 10 MΩ value for R1 works well, but note that R1 needs to be implemented as a series of two resistors. This is done to stay within the safe voltage range of 0805 resistors are used. The 125 kHz carrier ripple voltage, without R1, is about 2V peak-to-peak and is due to the junction capacitance and reverse leakage of D1. The addition of R1 has little effect on the ripple voltage, but does improve the detectors dynamic performance at the data rate. The carrier ripple voltage will be filtered out at a later stage where a more effective solution can be implemented. THE DC DECOUPLER CONFLICTS The HV_Env signal (Figure 2) consists of three main components: 1. A 150V DC signal, as a result of the peak detector. 2. 2V peak-to-peak ripple voltage at the carrier frequency. 3. The modulated data signal at a TE of 200µs and a 2 mV peak-to-peak amplitude, highest funda- mental harmonic content is at 2.5 kHz [1/(2*200 uS)], irrespective of the modulation scheme used (i.e., Manchester, PWM etc.). The aim of the decoupling stage is to reject the high DC voltage without adding unnecessary loading to the tank via the peak detector. It should also have a fast dynamic response and stabilize quickly after the tank is ener- gized. The dynamic response of the LF Talkback system is the major design hurdle to overcome as far as the decoupling stage is concerned. The problem is aggra- vated when the transponder needs to communicate on the LF link soon after the base station communicated with the transponder. The base station typically uses On Off Keying (OOK) modulation to communicate to the transponder. This means the tank resonance is completely halted and then started up to transfer data via the magnetic link. The decoupling stage experiences large “step” responses as data is transmitted to the transponder. The tank can ramp up to its full resonant amplitude in 100 µs to 400µs depending on the drive system used. FIGURE 3: The system can be simplified as shown in Figure 3. The output of the peak detector can be simplified as the step response source with a 150V amplitude that also has the carrier and data signals superimposed on it as described earlier. The output response of the decoupling stage is given by Equation 2. This is also the input signal to the low-pass filter. EQUATION 2: It is useful to think in terms of τ (RC time constant) because the voltage across the resistor reduces by a factor of 0.368 as every τ second elapses. The exponential decay curve, for the voltage across R, is shown in Figure 4 and indicates that the initial voltage decays rapidly, but settles out slower as the voltage is reduced across the resistor. The system must be allowed to settle for a long enough period so that the step response voltage has reduced to a voltage that is smaller than the modulation voltage. The required value for RC, or τ, can be calculated using Equation 3, based on the following assumptions:  The system needs to be able to start LF communi- cations 200µs after the resonant tank has stabilized.  The decoupler should settle to at least half the data modulation voltage. HV-Env C R LP Filter V = 150e-t/τ τ = RC

latter solution is shown in Figure 5. is around 3 volts, after the tank has stabilized.

150 VDC , but if the calculation is repeated with an initial

voltage of 3 VDC , then the required τ comes to 25µs. Figure 6. The switch is closed to reduce received via the LF Talkback link.

DS00912A-page 6  2004 Microchip Technology Inc. THE LOW PASS FILTER STAGE The output signal from the decoupling stage consists of the 125 kHz carrier ripple and the modulated data signal, if one ignores the dynamic response signal. The carrier ripple is about 300 mV peak-to-peak. The data is 4 mV peak-to-peak with 6 dB of gain of the decoupler and a cutoff frequency at about 10 kHz. The aim of the low-pass filter stage is to amplify the data signal at 2.5 kHz and to filter out the carrier ripple in the most effective manner. The three most common active filter topologies used are the Chebyshev, Butterworth and Bessel filters. The Chebyshev filter has the steepest transition from pass band to stop band, but has ripple in the pass band. The Butterworth filters have the flattest pass band response, but does not have such a steep transition as the Chebyshev. The Bessel filter has a linear phase response with a smooth transition from pass to stop band. It seems the Chebyshev filter would best be suited for this application, but the frequency response does not tell the whole story. The data signal is amplitude modulated and the tank has steep transient response dynamics. As a result, the filter should have a stable and flat transient response. The Chebyshev filter has a very sharp frequency cutoff response, but has the worst transient response of the three filter topologies. The Chebyshev filter also has an underdamped step response with overshoot and ringing. The Butterworth filter has a better transient response, but still some overshoot. The Bessel filter has the worst response from a frequency perspective, but has the best transient response as a result of its linear phase characteristics. There are of course other active filter topologies such as elliptical, state variable, biquad and more, but a Bessel filter has adequate performance for the application. The data signal, in this example, has maximum modulation frequency of 2.5 kHz or a T E of 200µs. A Bessel filter, with a cutoff frequency of 1/(2.2TE) = 2.27 kHz, would be ideal from a noise rejection point of view, but a 2.5 kHz cutoff was chosen to minimize sym- bol overlap. The target is to design a filter with sufficient performance using a single operational amplifier in order to reduce the system cost. A dual operational amplifier can then be used because the decoupling stage also uses an amplifier. A third order Bessel filter can now be implemented with the remaining amplifier. The filter gain is the final aspect to specifying the Bessel filter. Using Microchip's FilterLab ® program, one can get the response for a unity gain – 2.5 kHz, 3d order Bessel filter. At 125 kHz, the filter has 93 dB of attenuation and the input ripple amplitude is 300 mV peak-to-peak. Assuming the filter should have an output ripple of no more then 1 mV peak-to-peak with 12 dB of headroom for noise, coupled through the supply line, then one needs at least 62 dB of attenua- tion. This leaves 31 dB of allowable gain from the third order filter. For the design, a gain of 20 or 26 dB was chosen, leaving some additional headroom for ripple rejection. The 3d order low-pass Bessel filter is shown in Figure 8 and has a Fc = 2.5 kHz and 26 dB of gain. Please note that the circuit shown in Figure 8 has a fairly high output impedance at the data rate, but the output of the filter will be driving a high-impedance load, and this is therefore acceptable. FIGURE 8: Input 4.87k 10 nF Output 78.7k 16.5k 150 pF 10 nF3.92k

 2004 Microchip Technology Inc. DS00912A-page 9 AN912 SYSTEM MODIFICATIONS The system can be modified to better suit the user's requirements. The first aspect is to change the Bessel filter for a different LF Talkback TE. The rule of thumb is to set the filter's 3 dB cutoff frequency to Fc = 1/(2*TE). The new values for the Bessel filter, with a 400µs TE, is given in Table 1. TABLE 1: In addition to changing the filter cutoff frequency for a T E of 400µs, it is possible to increase the gain up to 18 dB and still maintain the carrier rejection chosen. It is also possible to increase C6 to a 4.7 nF capacitor, but please note that this will increase the transient response period. Increasing C6 will not have a dramatic influence on the overall system performance and it is not recommended. LONGER TRANSIENT STABILIZING PERIOD The example system was designed with the require- ment that LF Talkback communications should be able to start 200µS after the resonant tank has stabilized. The tank itself takes 100µS to 400µs to stabilize sufficiently, depending on the drive mechanism. The example circuit should be able to start LF Talkback communications with 2 mV of data modulation after 350 µs to 450µs, from when the tank is turned. The exact period depends on the residual charge in the peak detector from previous transmissions. The system can be simplified and improved if the system allows for a longer transient stabilizing period before LF Talkback communications are initiated. The peak detector capacitor, C5, can be increased propor- tionally to the longer stabilizing period, but not by more than a factor of about 3, otherwise it can influence data modulation sensitivity. R1 and R2 tank resistors should be reduced if C5 is increased, but not proportionally, it will effect sensitivity. The combined value for R1 and R2 should be no less then 4 MΩ . Capacitor C6 can also be increased, but it will not have a dramatic performance increase. The biggest advan- tage of a longer transient stabilizing period is that bias resistor R3 can be increased. Increasing the value of R3 will result in a slower change in the signal at point “A”, which means the tank can be controlled more accurately during the transient period. INCREASED DATA SENSITIVITY Increasing the system’s sensitivity to the modulated data signal can increase the LF Talkback range. A solu- tion has been partly described in the previous section; increase T E from 200µs to 400µs and then increase the gain by up to 18 dB. The component values for a system with a 400µs T E, or a center frequency of 1.25 kHz, and a gain of 100, or a 14 dB increase, is described in Table 2 below. This approach decreases the dynamic range that may or may not be used depending on how well the transponder loads the resonant tank. TABLE 2: Another solution is to remove resistor R11 to get the maximum sensitivity from the comparator, but this will also increase noise in the data. Another quick solution is to increase the gain of the decoupler buffer by up to 10 dB and lower the decoupler cutoff frequency by about half the gain increase ratio. The existing design makes use of a 3d order Bessel filter. For improved noise reduction, increase the order of the filter and add more gain per stage. This would typically be done if a T E of 200µs or 100µs, is desired with more sensitivity than can be reliably obtained with the example system. DRIVE SYSTEM The example circuit uses a half-bridge driver based on the TC range of FET drivers from Microchip. To increase the transient response period of the tank, start the tank in Full-bridge mode until the desired tank amplitude is reached and the tank oscillation is main- tained in Half-bridge mode. This method is described in AN232 Low Frequency Magnetic Transmitter Design. CONCLUSION This LF Talkback Design application note can be used to implement a cost-effective system to be used in RFID, passive keyless entry and other bidirectional transponder based technologies. The example circuit can be used as a basis for further hardware and firm- ware development to suit the user's requirements. R6 = 3.57k R7 = 15.0k R8 = 71.5k R9 = 4.42k R10 = 5.62k C9 = C12 = 22 nF C11 = 330 pF R6 = 2.26k R7 = 10.5k R8 = 226k R9 = 4.42k R10 = 5.62k C9 = 33 nF C11 = 100 pF C12 = 22 nF

DS00912A-page 10  2004 Microchip Technology Inc. APPENDIX A: SCHEMATICS FIGURE 11: LF BASE STATION NOTES: Unless otherwise specified; Resistance values are in ohms. Resistors are 1% tolerance. Capacitance values are in uF. SMT resistors are size 1206 and 1/8W. Device names and numbers shown here are for reference only and may differ from the actual number. Items labeled with A are unpopulated. Items labeled with B are socketed and populated. HIGH-VOLTAGE SECTION +5V 0.1 uF

1 INPWM

A COARSE_ENV_IN 10 nF 400V P3476-ND A .200LS .200LS UF1005 TP2 1.0 nF 500V 1412PH-ND A 2.2 nF 500V A 4.99M 4.99M A 4.99K BIAS TP3 80.6K R22 4.99M VREF VREF 2 -IN +IN VDD VSS OUT U2:A MCP6002/SN C8 100 pF 162K +5V 0.1 uF TP4 3.92K +5V R23 49.9K VREF R24 49.9K 10 nF 16.5K VREF C10 0.1 uF 6 -IN +IN OUT 7 78.7K C11 150 pF U2:B MCP6002/SN 4.87K TP5 C12 10 nF R10 5.11K R11 4.99M ENV_IN ENV_OUT COARSE_ENV_OUT TP6 TP7

 2004 Microchip Technology Inc. DS00912A-page 11 AN912 FIGURE 12: LF BASE STATION (Continued) +5V 1 uF C21 C20 1 uF MAX232CPE VCC TX RX R14 R15 T1IN T2IN R1OUT R2OUT C1+ C1- C18 1 uF C17 1 uF GND T1OUT T2OUT R1IN R2IN C2+ C2- C19 1 uF DE-9S (FEM) RESET SW2 MOM-NO +5V R20 4.7K R21 470 MCLR +5V R19 4.7K R18 470 RB0 SW1 MOM-NO A J3 RBO RX TX RB5 RB6 BIAS C23 20 pF

20.0 MHz

0.1 uF +5V COARSE_ENV_IN ENV_IN OSC1 OSC2 BIAS RB5 RB6 RF_IN PIC16F648A/P RA1 RA0 OSC1 OSC2 VDD RB7 RB6 RB5 RB4 RA2 RA3 RA4/TO MCLR VSS RBO/INT RB1 RB2 RB3 U3B VREF ENV_OUT COARSE_ENV_OUT MCLR RB0 RX TX PWM +12V 4P-DIN 6 7 POWER DYNAMIC MDC-034 C13 560 uF 25V P11220-ND VR1 78L05 OUTIN GND C14 47 uF 10V P11180-ND D2 GRN POWER +5V R12 270 RF_IN R13 WIRE ANTENNA 6.8" 1C16 0.1 uF C15 10 uF 6.3V +5V RF_+V CC RF_GND DATA_IN RF_GND NC AF_+VCC AF_GND AF_+VCC TP DATA_OUT AF_+VCC RR8

433.92 MHz

+5V +5V R16 270 RED RB5 RB6 GRN R17 270

 2004 Microchip Technology Inc. DS00912A-page 13 AN912 FIGURE 15: TOP MASK 05-01 XXXX REV. A TOP MASK C21 C20 C18 C19 RS232 C17 R14 R15U5 J1 J2D2 R12 C13 VR1 C14 POWER C15 C16 R13 C22C23 C24 RESET R20 R21 R19RB0 R18 A1 D3R16 D4R17 RB0 RB1 RB2 RB5 RB6 RB7 FILTER TP7 TP6 TP5 R11 R10 U3 R9 C11 TP4 TP3 C12 R23R24 HIGH VOLTAGE SECTIONU1 C10 R22 R2 D1 TP2 TP1

DS00912A-page 14  2004 Microchip Technology Inc. NOTES:

 2004 Microchip Technology Inc. DS00912A-page 15 Information contained in this publication regarding device applications and the like is intended through suggestion only and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. No representation or warranty is given and no liability is assumed by Microchip Technology Incorporated with respect to the accuracy or use of such information, or infringement of patents or other intellectual property rights arising from such use or otherwise. Use of Microchip’s products as critical components in life support systems is not authorized except with express written approval by Microchip. No licenses are conveyed, implicitly or otherwise, under any intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, Accuron, dsPIC, KEE LOQ , MPLAB, PIC, PICmicro, PICSTART, PRO MATE and PowerSmart are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. AmpLab, FilterLab, microID, MXDEV, MXLAB, PICMASTER, SEEVAL, SmartShunt and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Application Maestro, dsPICDEM, dsPICDEM.net, dsPICworks, ECAN, ECONOMONITOR, FanSense, FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP , ICEPIC, microPort, Migratable Memory, MPASM, MPLIB, MPLINK, MPSIM, PICkit, PICDEM, PICDEM.net, PICtail, PowerCal, PowerInfo, PowerMate, PowerTool, rfLAB, rfPIC, Select Mode, SmartSensor, SmartTel and Total Endurance are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. Serialized Quick Turn Programming (SQTP) is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2004, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. Note the following details of the code protection feature on Microchip devices:  Microchip products meet the specification contained in their particular Microchip Data Sheet.  Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions.  There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip's Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property.  Microchip is willing to work with the customer who is concerned about the integrity of their code.  Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Microchip received ISO/TS-16949:2002 quality system certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona and Mountain View, California in October 2003. The Company’s quality system processes and procedures are for its PICmicro® 8-bit MCUs, KEE LOQ ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified.

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