AN4501 STMICROELECTRONICS | Alldatasheet
Document overview
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Technical content
Datasheet sections
- 1 Overview
- 2 Guideline for the selection of external components
- 2.1 Guideline
- 2.2 Application example
- 3 General considerations
- 3.1 Operating input voltage range
- 3.2 Output voltage and current
- 3.3 Switching frequency
- 4 Selection of the inductor
- 4.1 Converter Duty cycle
- 4.2 Inductor average current
- 4.3 Inductor peak-to-peak current ripple
- 4.4 Inductor current ripple ratio
- 4.5 Inductor peak current
- 4.6 Input current limiter
- 4.7 Limit of the continuous conduction mode
- 4.8 Inductor RMS current
- 5 Selection of the freewheeling diode
- 6 Selection of the output capacitor
- 6.1 Output voltage ripple
- 6.1.1 Contribution of the charge and discharge of the bulk capacitance
- 6.1.3 Total output voltage ripple
- 6.2 Output current ripple
- 6.3 Calculation of the output capacitor
- 7 Selection of the input capacitor
- 7.1 Input voltage ripples
components and provides full diagnostics and protection for enhanced system reliability. watchdog and a limp home input support safety relevant functions. Figure 1. Circuit schematics in boost configuration
1 Overview
This application note provides a guideline and an example of dimensioning of the power component around the boost controller using the L99LD01. This guideline takes into account the wide input voltage range of automotive applications and its implication on the device selection. In a second part, additional information on the component dimensioning and the details of the calculations are provided.
2 Guideline for the selection of external components
The reader can find the list of abbreviations in Table 4.
2.1 Guideline
- Calculate the minimum and maximum duty cycles DMIN and DMAX
- Verify that the operation is compatible with the converter minimum on-time (tON,MIN)
- Verify that DMAX does not exceed the converter duty cycle limitation Step 2: Selection of the inductor
- Calculate the maximum inductor DC current IL,MAX
- Calculate the minimum inductance to comply with the requested ratio inductance current ripple to DC current at the minimum input voltage (IL,PP@VINMIN / IL,MAX)
- Select the standardized inductor value
- Recalculate IL,PP@VINMIN with the selected standardized inductance value Step 3: Selection of the freewheel diode
- Calculate the minimum breakdown voltage and the required current capability Step 4: Selection of the output capacitor
- Choose a voltage capability, which is higher than the overvoltage protection
- Calculate the required output capacitance fulfilling the maximum output current ripple Step 4: Selection of the input capacitor
- Calculate the minimum capacitance fulfilling the required input voltage ripple Step 5: Selection of the switching MOSFET
- Choose a MOSFET and determine the corresponding switching losses (at VIN,MIN)
- Calculate the MOSFET RMS current
- Make an iterative calculation of the total power dissipation (at VIN,MIN)
- Verify that the maximum junction temperature does not exceed the MOSFET maximum rating.
2.2 Application example
Table 1. Example of application conditions and requirements
Figure 2. Application schematics Table 1. Example of application conditions and requirements (continued)
AN4501 Guideline for the selection of external components The maximal duty cycle of the L99LD01 is respected as well (datasheet parameter: Duty Cycle, 88% minimum specification). Step 2: Selection of the inductor Required inductance value: The maximum inductor DC current is given by: According to the definition of the inductor current ripple rMAX, we have: The minimum inductance value is given by: An inductance of 6.8µH can be selected, considering a tolerance of +/- 20%. Inductor maximum peak and RMS current DMAX 1 VINMIN η× VOUT IL,MAX IOUT 1D– MAX IL,PP@VINMIN,TARGET rMAX IL,MAX× 0.5 4.81× 2.41A== = LMIN VOUT DMAX 1D MAX–()×× 450 10⋅ 3 2.41 0.9×× IL,PEAK,MAX IOUT VOUT DMAX 1D MAX–()×× IL,PEAK,MAX
Figure 3. Evolution of IL, IL,PEAK, IL,PP with VIN The next standardized breakdown voltage for a Schottky diode is 60 V. is equal to the LED current (1.0 A).
2 IL,PP@VIN,MIN
AN4501 Guideline for the selection of external components Two 4.7 µF MLCC with a voltage capability of 50 V and with an ESR of 4 mΩ can be placed in parallel. The resulting ESR is 2 mΩ (ESRCOUT). The resulting maximum output current ripple due to the bulk capacitance is: The additional current ripple caused by the ESR is: which is indeed negligible compared to IOUT,PP,MAX. The total output current ripple is 62 mA, fulfilling the target. Step 5: Selection of the input capacitor We assume that MLCC with a very low ESR is used. The contribution of the ESR to the input voltage ripple can be neglected as well, compared to the contribution of the charge and discharge of the input capacitor itself. The maximum inductor peak-to-peak current I L,PP,MAX is: The minimum input capacitor is calculated by: Using two 4.7 µF, 50 V input capacitor with an ESR of 4mΩ (ESRCIN), the maximum input voltage ripple caused by the bulk capacitance is: COUT,MIN IOUT DMAX× COUT,MIN 0.075 450 10 3⋅× IOUT,PP,MAX IOUT DMAX× 9.4 10 6–⋅ 450 10 3⋅× ESR COUT IL,PEAK,MAX× IL,PP,MAX VOUT D50× 1D 50–()× CIN,MIN IL,PP,MAX 8 450 10 3 0.100×⋅× IL,PP,MAX
Guideline for the selection of external components AN4501 The additional input voltage ripple caused by the ESR is: which is indeed negligible compared to 70 mV. The sum of both contributions is below the maximum target of 100 mV (ΔVIN,MAX). Step 6: Selection of the switching MOSFET The minimum required breakdown voltage is the same as the breakdown voltage of the diode. A 60 V, 16 mΩ max @ 25°C is considered. Estimation of the switching losses The turn-on and turn-off times of the considered MOSFET with is estimated to 20 ns. The maximum switching losses are given by: Calculation of the max. RMS current The MOSFET’s maximum RMS current is estimated by: Iterative calculation of the total power losses and of the maximum junction temperature Considering at maximum ambient temperature of 85°C, a thermal coefficient of the RDSON of 0.006 K-1, 20 ns of rise and fall times and a thermal resistance of 25 K/W, an iterative calculation leads to a junction temperature of 125°C and a RDSON of 25.6 mΩ with a This confirms that the MOSFET can operate in the worst case conditions, without exceeding the maximum rating of the junction temperature (175°C in general). ESR CIN IL,PP,MAX× 21 0 3– 2.36×⋅ 4.8mV== PM1,SW VOUT ILM A X,× FSW× PM1,SW 26 4.81× 450 10 3⋅× IM1,RMS,MAX IOUT IL,PP,VINMIN IL,MAX 2 ×= IM1,RMS,MAX 1.0 2 + × 4.30A==
Table 2. Results of the iterative calculation of the worst case junction temperature Table 3. Formula for currents and voltages of power components
Figure 4. Typical waveforms of a boost converter in continuous current mode
AN4501 General considerations
3 General considerations
3.1 Operating input voltage range
The input voltage range of automotive applications is usually wide, stretching from the cold cranking (below 5 V, depending on the car makers) or warm cranking (~7 V) to the jump start (~24 V). It is important to consider the component current, voltage and power dissipation over the whole operating range and not only for the minimum, typical or maximum input voltages. As we will see, some parameters reach their maximum value at a duty cycle of 50% and not at the minimum or maximum input voltages (V INMIN, VINMAX).
3.2 Output voltage and current
A slight variation of the voltage applied to a LED string results in a large variation of its forward current. As the light output and the color of the LEDs vary with the current, the best control strategy is a constant current generator to keep a constant brightness and the color. The forward voltage of the LEDs (V FLED) depends on the LED type, the process, the current, the temperature etc… The boost output voltage (V OUT) is given by the formula: VOUT = NbLED x VFLED + VRSENSE + VRON,M2 Where NbLED is the number of LEDs in the string, VRSENSE and VRON,M2 are the voltage drop across RSENSE and M2 (see Figure 2). VSENSE and VRON,M2 can be neglected compared to NbLED * VF,LED, therefore we will consider: VOUT - NbLED * VF,LED The maximum VOUT leads to the highest peak current in the inductor, in the switching transistor and in the diode. Therefore, a worst case calculation of those parameters must consider V OUT,MAX. The output current (IOUT) is set by the choice of the sense resistor RSENSE and a specific SPI control register of the L99LD01. To simplify, we consider only the case where this SPI register is set at its default value: I OUT = 150 mV (typ.) / RSENSE
3.3 Switching frequency
The switching frequency FSW is a key parameter in the design of a DC-DC converter. Increasing the frequency allows in general the use of smaller capacitors and inductors, but as a drawback, it also leads to higher switching losses. Therefore the choice of the switching frequency is a tradeoff between costs, PCB area and efficiency. The L99LD01 uses a constant frequency architecture, designed to operate from 100 kHz to 500 kHz. The switching frequency is set by the resistor R SF as shown in Figure 5.
Figure 5. Converter switching frequency versus RSF value disturbance over a wide frequency range, resulting in a reduction of the peak emission. For clarity, in the rest of the document, we assume that this function is disabled.
4 Selection of the inductor
for its choice, as it dictates the cost and the overall performance of the system. size, its cost and the inductor current ripple.
- Smaller inductance current ripples
- Smaller input voltage and current ripples
- Smaller output voltage and current ripples
- Smaller current peaks in the converter switching MOSFET
- Smaller diode peak currents However, a larger inductance value means higher cost, larger PCB surface and in general slower response time to transients.
Figure 6. Inductor waveforms of a boost converter in CCM
4.1 Converter Duty cycle
Selection of the inductor AN4501 Equation 2: where η is the converter efficiency. The duty cycle is a decreasing function of VIN. In particular, the maximal duty cycle, noted DMAX, is reached for the minimum input voltage, VIN,MIN. The duty cycle range must be compliant with two device parameters with the device minimum duty cycle (parameter TON_MIN, maximum specification: 14%) and maximum duty cycle (parameter Duty Max, minimum specification: 88%)
4.2 Inductor average current
In steady state, the average current of the output capacitor over one period must be equal to zero. Since the inductor delivers current to the load only during the converter’s off-phase (see Figure 7), inductor current averaged during t OFF is equal to the output current: Extracting IL from this equation gives: Equation 3: We can see that IL is independent from the inductor value. Moreover, the worst case average inductor DC current is maximal for the maximal duty cycle. As a consequence, a worst case calculation of the inductor DC current must consider V IN,MIN. Equation 4: with Deffective D1 VIN η× VOUT IL tOFF IL IOUT ILM A X, IOUT DMAX 1 VIN MIN, η× VOUT,MAX
Figure 7. Inductor current flow during the converter on-time and off-time
4.3 Inductor peak-to-peak current ripple
Selection of the inductor AN4501 Equation 7: This shows that the maximum value of the inductor peak-to-peak current is reached for a duty cycle of D50 = 50%.
4.4 Inductor current ripple ratio
The inductor current ripple ratio r is defined as the ratio between the peak-to-peak current ripple and the average current. Equation 8: Increasing the value of the inductance, we reduce the inductor current ripple and the output voltage ripple, as we will see in the section Section 6.1.2. In general, the max allowed inductor current ripple ratio is optimal for a value in the range of 0.3 to 0.5, from the standpoint of the cost / current ripple. Indeed, reducing r to a value much lower than 0.3 leads a very large inductor size. Increasing r to a value which is much higher than 0.5, does not lead to a significant size reduction (see Appendix C: Document management). Therefore setting r to 0.4 or 0.5 is a good starting point. Once the maximal inductor peak-to-peak current is fixed, we can estimate the minimum required inductance value, using Equation 6 applied at V IN,MIN which corresponds to a duty cycle DMAX. Equation 9:
4.5 Inductor peak current
The inductor peak current must be calculated to make sure that in all cases, the inductor saturation current is not reached. The inductor peak current, ILPEAK is given by: dlLP P, VOUT 12 D–()× r ILP P, IL IL,PP,MAX rI LM A X,× r IOUT LMIN VOUT DMAX× 1D MAX–()×
AN4501 Selection of the inductor Equation 10: Using the expression of IL,PP from Equation 5 gives: Equation 11: As we will see, in most of the cases, the variation of IL,PEAK with the duty cycle is dictated by the term IOUT/(1 - D) if the IL,PP is lower than IL. Therefore, in general IL,PEAK MAX is reached at the DMAX (and VINMIN). This property can be verified by calculating the derived function of IL,PEAK: Equation 12: In general the ratio between the inductor current peak-to-peak current and the inductor average current is kept below 1. Therefore: Equation 13: Therefore: Under these conditions, therefore the peak current is a monotonically increasing function which reaches its maximum at D MAX (and VIN,MIN): Equation 14: IL,PEAK IL ILP P, IL,PEAK IOUT VOUT D1 D –()×× dIL,PEAK IOUT VOUT 12 D–()× VOUT D1 D –()×× IOUT VOUT D× IOUT dIL,PEAK IOUT VOUT D× VOUT VOUT dIL,PEAK IL,PEAK,MAX IOUT VOUT DMAX 1D MAX–()××
4.6 Input current limiter
Figure 8. A shunt resistor is used to monitor the inductor current during the on-state
4.7 Limit of the continuous conduction mode
works in CCM, in other words, inductor current does not decay to zero.
diode and of the switching MOSFET, when the inductor current is close to zero. CCM over the whole input voltage range. Figure 9. Inductor current at the boundary between CCM and DCM
- an operation in CCM at a duty cycle of 33% guarantees the CCM over the whole duty cycle range
- an inductance value higher than ensures the operation in CCM over the whole operating range ,/33,/ W ("1($'5 VOUT D1 D –()×× 2I× OUT LBOUNDARY VOUT D1 D –() 2×× dLBOUNDARY VOUT 1D–() 13 D–()×× LBOUNDARY 2V× OUT
Selection of the inductor AN4501
4.8 Inductor RMS current
The inductor RMS current (IL,RMS) is needed to calculate the inductor copper loss (power dissipation caused by resistance of the inductor wires, noted DCR). The waveform of the inductor current in CCM is a triangular signal with an average current of IL and a peak-to-peak current IL,PP (see Figure 4). IL,RMS is given by (see Section 8.7 for the details of the calculations): Equation 16: Similarly to IL,PEAK, IL,RMS is also in general reached at DMAX (and VIN,MIN) The maximum copper loss is PCOPPER,MAX = DCR x I2 L,RMS,MAX ILR M S, IL
2 ILP P,
IL RMS MAX,, ILM A X,
2 ILP PV INMIN(),
ILP PV INMIN(),
5 Selection of the freewheeling diode
thanks to their low forward voltage and their fast recovery time. Figure 10. Typical waveforms of the freewheeling diode in CCM the output current. However, the diode peak current is equal to the inductor’s peak current. The maximum rating of the Schottky diode must be chosen accordingly. higher in case of open load. output voltage until an over-voltage condition on the output is detected.
VOUT,OVTH must be set to a value which is higher than VLED,MAX. where OV_TH1 is typically 3.5 V (refer to datasheet of the L99LD01). Figure 11. Feedback resistors for output over-voltage detection
AN4501 Selection of the output capacitor
6 Selection of the output capacitor
The output capacitor determines the output voltage and current ripples. For a current source, the choice of the output capacitor begins with the specification of the maximum output current ripple IOUT,PP ,MAX. The capacitor voltage capability must be higher than the maximum output voltage. Note that some margin must be taken, as this parameter has a non-negligible tolerance and varies with the temperature and the applied DC voltage (if a MLCC capacitor is used).
6.1 Output voltage ripple
The main causes of the output voltage ripples are:
- The charge, respectively the discharge, of the ideal capacitor without equivalent series resistor (ESRCOUT) during tON and tOFF. The ideal capacitance is called the bulk capacitance in the rest of the document.
- The voltage drop caused by the output capacitor’s current ripple across ESRCOUT.
6.1.1 Contribution of the charge and discharge of the bulk capacitance
During the on-phase, the output current is exclusively delivered by the output capacitor COUT. During tON: iCOUT = -iOUT ~ -IOUT (seeFigure 12) As a first approximation, we assume that the output current ripples are negligible compared to the average value. This assumption is justified by the fact that the selection of the output capacitor should limit the conducted emission at the output in order to fulfill stringent specifications on the electromagnetic emissions. The integration of the output voltage over the on-phase gives:
6.1.2 Contribution of the output capa citor ESR to the output voltage ripple
Considering that:
- During tON, iCOUT = -iOUT ~ -IOUT and the diode is reverse biased: iD = 0
- During tOFF, the diode is conducting and charges COUT and delivers current to the LED strings: iCOUT ~ iD - IOUT We can conclude that at anytime, iCOUT ~ iD - IOUT and ICOUT,PP = ID,PP = IL,PEAK (see Figure 12). iCOUT COUT dVCOUT ΔVCOUT IOUT COUT IOUT D×
Figure 12. Typical current waveforms of the output capacitor current
6.1.3 Total output voltage ripple
6.2 Output current ripple
the dynamic resistance of the LED string). the dynamic resistance one LED at the considered output current.
AN4501 Selection of the output capacitor The resulting output current ripple is: Equation 21:
6.3 Calculation of the output capacitor
Ceramic capacitors are recommended. As they have a low ESR, we can first select the output capacitance, neglecting the contribution of ESRCOUT. The Equation 21 becomes: Extracting COUT yields: Equation 22: Once the output capacitor is chosen based on Equation 22, its ESR is known, and the contribution of the ESR to the output current ripple can be calculated, so that the assumption can be confirmed. IOUT PP, ΔVCOUT ΔVCOUT,ESR+ RDOUT IOUT D× × IOUT PP MAX,, IOUT DMAX× COUT MIN, IOUT DMAX×
7 Selection of the input capacitor
buffer the input voltage, for example in case of line transients. Figure 13. Current waveforms of the inductor and of the input capacitor for a boost
7.1 Input voltage ripples
discharge of the (ideal) input bulk capacitance and of the current ripple across the ESR.
7.1.1 Contribution bulk capacitance to the input voltage ripple
voltage increase during the charge of the capacitor (iCIN > 0). is represented by the blue area (see Figure 14).
Figure 14. Current waveforms of the input capacitor for a boost converter in CCM Similarly to IL,PP, VCIN,PP reaches its maximum when the duty cycle is 50%.
7.1.2 Contribution of the capacitor ESR to the input voltage ripple
Here again, the worst case corresponds to a duty cycle of 50%.
7.2 Maximum input voltage ripple
Selection of the switching MOSFET AN4501
8 Selection of the switching MOSFET
The MOSFET M1 is the main switching element of the boost converter. The most important parameters for its selection are:
- the breakdown voltage
- the peak and the RMS currents
- the RDSON
- the thermal resistance RTH-J-AMB
- the turn-on and the turn-off time Note: The gate driver of the L99LD01 controls M1 with a typ. voltage, V G1, of ~ 10V, provided that the supply voltage is high enough (10 V + Dropout of the VCC2 internal regulator ~ 10.2 V). If VIN is below ~10.2 V, VCC2 will be ~VIN – 0.2 V and so does VG1. Therefore, a logic level MOSFET is required to keep the converter’s performance at VIN below ~ 10.2 V.
8.1 Breakdown voltage
During tOFF the drain-source voltage of the MOSFET M1 is equal to VOUT + VF,DIODE. Some margin must be added in case of ringing at the switching node.
8.2 MOSFET peak current
During tON, the MOSFET M1 is turned on and the inductor current flows into M1 (see Figure 4 and Figure 7). Therefore, the MOSFET maximum peak current is equal to the inductor maximum peak current:
8.3 MOSFET power dissipation
The MOSFET’s power dissipation mainly comes from the the conduction losses and the switching losses. IM1 PEAK MAX,, IL PEAK MAX,, IOUT VOUT DMAX× 1D MAX–()×
8.3.1 Conduction losses
Figure 15. Typical current waveform of M1 in CCM
Selection of the switching MOSFET AN4501 The Rdson is temperature dependant: RON,M1 = RON,M1@T25°C x (1 + α (Tj - 25)) Where:
- RON,M1@T25°C is the RDSON at 25°C
- Tj is the junction temperature of M1
- α is the temperature coefficient, which is in general in the range of 6·10-3 K-1 In return the junction temperature depends on the power losses. Therefore, an iterative calculation is necessary for an accurate estimation of the conduction loss. Figure 18 displays the flowchart for the estimation of the MOSFET power losses and R DSON.
8.3.1 Switching losses of M1
Some care must be taken for the calculation of the switching losses. Often, the turn-on and the turn-off times which are specified in datasheets of the MOSFET are applicable for a resistive load under very specific conditions (current and drain-source voltage).They are not valid for the switching of inductive loads. Switching losses in the M1 MOSFET occur when the drain-source voltage is high, while the M1 current is not negligible. We can split the switching losses between the turn-on and the turn-off transitions: P M1,SWON and PM1,SWOFF. Switch-on phase During t1, the gate-source voltage of M1 (vM1,GS) ramps up to the gate-source threshold voltage, VM1,GS,TH. During this phase, there is no change in the drain-source voltage of M1 (VM1,DS) and M1 is not yet conducting. The inductor’s current (IL,VALLEY) still flows through the diode and there is no switching loss in M1 (see Figure 16). PM1 COND, RON M1, IOUT 2 DMAX××∼
Figure 16. Waveforms during the switch-on of M1 allows M1 to drive the whole inductor current (called VM1,GS,ILVALLEY). which represents the area of the red triangle on Figure 16.
AN4501 Selection of the switching MOSFET The related switching energy is estimated by: At the end of the t6 phase, vM1,GS reaches vM1,GS,TH. Therefore, the drain current of M1 is zero and the diode conducts the inductor current IL,PEAK. The switching loss in this interval is zero. To sum up, the switching losses during the turn-off of M1 are equal to the total switching energy during this phase, multiplied by the switching frequency. Equation 27: Where tM1,SWOFF = t5 + t6. Total power switching losses of M1 From Equation 26 and Equation 27, we can estimate the total switching losses of M1: Equation 28: If the switch-on and switch-off times do not significantly differ, we can approximate the term IL,VALLEY x tM1,SWON + IL,PEAK x tM1,SWOFF by IL (tM1,SWON + tM1,SWOFF): Equation 29: We see that the switching losses reach the maximum value at VIN,MIN, which corresponds to IL,MAX.
8.3.2 Iterative calculation of the junction temperature of M1
The temperature dependence of the Rdson requires an iterative calculation. We propose here a way how to proceed. To have a first approximation of the junction temperature of M1, we consider that the switching losses are temperature independent. The Figure 18 sums up the proposed procedure: VOUT IL PEAK,× t6× PM1,SWOFF VOUT ILP E A K,× tM1,SWOFF FSW×× PM1,SW PM1,SWON PM1,SWOFF+ == VOUT FSW× PM1,SW VOUT IL× FSW×
- Initial estimation of T J (noted TJ1). For example, we can use TAMB as a starting point.
- Calculation of the corresponding Rdson, using the Rdson at 25°C and the thermal
- Calculation of the conduction losses in M1
- Calculation of the total power losses in M1
- Calculation of the resulting junction temperature (noted T J2)
- Calculation of the R DSON corresponding to TJ2
- The calculation is finished if the difference between R DSON at TJ1 and TJ2 is smaller
substitute TJ1 by TJ2 and we restart a new iteration. Figure 18. Flowchart for the iterative calculation of the junction temperature of M1
Table 4. Notations and abbreviations
Table 4. Notations and abbreviations (continued)
8.4 Calculation of the duty cycle in CCM
Figure 19. Inductor voltage during the on-phase Figure 20. Inductor voltage during the off-phase
Figure 21. Typical inductor waveforms in continuous conduction mode
8.5 Calculation of the mosfet RMS current
Figure 22. Typical current waveform of M1 in CCM
8.6 Calculation of the freewheeling diode RMS current
the diode RMS current can be derived from Equation 30. diode, D must be replaced by 1 - D in Equation 30. Figure 23. Typical current waveform of the freewheeling diode in CCM
8.7 Calculation of the inductor RMS current
2 IL,PP
2 DI L
AN4501 Calculation details iL = iD and (iM1 = 0) for DT < t < T By definition, Using Equation 30 and Equation 31, we have: Equation 32: Equation 33: or Equation 34: IL,RMS 2 1 T--- iL 2 td T T--- iM1 2 td T T--- iD 2 td T + IM1,RMS
2 ID,RMS
2+== = ID,RMS 2 IL IL,RMS IL IL,PP IL,RMS IOUT IL,PP IL 2
Document management AN4501 Appendix C Document management 1. High efficiency constant current LED driver (L99LD01, DocID025319) 2. Switching power supplies A to Z. by S. Maniktala (ISBN13: 978-0-7506-7970-1) 3. Fundamental of Power Electronics by R. Erickson, D. Maksimovic (ISBN 0-7923-7270-0) 4. Switch-Mode Power Supplies by C. Basso (ISBN 978-0-07-150859-9)
Revision history
Table 5. Document revision history 29-May-2014 1 Initial release.