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Design Considerations for Switched Mode Power Supplies Using A Fairchild Power Switch (FPS) in a Flyback Converter www.fairchildsemi.com ©2002 Fairchild Semiconductor Corporation Introduction Flyback switched mode power supplies (SMPS) are among the most frequently used power circuits in household and consumer electronics. The basic function of an SMPS is to supply regulated power to the load on the secondary, or output side. An SMPS typically incorporates a power transformer, secondary-side rectifier diodes, a switching semiconductor device with control IC, and peripheral circuitry. If the level of integration of the switching and control circuitry is not high enough, then additional, separate circuits will be required to accommodate all functions. Such additional components raise the overall SMPS cost and not uncommonly reduce reliability. Fairchild Power Switches are highly-integrated ICs for power supply applications. They combine a high-voltage power MOSFET (SenseFET) and pulse width modulation (PWM) based control IC in one package. Moreover, they provide enhanced IC functionality, thereby minimizing the number of additional components needed in an SMPS. Fairchild Power Switch (SPS) ICs are widely used in the power circuits of a variety of equipment, such as color TVs, printers, PCs, monitors, battery chargers and ac adapters. They typically incorporate a variety of enhanced protection functions and they permit a much reduced power consumption in standby modes. This application note considers the three major functional blocks of an SMPS: Fairchild Power Switch (SPS) , flyback converter, and transformer. It discusses a variety of issues important to their design and use in the overall SMPS. Figure1. Internal block diagram of a Fairchild Power Switch(FPS). Source GND Drain Sense Feedback Soft Start & Sync. LEB Rsense 2.5R R OSC. Reset 1mA2uA Vck R S Q 6.3V Sync. 7.5V R S Q Reset Voffset #3 Vcc Voltage Ref.UVLO 32V OVP Control IC Sense FET Thermal Protection Rev. 1.0.2
- Block Diagram and Basic Operation
1.1 Block Diagram
block; protection circuits; and gate driving circuits.
1.2 Under Voltage Lockout (UVLO)
Figure 2. Detail of the undervoltage lockout (UVLO) thereby maintaining the SenseFET at turnoff. Figure 3. Fairchild Power Switch(FPS) control circuit status vs. Vcc.
1.3 Feedback Control Circuit
isolation is required. C fb improves the noise characteristics. SMPS secondary side error amplifiers. Figure 4. Fairchild Power Switch(FPS) feedback circuit appropriate for off line SMPS use (current mode PWM).
1.4 Example Fairchild Power Switch (FPS)
the LM431 regulator, and that of Figure 5b uses a Zener diode. regulation is relatively poor. fixed current to the Zener diode to stabilize its voltage.
Figure 5. Fairchild Power Switch(FPS) feedback control circuit.
1.5 Soft Start Operation
continues to charge to 5V through the 50kΩ resistor.
Figure 6. Soft start circuit.
1.6 Synchronization
the far left of the monitor display where it cannot be seen. s, cannot drop below 0.6V because of diode D sync. comparator output waveform, Vcomp of Figure 8. Ck in Figure 8, which goes low. Figure 7. Synchronization circuit. Figure 8. Synchronization circuit operation.
- Fairchild Power Switch(FPS) Built
(Auto Restart Mode protection).
2.1 Output Overload Protection
determines whether the overload is true or merely transient. Only a true overload will trigger the overload protection. the maximum input power is limited at any given voltage. Fairchild Power Switch (FPS) shuts down. Figure 9. Fairchild Power Switch(FPS) overload protection circuit. The shutdown delay interval is therefore determined by C fb. d and the Zener, thereby eliminating their added costs.
Figure 10. Long delay shutdown.
2.2 Output Short Circuit Protection
short circuit protection operates as follows. peak because the inductor’s magnetic core is unable to reset. secondary coils and the secondary side rectifiers. can be operated in either latch mode or auto restart mode. protection circuit operates, entering the latch mode.
4 D1 D2
Figure 11. Operation of the SMPS flyback converter’s output short circuit protection (latch mode ).
2.3 Fast Protection Without Delay
protection with no time delay. Figure 12. A fast protection circuit without a shutdown delay.
2.4 Overvoltage Protection
Switch (FPS) is unable to start switching. serious, a fire could start. connected to the secondary output could be destroyed. operation Vcc must be set below 25V .
- Noise Considerations at switch Turn On
3.1 SMPS Current Sensing
3.2 Current Sensing Waveform Noise After
capacitance; and (3) MOSFET gate operating current. current also flows through the current sensing resistor. Figure 13. Current sensing waveform noise after turn on.
©2002 Fairchild Semiconductor Corporation
3.3 Dealing With Leading Edge Noise
Among the measures taken to reduce leading edge noise, the most commonly used technique is the RC filter. As shown in Figure 14a the RC filter is effective against the noise, but it has the disadvantage that it distorts the current sensing signal so that accurate current sensing becomes difficult. Furthermore, a large RC value may be difficult to implement on an IC and may even require a bigger chip. The technique of leading edge blanking, as presented in Figure 14b and 14c, overcomes the distortion disadvantage of the RC technique and works as follows. Since the problem noise arises just after turn on, if a circuit is inserted that ignores the current sensing line for a fixed time just after turn on, operation can continue normally regardless of the noise. Whatever the details of the location and type of circuit used, the basic idea is to maintain a minimum turn on time i.e., touse the shortest turn on time that cannot be terminated once turn on starts. Duty ratio control with a minimum turn on time is implemented through a non linear control method having a very wide control range relative to a linear control. The non linear control operates such that if load conditions require a turn on time of 400ns when the minimum turn on time is set at 500ns, then one switching cycle will turn on at 800ns. The next cycle will be missed, ensuring that the average turn on time is 400ns. In this case every other cycle is missed. This is pulse skipping. In this case the switching frequency will be half that of a linearly controlled system, thereby improving SMPS efficiency at light loads. The input power is therefore minimised. The Fairchild KA34063 dc/dc converter is an example of a non linear control IC.
3.3.1 Burst Mode Operation
The aforementioned method can be viewed as an example of burst mode operation. Burst mode operation, by reducing the switching frequency, is one of the most useful ways to improve SMPS efficiency at light loads and to reduce the standby input power of household appliances, etc. Note that burst mode operation is not a burst oscillation (as in ringing choke conversion circuits), which can bring about reliability problems. There are mainly two types of true burst mode operation: one type lowers the switching frequency equally. The other switches at normal frequency for a fixed time and stops the control IC operation for a large number of cycles. Even though in the first method the control IC continues to consume power, the output voltage ripple is minimized. The second method can be a useful way to reduce the minimum input power at standby (since obviously the standby power is greatly reduced when the IC is stopped). Indeed, it is often used in cell phones to reduce the dc/dc converter’s power consumption in standby mode. However, it has the disadvantage of a larger output voltage ripple. Currently, Europe restricts a household appliance’s standby input power to less than 5W, and in time it will be required to be less than 3W. For such needs, burst mode operation will be a powerful method to satisfy the requirement for reduced standby input power.
Figure 14. Leading edge blanking.
- Flyback Converter Operation
4.1 Operation In Continuous Conduction Mode
waveforms of CCM operation, which operates as follows.
4.1.1 For t0 ~ t1 = TON
power source supplies energy to Lm while the MOSFET is on. that has to supply the output current during this interval.
4.1.2 For t1 ~ t2 = TOFF
4.1.3 Relationship Between Input And Output
Figure 15. A typical flyback converter. Figure 16. Flyback converter operating waveforms in continuous current mode (CCM).
4.2 Discontinuous Conduction Mode (DCM)
output rectifier diode is actually conducting. Figure 17. Flyback converter operating waveforms in discontinuous current mode (DCM). average inductor (or transformer) voltage is always zero. where fsw is the switching frequency.
4.3 Flyback Converter Design
4.3.1 Turns Ratio Considerations
conditions change as listed immediately below. cycle to deliver equal average current reduces efficiency. secondary side outputs, it is advantageous to increase n.
©2002 Fairchild Semiconductor Corporation
4.3.2 Deciding On The Operating Current Mode
As discussed in Sections 4.1 and 4.2, there are two different operating current modes possible in a flyback converter: the continuous conduction mode (CCM); and the discontinuous conduction mode (DCM). Here the advantages and disadvantages of each are reviewed, to help the designer make a proper choice between them.
4.3.2.1 Characteristics Of The Discontinuous
In a flyback converter design, if discontinuous conduction occurs just at minimum input voltage and maximum output power, then discontinuous conduction must be considered to be the case for all input conditions. The flyback converter’s input power in discontinuous conduction mode can be expressed as: Regardless of any changes in input voltage, the power equation indicates that the input current is limited by the peak value of the current flowing through the MOSFET in the transformer primary. A Fairchild Power Switch (FPS) has an integrated overcurrent protection feature (see Section 2.1, above). This feature does not require external components and operates across the range of input current. However, the fixed operating current of DCM, tends somewhat to offset the effect of the larger effective primary side current. The gain is at the low frequency end where core loss is not a problem since only a minimum number of turns need be wound. Also, turn on loss is not a serious problem due to the low input current. Other losses such as eddy current, skin effect, proximity effect, etc., are not significant. A more clearly defined advantage of DCM operation is that it permits the use of a slow and hence, low cost secondary rectifier diode. In contrast to the continuous conduction mode, in the discontinuous conduction mode the effective current is higher, requiring the use of heavier wire and hence thicker coils. Therefore DCM does not bring an advantage insofar as transformer construction is concerned. Moreover, DCM causes the MOSFET operating temperature to increase because of the large effective primary side current, as was described above (Section 4.3.1).
4.3.2.2 Characteristics Of The
Continuous Conduction Mode Since the coils’ effective current is decreased CCM brings the advantage of lighter wire. The smaller effective current also reduces MOSFET heating. This is a definite advantage for average input current. On the other hand, CCM operation brings with it a need to consider the rectifier diode’s reverse recovery current. Depending on the diode’s reverse recovery time (t rr), the reverse recovery current may stress the diode and increase the loss at its end terminals. It is therefore necessary to use a diode with the minimum t rr possible within the allowable cost range. PI 2---LmIP 2 fsw=
Figure 18. The current and voltage ratings required on the primary side switching device and the secondary side rectifier diode depend on the turns ratio (n) selected.
4.3.2.3 Designer’s Choice
through the primary side switch (MOSFET) current.
5.1 Why a Transformer Is Needed
19b) switching at 50kHz is used to obtain 5V from 100V . Figure 19. Why a transformer is needed.
5.2 The Ideal Transformer
2.The coil loss is zero (the device has no losses). 3.The inductance of each coil is infinite.
Figure 20. Ideal transformer. tion is such that it enters on one side and leaves at another. mary side also goes to zero.
5.3 The Real Transformer
- The coupling coefficient between each coil is finite, and
- There are losses, such as iron (hysteresis) loss, eddy current
- The inductance of each coil is finite. When a gap is
placed in the core the inductance becomes still smaller. an actual transformer and shows the magnetizing inductance. Figure 21. A model of an actual transformer showing the Figure 22. A more complete equivalent circuit of an
©2002 Fairchild Semiconductor Corporation 6. Transformer Design
6.1 Core Selection
The maximum power that a transformer core can deliver and the maximum energy a transformer inductor can store depends on the shape and size of the core. In general, as the effective cross sectional area (Ae) increases, more power can be delivered. Also, as the window area (Aw) on which the coils are wound increases, more and thicker windings can be used, allowing a further increase in the power that can be delivered. The product of Aw and Ae is called the area product, AP, and the maximum power a transformer can deliver is proportional to an exponential power of AP. Indeed, recent transformer theory shows designs depending almost entirely on AP. In the broader view, a flyback converter transformer can be viewed as a coupled inductor, so it's common to design a flyback transformer using inductor design methods. The two equations below, (a) and (b), represent two ways to calculate AP. Equation (a) below, is a method based on whether or not the core is saturated, is appropriate at low operating frequencies. Equation (b), limited by core loss, is appropriate at high frequencies. For any given design, it is necessary to calculate AP using both equations, and the equation that gives the higher value is the one that must be considered correct. Equation (a) assumes that all losses are wire losses and ignores the core (iron) loss. L and B MAX are in Henries and Tesla units, respectively, and K is listed in Table 1, below. From the above equation the current density (J) per unit area of wire is obtained from the current by the relationship below, which assumes that the temperature of the inductor’s “hot spot” is 30°C above ambient. At any operating frequency high enough so that the core losses become large, the following equation should be used. Specifically, it assumes that the total transformer losses are split equally (50/50) between the wire and the core In equation (b), KH is the hysteresis coefficient (typically 4 x 10-5 for ferrite cores) and K E is the eddy current coefficient (typically 4 x 10-10 for ferrite cores). The current density relationship here, represented by the equation for J 30, below, assumes a hot spot temperature of 15°C above ambient, with the iron loss adding on an additional 15°C (again, a total of 30°C above ambient). The parameter K in equation (a) is the product of the window utilization factor K U with the primary area factor KP. (See Table 1.) K U is the ratio of the cross sectional area of the winding’s copper to the entire window area, and it is significant in setting the isolation between the primary and secondary sides. Because it is related to the transformer shape and winding method, the designer should know the value of K U for the transformer usually used. The value of K U can vary greatly, depending particularly on how closely the isolation safety standards (re isolation) are followed; the K U of Table 1 assumes a general bobbin is used. K P is the ratio of the area of the primary winding to that of the total winding. In Table 1, it is unity for the inductors because a buck boost inductor has no secondary windings. For the flyback transformer coupled inductor, K P is usually 0.5, as Table 1 shows; such an inductor has the highest efficiency when the primary and secondary winding areas are equal. When there are more secondary windings, however, the K P for a flyback transformer coupled inductor can be lower than 0.5. Note that the ease and speed of obtaining an accurate AP from equations (a) and (b) depends on the designer's experience. A reasonably good knowledge of the probable values of the three parameters K U, KP, and J for the flyback transformer being designed will reduce the number of trial and error attempts necessary. AP LI PIRMS 104 420KB MAX 1.31 cm2[] a) = L = Inductance of Transformer Ip = Operating peak current Bmax = MAximum operating flux density Irms = RMS current J30 420 AP 0.24– Ac m 2⁄[]= AP L∆ ImIRMS 104 1.58 KHfSW KE fsw() 2+() 0.66 cm2[] b )= L = Inductance of Transformer Ip = Operating peak current Irms = RMS current J30 297AP 0.24– Ac m 2⁄[]=
6.2 Determining The Number of Turns
6.3 The Windings
determining the coil cross sectional area.
6.4 Determining The Gap
It is not easy to precisely calculate the required gap.
- Transformer and Inductor Design Handbook. 2nd ed.
- Switch Mode Power Supply Handbook. Keith H.
Billings. McGraw-Hill, Inc., 1989. Table 1. KU, KP, and K
©2002 Fairchild Semiconductor Corporation
3/19/04 0.0m 002 2002 Fairchild Semiconductor Corporation DISCLAIMER FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION, OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS. LIFE SUPPORT POLICY FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICODUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which (a) are intended for surgical implant into the body, or (b) support or sustain life, or (c) whose failure to per-form when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in significant injury to the user. 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. www.fairchildsemi.com