AN2252 STMICROELECTRONICS | Alldatasheet

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Technical content

Datasheet sections

  • 1 Theory of ESBT and quasi-resonant operation
  • 2 Application circuit description
  • 2.1 Operating conditions
  • 2.2 Circuit operation
  • 2.2.1 Bill of materials
  • 2.2.2 Transformer design
  • 2.3 PCB Layout
  • 2.3.1 Evaluation and measurements
  • 3 Conclusion
  • 4 References
  • 5 Revision history

Zero-voltage switching and emitter-switched bipolar transistor in a 3-phase auxiliary power supply Introduction The flyback converter is a popular choice in applications where the required power is normally less than 200W. The main reasons explaining its popularity are its simplicity, low cost and high efficiency for a small number of active components. In switching converters power loss is caused by power dissipation within the parasitic elements of both passive and active components. Power loss in passive components can be reduced by selecting suitable passive components and carefully designing the transformer. Power loss in active components can be improved by selecting suitable active components and making sure that they are used correctly. Power loss generated by active components can be divided into two categories:

  • conduction loss
  • switching loss The aim of the proposed zero-voltage switching control is to reduce switching loss (in this application, the primary switch turn-on loss). The zero-voltage switching control also greatly reduces the EMI generated by primary switch turn-on. Conduction loss is generated with the device fully turned-on, by the voltage drop across the conducting device. The proposed use of the Emitter-Switched Bipolar Transistor (ESBT) as the primary switch reduces conduction loss efficiently. Moreover, with its low saturation voltage and fast switching capability compared to an IGBT or a bipolar junction transistor (BJT), the ESBT is well suited for this use. These characteristics are essential in applications where a high breakdown voltage capability is required. The reference board presented in this Application Note gives a solution of a power supply for 3-phase applications like inverters for induction motors, welding machines, UPS etc. Very commonly in this kind of applications, the neutral line is not available or its use is not allowed, and only phase-to-phase voltage is available. The nominal European phase-to- phase voltage is 400VAC. Taking into account a ±20% tolerance, the rectified input bulk capacitor voltage can reach up to 680VDC. The zero-voltage switching topology requires a reflected flyback voltage equal to the input bulk capacitor voltage. For this reason it is necessary to use a switch which will accept at least 1500V and exhibits a low conduction loss during the ON-time. The high-voltage MOSFET switches rated for this voltage, available on the market today are rather expensive due to their large die size. The ESBT, thanks to its low voltage drop, high speed, square reverse bias safe operating area, smaller die size and lower price is well suited for use as a high-voltage power switch.

Theory of ESBT and quasi-resonant operation AN2252

1 Theory of ESBT and quasi-resonant operation

As mentioned in Introduction, the application studied in this Application Note implements zerovoltage switching (ZVS). This principle of operation is also known as quasi-resonant or valley switching. These names come from the waveform shape of the voltage across the primary side switch during or just before switch turn-on. Figure 1. shows the switch voltage, which is the sum of VIN, the DC bulk capacitor voltage, and Vflyback, the reflected voltage across the primary winding. The winding voltage depends on the state of the switch and the amount of magnetizing energy stored in the magnetic circuit of the transformer. One switching period can be divided into three basic areas determined by the state of the primary switch and the output diode conduction: the ON time, the OFF time and the DEAD time areas.

  • The “ON time” area corresponds to the time during which the primary switch is on and the transformerfs magnetizing inductance stores energy.
  • During the “OFF time”, the primary switch is off and the magnetizing inductance energy is discharged through the conducting output diode to the output capacitor. A ringing voltage of amplitude V spike also occurs during this phase. It is generated by the layout- related track inductance and by the leakage inductance created by the imperfect magnetic field coupling between the transformer primary and secondary windings. The ringing voltage amplitude is controlled and limited by a clamp circuit.
  • The “DEAD time” starts once all the stored magnetizing inductance energy has been discharged to the output capacitor. It is called “DEAD time” because neither the primary switch nor the output diode is conducting. So there is no energy transfer between the primary side and the secondary side. The primary winding voltage during this phase is resonating and has a cosine waveform starting from a voltage equal to the OFF-time plateau voltage. The DEAD time is used for the initiation of the next switching cycle, only there is no energy conversion, which is why this concept is called QUASI resonant, in comparison with pure resonant converters where the resonance of the primary current or voltage is the means of energy conversion. The voltage waveform has a negative slope and approaches or can even cross zero. The suitable moment to turn the primary switch on again is when the voltage across the primary switch is lowest. The shape of the switch voltage waveform at this point evokes a valley. This is why quasi-resonant or zero-voltage switching is also called valley switching. The resonance frequency during the DEAD time is determined by the magnetizing inductance and parasitic capacitance. The parasitic capacitance consists of the primary switch capacitance, the transformer winding capacitance, the inter-winding capacitance, the capacitances of the diodes located in the secondary, auxiliary and clamp circuits transformed to the primary side. The PCB tracks also generate some parasitic capacitance depending on the layout.

pin is usually connected to the application ground. AN1889 (see Section 4), available from the STMicroelectronics website: www.st.com.

2 Application circuit description

2.1 Operating conditions

2.2 Circuit operation

Figure 5. shows the schematic of the power supply. The power supply can be supplied from either an AC or a DC voltage source. by a voltage doubler consisting of diodes D2 and D3, and of capacitors C1 and C2. Table 1. Input/Output specifications

inductance generates ringing on the auxiliary winding voltage. Figure 5. Schematic diagram

The VCC capacitors are charged during the start-up phase by the permanent current source made up of resistors R1, R2 and R3. Thanks to the very low current consumption of L6565 during the start-up phase, the start-up current is in the order of hundreds of microamperes which significantly reduces the power dissipated in the start-up resistors. The complete demagnetization of the transformer core is detected by the auxiliary winding voltage crossing zero. Resistor R6 delivers this information to the IC's internal zero crossing detector through pin 5. Resistor R7 shifts the zero-crossing detector threshold towards a value closer to zero for reliable zero crossing during converter start-up or under overload conditions. Capacitor C8 delays the power switch turn-on to the moment when the collector voltage reaches a valley point. The primary current control circuit consists of current-sense resistors R11, R12 and low-pass filter R9, C7 connected to the CS pin 4 of the control IC. The primary power switch is the STC04IE170HV. It is an ESBT rated for a maximum current of 5A and collector-to-source voltage of 1500V. The gate of the ESBT is driven directly by the internal gate driver of U1 through pin 7. The ESBT also requires a bias current for the base of the internal BJT. It is provided by current transformer T2 through the diode D7. During the storage time, the collector current flows through the B-C junction for the time required by the junction to recover from conduction. The collector current flows then through capacitor C10 which stores the energy that will generate the initial base current spike necessary for the next switching cycle. The value of this current spike is determined by the voltage across capacitor C10 (which is limited by Zener diode D9), by resistor R10 and by the resistance of the B-E junction of the internal BJT of the ESBT. Diode D6 and resistor R4 provide the bias current required to precharge C10 during the first switching cycle and properly start the converter operation. Since the current transformer operation may be affected by core saturation when the voltsecond product exceeds the limit, a protection circuit consisting of R8, C9, D8 and Q2 is inserted in the current-sense path. This circuit is a timer which watches the maximum ON time. If the latter goes beyond a certain limit, the current-sense voltage is suddenly increased to its maximum threshold, thus stopping the gate driver and turning off the ESBT through the gate. Without this circuit, the current transformer core saturation would cause the ESBT to be unsafely turned off whenever there is a lack of base current. This condition may happen in case of an undervoltage at the input, for instance if a mains voltage drop occurs or the power supply is unplugged. As a consequence, the ON-time would be increased above the specified current transformer volt-second product limit. Clamp circuit D10, D11, C3, R13 and R14 protects the ESBT switch from the voltage spikes induced by the transformer leakage inductance. The output voltage is controlled by an opto-isolated feedback loop consisting of U2, voltage divider R18, R19, R20 and frequency response compensation components R17, C13 and C14. Since most of the voltage stress was moved (by the increased flyback voltage provided by an appropriate transformer's turns ratio) to the primary side, a 100V Schottky diode can be used as a rectifier on the secondary side even if the nominal output voltage is 24V. This is one of the advantages of using the quasi-resonant mode, which further helps decrease the output rectifier loss and increases the overall converter power efficiency.

2.2.1 Bill of materials

The list of components required to build the demonstration board is shown in Table 2. Most of the used active components are available from STMicroelectronics. Thanks to the outstanding performance of the ESBT, the switch does not require any heat sink for this

supplied by VOGT Electronic Components GmbH. Table 2. Bill of materials

2.2.2 Transformer design

  • Minimum input voltage 320VAC = 450VDC
  • Minimum switching frequency at full load and minimum input voltage 50kHz
  • Reflected flyback voltage 500V
  • Converter efficiency at full load and minimum input voltage 90% The initial power transformer design was further optimized by VOGT Electronic Components GmbH to cope with such a high voltage. Special attention was paid to guarantee VDE distances by padding the winding ends. To improve magnetic coupling, not only the primary winding but also the secondary was split. The two primary windings so obtained have half the total number of turns each and are connected in series while the two secondary windings have the nominal number of turns and are connected in parallel. The windings are laid out as follows starting from the winding nearest the core: W1 (Primary 1), W2 (Secondary 1), W3 (Primary 2), W4 (Secondary 2) and W5 (Auxiliary). The transformer is designed so as to comply with the EN60950 safety standard for CE certification. If the transformer must be compliant with the UL standard – on the flammability of isolation material – whose compliance is mandatory for applications implemented in the USA, then some modifications are required. The transformer’s physical appearance, dimensions and winding arrangement are shown in Figure 6.

Figure 6. Power transformer dimensions and winding arrangement

secondary windings are made of Litz wire. Table 5. The physical appearance, dimensions and winding arrangement of the current transformer are shown in Figure 7. Figure 7. Current transformer dimensions and winding arrangement Table 3. Power transformer core parameters Table 4. Power transformer winding parameters

2.3 PCB Layout

Figure 8. Assembly schematic Figure 9. represents the PCB layout of the copper connections. The holes for through hole Table 5. Current transformer core parameters Table 6. Current transformer winding parameters

Figure 9. PCB layout Figure 10. shows the converter. Figure 10. Picture of the converter

2.3.1 Evaluation and measurements

conduction loss prevail over the turn-on loss.

3 Conclusion

be used as a primary switch. the board and the power supply was not tested for EMC compliance.

4 References

  • AN1699, “Efficient driving network for ESBT to reduce the dynamic VCESAT and enhance the switching performances”
  • AN1889, “ESBT STC03DE170 IN 3-PHASE AUXILIARY POWER SUPPL Y”

5 Revision history

Table 7. Temperatures of the power switches at full load Table 8. Revision history