ADP3050

ADP3050 (Rev. C)

Document overview

• Manufacturer or author: Analog Devices, Inc.
• PDF pages: 20

Technical content

200 kHz, 1 A High Voltage Step-Down Switching Regulator Data Sheet ADP3050 Rev. C Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2008–2012 Analog Devices, Inc. All rights reserved.

FEATURES

Wide input voltage range: 3.6 V to 30 V Adjustable and fixed (3.3 V, 5 V) output options Integrated 1 A power switch Uses small surface-mount components Cycle-by-cycle current limiting Peak input voltage (100 ms): 60 V Configurable as a buck, buck-boost, and SEPIC regulator Available in 8-lead SOIC package Supported by ADIsimPower™ design tool

APPLICATIONS

PC peripheral power systems Preregulator for linear regulators Distributed power systems Automotive systems Battery chargers FUNCTIONAL BLOCK DIAGRAM 200kHz OSCILLATOR FREQUENCY AND CURRENT LIMIT FOLDBACK 2.50V REGULATOR CURRENT SENSE AMPLIFIER BOOSTFB SD GND BIAS CURRENT LIMIT gm 1.2V CMP R S Q DRIVER SWITCH IN COMP ADP3050 00125-001 Figure 1. GENERAL DESCRIPTION The ADP3050 is a current mode monolithic buck (step down) PWM switching regulator that contains a high current 1 A power switch and all control, logic, and protection functions. It uses a unique compensation scheme allowing the use of any type of output capacitor (tantalum, ceramic, electrolytic, OS-CON). Unlike some buck regulators, the design is not restricted to using a specific type of output capacitor or ESR value. A special boosted drive stage is used to saturate the NPN power switch, providing a system efficiency higher than conventional bipolar buck switchers. Further efficiency improvements are obtained by using the low voltage regulated output to provide the internal operating current of the device. A high switching frequency allows the use of small external surface-mount compo- nents. A wide variety of standard off-the-shelf devices can be used, providing a great deal of design flexibility. A complete regulator design requires only a few external components. The ADP3050 includes a shutdown input that places the device in a low power mode, reducing the total supply current to under 20 µA. Internal protection features include thermal shutdown circuitry and a cycle-by-cycle current limit for the power switch to provide complete device protection under fault conditions. The ADP3050 provides excellent line and load regulation, maintaining typically less than ±3% output voltage accuracy over temperature and under all input voltage and output current conditions. The ADP3050 is specified over the industrial temperature range of −40°C to +85°C and is available in a thermally enhanced 8-lead (not Pb-free only) SOIC package and a standard 8-lead (Pb-free only) RoHS-compliant SOIC package.

Rev. C | Page 2 of 20 TABLE OF CONTENTS

REVISION HISTORY

6/12—Rev. B to Rev. C 3/08—Rev. A to Rev. B Changes to Boosted Drive Stage Section and Thermal

Rev. C | Page 3 of 20 SPECIFICATIONS VIN = 10 V , TA = −40°C to +85°C, unless otherwise noted. Table 1. Parameter1 Symbol Conditions Min Typ Max Unit FEEDBACK Feedback Voltage VFB Over line and temperature ADP3050 1.16 1.20 1.24 V ADP3050-3.3 3.20 3.30 3.40 V ADP3050-5 4.85 5.00 5.15 V Line Regulation VIN = 10 V to 30 V, no load 0.005 %/V Load Regulation ILOAD = 100 mA to 1 A, ADP3050AR only −1.0 +0.1 +1.0 %/A Input Bias Current IFB ADP3050AR only 0.65 2 μA ERROR AMPLIFIER Transconductance2 gm 1250 μMho Voltage Gain2 AVOL 300 V/V Output Current ADP3050 COMP = 1.0 V, FB = 1.1 V to 1.3 V ±115 μA ADP3050-3.3 COMP = 1.0 V, FB = 3.0 V to 3.6 V ±120 μA ADP3050-5 COMP = 1.0 V, FB = 4.5 V to 5.5 V ±135 μA OSCILLATOR Oscillator Frequency3 fOSC 170 200 240 kHz Minimum Duty Cycle DMIN 10 % Maximum Duty Cycle DMAX 90 % SWITCH Average Output Current Limit4 ICL(AVG) ADP3050 BOOST = 15 V, FB = 1.1 V 1.0 1.25 1.5 A ADP3050-5 BOOST = 15 V, FB = 4.5 V 1.0 1.25 1.5 A Peak Switch Current Limit5 ICL(PEAK) 1.5 1.7 2.1 A Saturation Voltage BOOST = 15 V, ILOAD = 1 A 0.65 0.95 V Leakage Current 50 nA SHUTDOWN Input Voltage Low 0.4 V Input Voltage High 2.0 V SUPPLY Input Voltage Range6 VIN 3.6 30 V Minimum BIAS Voltage VBIAS 3.0 V Minimum BOOST Voltage VBOOST 3.0 V IN Supply Current IQ Normal Mode BIAS = 5.0 V 0.7 1.5 mA Shutdown Mode SD = 0 V, VIN ≤ 30 V 15 40 μA BIAS Supply Current IBIAS BIAS = 5.0 V 4.0 6.0 mA BOOST Supply Current IBOOST BOOST = 15 V, ISW = 0.5 A 18 mA BOOST = 15 V, ISW = 1.0 A 20 40 mA 1 All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). 2 Transconductance and voltage gain measurements refer to the internal amplifier without the voltage divider. To calculate the transcondu ctance and gain of the fixed voltage parts, divide the values shown by FB/1.20. 3 The switching frequency is reduced when the feedback pin is lower than 0.8 × FB. 4 See Figure 24 for typical application circuit. 5 Switch current limit is measured with no diode, no inductor, and no output capacitor. 6 Minimum input voltage is not measured directly, but is guaranteed by other tests. The actual minimum input voltage needed to keep the output in regulation depends on output voltage and load current.

Rev. C | Page 4 of 20 ABSOLUTE MAXIMUM RATINGS Table 2. Parameter Rating IN Voltage Continuous −0.3 V to +40 V Peak (<100 ms) −0.3 V to +60 V BOOST Voltage Continuous −0.3 V to +45 V Peak (<100 ms) −0.3 V to +65 V SD, BIAS Voltage −0.3 V to IN + 0.3 V FB Voltage −0.3 V to +8 V COMP Voltage −0.3 V to IN + 0.3 V SWITCH Voltage −0.3 V to IN + 0.3 V Operating Ambient Temperature Range −40°C to +85°C Operating Junction Temperature Range −40°C to +125°C Storage Temperature Range −65°C to +150°C θJA (4-Layer PCB)1 60.6°C/W θJA (4-Layer PCB)2 87.5°C/W Lead Temperature (Soldering, 60 sec) 300°C 1 Applied to all models that are not Pb-free. 2 Applied to all Pb-free models. Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ESD CAUTION

Figure 2. Pin Configuration Table 3. Pin Function Descriptions VIN and approximately −0.5 V. 6 SD Shutdown Pin. Use this pin to turn the device on and off. If this feature is not needed, tie this pin directly to IN. 7 GND Ground Pin. Connect this pin to local ground plane. this pin to ensure proper regulator operation.

Figure 15. Switch Saturation Voltage vs. Load Current

190 SWITCHING FREQUENCY (kHz)

Figure 16. Switching Frequency vs. Temperature Figure 17. Frequency Foldback Figure 18. Continuous Conduction Mode Waveforms Figure 19. Discontinuous Conduction Mode Waveforms Figure 20. Transient Response

Rev. C | Page 11 of 20 APPLICATIONS INFORMATION ADIsimPower DESIGN TOOL The ADP3050 is supported by the ADIsimPower design tool set. ADIsimPower is a collection of tools that produce complete power designs optimized for a specific design goal. The tools enable the user to generate a full schematic, bill of materials, and calculate performance in minutes. ADIsimPower can optimize designs for cost, area, efficiency, and parts count while taking into considera- tion the operating conditions and limitations of the IC and all real external components. For more information about ADIsimPower design tools, refer to www.analog.com/ADIsimPower. The tool set is available from this website, and users can request an unpopulated board through the tool. The complete process for designing a step-down switching regulator using the ADP3050 is provided in the following sections. Each section includes a list of recommended devices. These lists do not include every available device or manufacturer. They contain only surface-mount devices. Equivalent through- hole devices can be substituted if needed. In choosing components, keep in mind what is most important to the design, for example, efficiency, cost, and size. These ultimately determine which compo- nents are used. It is also important to ensure that the design specifications are clearly defined and reflect the worst-case conditions. Key specifications include the minimum and maximum input voltage, the output voltage and ripple, and the minimum and maximum load current. INDUCTOR SELECTION The inductor value determines the mode of operation for the regulator: continuous mode, where the inductor current flows continuously; or discontinuous mode, where the inductor current reduces to zero during every switch cycle. Continuous mode is the best choice for many applications. It provides higher output power, lower peak currents in the switch, inductor, and diode, and a lower inductor ripple current, which means lower output ripple voltage. Discontinuous mode allows the use of smaller magnetics, but at a price: lower available load current and higher peak and ripple currents. Designs with a high input voltage or a low load current often operate in discontinuous mode to minimize inductor value and size. The ADP3050 is designed to work well in both modes of operation. Continuous Mode The inductor current in a continuous mode system is a triangular waveform (equal to the ripple current) centered around a dc value (equal to the load current). The amount of ripple current is determined by the inductor value, and is usually between 20% and 40% of the maximum load current. To reduce the inductor size, ripple currents between 40% and 80% are often used in continuous mode designs with a high input voltage or a low output current. The inductor value is calculated using the following equation: )( 1 MAXIN OUT SWRIPPLE OUTMAXIN V V fI VV L ×× = (2) Where VIN(MAX) is the maximum input voltage, VOUT is the regulated output voltage, and fSW is the switching frequency (200 kHz). The initial choice for the amount of ripple current may seem arbitrary, but it serves as a good starting point for finding a standard off-the-shelf inductor value, such as 10 μH, 15 μH, 22 μH, 33 μH, and 47 μH. If a specific inductance value is to be used, simply rearrange Equation 2 to find the ripple current. For an 800 mA, 12 V to 5 V system, and a ripple current of 320 mA (40% of 800 mA) is chosen, the inductance is μH45.512 10200 0.32 512 3 =×××−=L A 47 μH inductor is the closest standard value that gives a ripple current of about 310 mA. The peak switch current is equal to the load current plus one-half the ripple current (this is also the peak current for the inductor and the catch diode). A95.0155.08.02 )()( =+=+= RIPPLEMAXOUTPKSW III (3) Pick an inductor with a dc (or saturation) current rating about 20% larger than ISW(PK) to ensure that the inductor is not running near the edge of saturation. For this example, 1.20 × 0.95 A = 1.14 A, use an inductor with a dc current rating of at least 1.2 A. The maxi- mum switch current is internally limited to 1.5 A, and this limit, along with the ripple current, determines the maximum load current the system can provide. If the load current decreases to below one-half the ripple current, the regulator operates in discontinuous mode. Discontinuous Mode For load currents less than approximately 0.5 A, discontinuous mode operation can be used. This allows the use of a smaller inductor, but the ripple current is much higher (which means a higher output ripple voltage). If a larger output capacitor must be used to reduce the output ripple voltage, the overall system may take up more board area than if a larger inductor is used. The operation and equations for the two modes are quite different, but the boundary between these two modes occurs when the ripple current is equal to twice the load current (when I RIPPLE = 2 × IOUT). From this, Equation 2 is used to find the minimum inductor value needed to keep the system in continuous mode operation (solve for the inductor value with IRIPPLE = 2 × IOUT). )( 1

2 MAXIN

V V fI VV L ××× = (4) Using an inductor below this value causes the system to operate in discontinuous mode.

Rev. C | Page 12 of 20 For a 400 mA, 24 V to 5 V system μH7.2424 10200 4.02 524 −≤DISL If the chosen inductor value is too small, the internal current limit trips each cycle and the regulator has trouble providing the necessary load current. Inductor Core Types and Materials Many types of inductors are currently available. Numerous core styles along with numerous core materials often make the selection process seem even more confusing. A quick overview of the types of inductors available makes the selection process a little easier to understand. Open core geometries (bobbin core) are usually less expensive than closed core geometries (toroidal core) and are a good choice for some applications, but care must be taken when they are used. In open core inductors, the magnetic flux is not completely contained inside the core. The radiating magnetic field generates electromagnetic interference (EMI), often inducing voltages onto nearby circuit board traces. These inductors may not be suitable for systems that contain very high accuracy circuits or sensitive magnetics. A few manufacturers have semiclosed and shielded cores, where an outer magnetic shield surrounds a bobbin core. These devices have less EMI than the standard open core and are usually smaller than a closed core. Most core materials used in surface-mount inductors are either powdered iron or ferrite. For many designs, material choice is arbitrary, but the properties of each material should be recognized. Ferrites have lower core losses than powdered iron, but the lower loss means a higher price. Powdered iron cores saturate softly (the inductance gradually reduces as current rating is exceeded), whereas ferrite cores saturate much more abruptly (the inductance rapidly reduces). Kool Mμ® is one type of ferrite that is specially designed to minimize core losses and heat generation (especially at switching frequencies above 100 kHz), but again, these devices are more expensive. The winding dc resistance (DCR) of the inductor must not be overlooked. A high DCR can decrease system efficiency by 2% to 5% for lower output voltages at heavy loads. To obtain a lower DCR means using a physically larger inductor, so a trade- off in size and efficiency must be made. The power loss due to this resistance is I OUT2 × DCR. For an 800 mA, 5 V to 3.3 V system with an inductor DCR of 100 mΩ, the winding resistance dissipates (0.82 A)2 × 0.1 Ω = 64 mW . This represents a power loss to the system of 64 mW/(3.3 V × 800 mA) = 2.4%. Typical DCR values are between 10 mΩ and 200 mΩ. Choosing an Inductor Several considerations must be made when choosing an inductor: cost, size, EMI, core and copper losses, and maximum current rating. Use the following steps to choose an inductor that is right for the system (refer to the calculations and descriptions in the Inductor Selection section). Contact the manufacturers for their full product offering, availability, and pricing. The manufacturers offer many more values and package sizes to suit numerous applications. 1. Choose a mode of operation, then calculate the inductor value using the appropriate equation. For continuous mode systems, a ripple current of 40% of the maximum load current is a good starting point. The inductor value can then be increased or decreased, if desired. 2. Calculate the peak switch current (this is the maximum current seen by the inductor). Make sure that the dc (or saturation) current rating of the inductor is high enough (around 1.2× the peak switch current). Inductors with dc current ratings of at least 1 A should be used for all designs. This provides a safety margin for start-up and fault conditions where the inductor current is higher than normal. If the current rating of an inductor is exceeded, the core saturates, causing the inductance value to decrease and the temperature of the inductor to increase. 3. Estimate the dc winding resistance based on the inductance value. A general rule is to allow approximately 5 mΩ of resistance per μH of inductance. 4. Pick the core material and type. First, decide if an open- core inductor can be used with the design. If this cannot be determined, try a few samples of each type (open core, semi closed core, shielded core, and closed core). Do not be discouraged from using open core inductors because they require extra care; just be aware of what to look for if used. They are quite small and inexpensive, and are used successfully in many different applications. OUTPUT CAPACITOR SELECTION The ADP3050 can be used with any type of output capacitor. The trade-offs between price, component size, and regulator performance can be evaluated to determine the best choice for each application. The effective series resistance (ESR) of the capacitor plays an important role in both the loop compensation and the system performance. The ESR provides a 0 in the feedback loop; therefore, the ESR value must be known so the loop can be compensated correctly (most manufacturers specify maximum ESR in their data sheets). The capacitor ESR also contributes to the output ripple voltage (V RIPPLE = ESR × IRIPPLE). Solid tantalum or multilayer ceramic capacitors are recommended, providing good performance with a small size and reasonable cost. Solid tantalum capacitors have a good combination of low ESR and high capacitance, and are available from several different manufacturers. Capacitance values from 22 μF to more than 500 μF can be used, but values of 47 μF to 220 μF are sufficient for most designs. A smaller value can be used, but ESR is size-dependent, so a smaller device has a higher ESR. Ensure that the ripple current of the capacitor rating is larger than the inductor ripple current (the ripple current flows into the output capacitor). Multilayer ceramic capacitors can be used in applications where minimum output voltage ripple is a priority. They have a very

Use the following steps to choose an appropriate capacitor.

1. Decide the maximum output ripple voltage for the design,
2. Decide what type of capacitor to use (tantalum, ceramic, or
3. Check the capacitor voltage rating and ripple current rating

the manufacturer’s data sheet.

1. Make sure the final choice for the output capacitor has

capacitance is in the 47 μF to 220 μF range.

1 A) and fast switching speed of a Schottky rectifier provide the

For this system, a 1N5817 is a good choice (rated at 20 V and 1 A). conditions. A larger 2 A or 3 A diode can be used if necessary. Table 4. Manufacturers Coilcraft Kemet Diodes, Inc.

Use the following steps to pick an appropriate catch diode. voltage and forward current ratings. average diode current is much lower. (1.2× the maximum input voltage) to provide a safety margin. Table 5. Schottky Diode Selection Guide

15 V 10BQ15 30BQ15

20 V 1N5817 B220 SK32

30 V V1N5818 B230 SK33

40 V 1N5819 B240 SK34

current rating, but they are typically larger and more costly. Refer to Table 4 for a list of capacitor manufacturers. reacting with the inductor to form a damped sinusoidal ringing. not harmful to normal circuit operation. version, the output voltage is set using two external resistors. placed close to the FB pin to prevent noise pickup. limited to a specific type of capacitor or a specific ESR range. to compensate the regulator loop. capacitors require different values of CC between 0.5 nF and 10 nF.

Rev. C | Page 15 of 20 shows the approximate unity-gain frequency of the loop. Again, always check the design over its full operating range of input voltage, output current, and temperature to ensure that the loop is compensated correctly. In addition to setting the zero location, RC also sets the high frequency gain of the error amplifier. If this gain is too large, output ripple voltage appears at the COMP pin (the output of the error amplifier) with enough amplitude to interfere with normal regulator operation. If this occurs, subharmonic switching results (the pulse width of the switch waveform changes, even though the output voltage stays regulated). The voltage ripple at the COMP pin should be kept below 100 mV to prevent subharmonic switching from occurring. The amount of ripple can be estimated by the following formula, where gm is the error amplifier transconductance (gm = 1250 μMho): ( ) ( ) OUT FB RIPPLECmRIPPLECOMP V VESRIRgV ××××=, (10) For example, a 12 V to 5 V , 800 mA regulator with an inductor of L = 47 μH has IRIPPLE = 310 mA (see example from the Continuous Mode section) if a 100 μF tantalum output capacitor with a maximum ESR of 100 mΩ and compensation values of RC = 4 kΩ and CC = 1 nF are used. The ripple voltage at the COMP pin is ( ) ( ) mV2.37 0.5 20.11.0310.0104101250 36 RIPPLECOMPV (11) If this ripple voltage is more than 100 mV , RC needs to be decreased to prevent subharmonic switching. Typical values for RC are in the range of 2 kΩ to 10 kΩ. For output voltages greater than 5 V , it may be necessary to add a small capacitor in parallel with R2, as shown in Figure 25. This improves stability and transient response. For tantalum output capacitors, the typical value for CF is 100 pF. For ceramic output capacitors, the typical value for CF is 400 pF. CURRENT LIMIT/FREQUENCY FOLDBACK The ADP3050 uses a cycle-by-cycle current limit to protect the device under fault and high stress conditions. When the current limit is exceeded, the power switch turns off until the beginning of the next oscillator cycle. If the voltage on the feedback pin drops below 80% of its nominal value, the oscillator frequency starts to decrease (see Figure 17 in the Typical Performance Characteristics section). The frequency gradually reduces to a minimum value of approximately 80 kHz (this minimum occurs when the feedback voltage falls to 30% of its nominal value). This reduces the power dissipation in the IC, the external diode, and the inductor during short-circuit conditions. This frequency foldback method provides complete device fault protection without interfering with the normal device operation. BIAS PIN CONNECTION To help improve efficiency, most of the internal operating current can be drawn from the lower voltage regulated output voltage instead of the input supply. For example, if the input voltage is 24 V and the output voltage is 5 V , a quiescent current of 4 mA wastes 96 mW if drawn from the input supply, but only 20 mW is drawn from the regulated 5 V output. This power savings is most evident at high input voltages and low load currents. The output voltage must be 3 V or higher to take advantage of this feature. BOOSTED DRIVE STAGE An external capacitor and diode are used to provide the boosted voltage needed for the special drive stage. If the output voltage is above 4 V , connect the anode of the boost diode to the regulated output; for output voltages less than or equal to voltages of ≤3 V, connect it to the input supply. For some low voltage systems, such as 5 V to 3.3 V converters, the anode of the boost diode can be connected to either the input or output voltage. During switch off time, the boost capacitor is charged up to the voltage at the anode of the boost diode. When the switch turns on, this voltage is added to the switch voltage (the boost diode is reverse- biased), providing a voltage higher than the input supply. The peak voltage appearing on the BOOST pin is the sum of the input voltage and the boost voltage (either V IN + VOUT or 2 × VIN). Ensure that this peak voltage does not exceed the BOOST pin maximum rating of 45 V . For most applications, a 1N4148 or 1N914 type diode can be used with a 220 nF capacitor. A 470 nF capacitor may be needed for output voltages between 3 V and 4 V . The boost capacitor should have an ESR of less than 2 Ω to ensure that it is adequately charged up during switch off time. Almost any type of film or ceramic capacitor can be used. START-UP/MINIMUM INPUT VOLTAGE For most designs, the regulated output voltage provides the boosted voltage for the drive stage. During startup, the output voltage is 0, so there is no boosted supply for the drive stage. To deal with this problem, the ADP3050 contains a backup drive stage to get everything started. As the output voltage increases, so does the boost voltage. When the boost voltage reaches approx- imately 2.5 V , the switch drives transition smoothly from the backup driver to the boosted driver. If the boost voltage decreases below approximately 2.5 V, resulting in a short-circuit or overload condition, the backup stage takes over to provide switch drive. The minimum input voltage needed for the ADP3050 to function correctly is about 3.6 V (this ensures proper operation of the internal circuitry), but a small amount of headroom is needed for all step-down regulators. The following formula gives the approximate minimum input voltage needed for a given system, where V SAT is the switch saturation voltage (see Figure 15 for the appropriate value of VSAT). Figure 13 also shows the typical minimum input voltage needed for 3.3 V and 5 V systems.

Rev. C | Page 16 of 20 85.0)( SATOUT MININ VVV += (12) THERMAL CONSIDERATIONS Several factors contribute to IC power dissipation: ac and dc switch losses, boost current, and quiescent current. The following formulas are used to calculate these losses to determine the power dissipation of the IC. These formulas assume continuous mode operation, but they provide a reasonable estimate for disconti- nuous mode systems (do not use these formulas to calculate efficiency at light loads). Switch loss ( )SWINOUTOV IN OUT SATOUTSW fVItV  ××= (13) Boost current loss IN OUT SW OUT BOOST V V β IP ×= (14) Quiescent current loss ( ) ( )BIASOUTQINQ IVIVP ×+×= (15) where: VSAT is ~0.6 V at IOUT = 800 mA (taken from Figure 15). fSW is the switch frequency (200 kHz). tOV is the switch current/voltage overlap time (~50 ns). βSW is the current gain of the NPN power switch (~50). IQ is the quiescent current drawn from VIN (~1 mA). IBIAS is the quiescent current drawn from VOUT (~4 mA). For example, a 5 V to 3.3 V system with IOUT = 800 mA ( ) mW357102000.58.01050 0.5 3.36.08.0 +   ××= SWP mW350.5 3.3 8.0 2 =×=BOOSTP QP For a total IC power dissipation of mW410=++= QBOOSTSWTOTAL PPPP (16) The ADP3050 is offered in a thermally enhanced (not Pb-free) 8-lead SOIC package with a thermal resistance, θJA, of 60.6°C/W, and in a standard Pb-free 8-lead SOIC package with θJA of 87.5°C / W. The maximum die temperature, TJ, is calculated using the thermal resistance and the maximum ambient temperature TOTALJAAJ PθTT ×+= (17) For the previous example (5 V to 3.3 V at 800 mA system, Pb- free 8-lead SOIC package using good layout techniques) with a worst-case ambient temperature of 70°C The maximum operating junction (die) temperature is 125°C, therefore this system operates within the safe limits of the ADP3050. Check the die temperature at minimum and maximum supply voltages to ensure proper operation under all conditions. Although the PCB and its copper traces provide sufficient heat sinking, it is important to follow the layout suggestions in the Board Layout Guidelines section. For any design that combines high output current with high duty cycle and/or high input voltage, the junction temperature must be calculated to ensure normal operation. Always use the equations in this section to estimate the power dissipation.

Figure 26. Main Switching Path away from the BOOST and SWITCH traces.

5 V input voltage with the following specifications:

Figure 27. Recommended Board Layout Figure 28. 5 V to 3.3 V Buck Regulator

Figure 29. Inverting (Buck-Boost) Regulator The average current diode is equal to the load current. maximum voltage rating of these pins is not exceeded.

to generate the −5 V output along with D2, C6, and C7. Figure 30. Dual Output +5 V and −5 V Regulator the buck winding is equal to VO + VD (VD is the diode drop). regulation of the −5 V output at light loads. regulator as described in the Applications Information section. output voltage ripple is a 30 mV peak-to-peak triangular wave.

REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. Figure 31. 8-Lead Standard Small Outline Package [SOIC_N] 2 Operating junction temperature is −40 to +125°C. registered trademarks are the property of their respective owners.