ADP1111 AD | Alldatasheet
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REV. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. a ADP1111 Tel: 617/329-4700 World Wide Web Site: http://www.analog.com Fax: 617/326-8703 © Analog Devices, Inc., 1996 Micropower, Step-Up/Step-Down SW Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V FUNCTIONAL BLOCK DIAGRAMS DRIVER ILIM SW1 SW2 VIN GND SET GAIN BLOCK/ ERROR AMP COMPARATOR FB 1.25V REFERENCE OSCILLATOR ADP1111 DRIVER ILIM SW1 SW2 VIN GND SET GAIN BLOCK/ ERROR AMP COMPARATOR SENSE 1.25V REFERENCE OSCILLATOR ADP1111-5 ADP1111-12 R1 R2 220k
FEATURES
Operates from 2 V to 30 V Input Voltage Range 72 kHz Frequency Operation Utilizes Surface Mount Inductors Very Few External Components Required Operates in Step-Up/Step-Down or Inverting Mode Low Battery Detector User Adjustable Current Limit Internal 1 A Power Switch Fixed or Adjustable Output Voltage 8-Pin DIP or SO-8 Package
APPLICATIONS
3 V to 5 V, 5 V to 12 V Step-Up Converters
9 V to 5 V, 12 V to 5 V Step-Down Converters
Laptop and Palmtop Computers Cellular Telephones Flash Memory VPP Generators Remote Controls Peripherals and Add-On Cards Battery Backup Supplies Uninterruptible Supplies Portable Instruments GENERAL DESCRIPTION The ADP1111 is part of a family of step-up/step-down switch- ing regulators that operates from an input voltage supply of 2 V to 12 V in step-up mode and up to 30 V in step-down mode. The ADP1111 can be programmed to operate in step-up/step- down or inverting applications with only 3 external components. The fixed outputs are 3.3 V, 5 V and 12 V; and an adjustable version is also available. The ADP1111 can deliver 100 mA at
5 V from a 3 V input in step-up mode, or it can deliver 200 mA
at 5 V from a 12 V input in step-down mode. Maximum switch current can be programmed with a single resistor, and an open collector gain block can be arranged in multiple configuration for low battery detection, as a post linear regulator, undervoltage lockout, or as an error amplifier. If input voltages are lower than 2 V, see the ADP1110.
–2– REV. 0 ADP1111–SPECIFICATIONS Parameter Conditions V S Min Typ Max Units QUIESCENT CURRENT Switch Off I Q 300 500 µA INPUT VOLTAGE Step-Up Mode V IN 2.0 12.6 V Step-Down Mode 30.0 V COMPARATOR TRIP POINT VOLTAGE ADP1111 1 1.20 1.25 1.30 V OUTPUT SENSE VOLTAGE ADP1111-3.3 V OUT 3.13 3.30 3.47 V ADP1111-52 4.75 5.00 5.25 V ADP1111-122 11.40 12.00 12.60 V COMPARATOR HYSTERESIS ADP1111 8 12.5 mV OUTPUT HYSTERESIS ADP1111-3.3 21 50 mV ADP1111-5 32 50 mV ADP1111-12 75 120 mV OSCILLATOR FREQUENCY f OSC 54 72 88 kHz DUTY CYCLE Full Load DC 43 50 65 % SWITCH ON TIME I LIM Tied to VIN tON 57 9 µs SW SATURATION VOLTAGE T A = +25°C STEP-UP MODE V IN = 3.0 V, ISW = 650 mA V SAT 0.5 0.65 V VIN = 5.0 V, ISW = 1 A 0.8 1.0 V STEP-DOWN MODE V IN = 12 V, ISW = 650 mA 1.1 1.5 V FEEDBACK PIN BIAS CURRENT ADP1111 V FB = 0 V I FB 160 300 nA SET PIN BIAS CURRENT V SET = VREF ISET 270 400 nA GAIN BLOCK OUTPUT LOW I SINK = 300 µA VSET = 1.00 V V OL 0.15 0.4 V REFERENCE LINE REGULATION 5 V ≤ VIN ≤ 30 V 0.02 0.075 %/V 2 V ≤ VIN ≤ 5 V 0.4 %/V GAIN BLOCK GAIN R L = 100 kΩ 3 AV 1000 6000 V/V CURRENT LIMIT T A = +25°C 220 Ω from ILIM to VIN ILIM 400 mA CURRENT LIMIT TEMPERATURE COEFFICIENT –0.3 %/ °C SWITCH OFF LEAKAGE CURRENT T A = +25°C Measured at SW1 Pin VSW1 = 12 V 1 10 µA MAXIMUM EXCURSION BELOW GND T A = +25°C ISW1 ≤ 10 µA, Switch Off –400 –350 mV NOTES 1This specification guarantees that both the high and low trip points of the comparator fall within the 1.20 V to 1.30 V range. 2The output voltage waveform will exhibit a sawtooth shape due to the comparator hysteresis. The output voltage on the fixed output versions will always be within the specified range. 3100 kΩ resistor connected between a 5 V source and the AO pin. All limits at temperature extremes are guaranteed via correlation using standard statistical methods. Specifications subject to change without notice. (08C ≤ TA ≤ +708C, VIN = 3 V unless otherwise noted)
accumulate on the human body and test equipment and can discharge without detection. precautions are recommended to avoid performance degradation or loss of functionality. *N = Plastic DIP, SO = Small Outline Package. should be connected between I LIM and VIN. by connecting a 220 Ω resistor. configuration, connect to an inductor/diode. down configuration, connect to inductor/diode. For step-up configuration, connect to ground. can sink 300 µA. It can be left open if unused. resistor that sets output voltage. Figure 1. 3 V to 5 V Step-Up Converter
Figure 2. Saturation Voltage vs. I SWITCH Current in Figure 3. Switch ON Voltage vs. I SWITCH Current In Figure 4. Quiescent Current vs. Input Voltage Figure 5. Oscillator Frequency vs. Input Voltage
0.3 SWITCH CURRENT – A
Figure 6. Maximum Switch Current vs. R LIM Figure 7. Oscillator Frequency vs. Temperature
Figure 8. Switch ON Time vs. Temperature Figure 9. Duty Cycle vs. Temperature Figure 10. Saturation Voltage vs. Temperature in Step-Up Figure 11. Switch ON Voltage vs. Temperature in Step-
350 QUIESCENT CURRENT
Figure 12. Quiescent Current vs. Temperature Figure 13. Feedback Bias Current vs. Temperature
nents for frequency compensation. transistor that can sink 300 µA. opposite polarity than the input voltage. a lower-cost alternative if EMI is not a problem.
- Define the operating parameters: minimum input voltage,
maximum input voltage, output voltage and output current.
- Select the appropriate conversion topology (step-up, step-
- Calculate the inductor value using the equations in the
Figure 14. Set Pin Bias Current vs. Temperature
–7–REV. 0 INDUCTOR SELECTION–STEP-UP CONVERTER In a step-up or boost converter (Figure 18), the inductor must store enough power to make up the difference between the input voltage and the output voltage. The power that must be stored is calculated from the equation: PL = VOUT +VD −VIN(MIN)() • IOUT() (Equation 1) where VD is the diode forward voltage (0.5 V for a 1N5818 Schottky). Because energy is only stored in the inductor while the ADP1111 switch is ON, the energy stored in the inductor on each switching cycle must be equal to or greater than: P f L OSC (Equation 2) in order for the ADP1111 to regulate the output voltage. When the internal power switch turns ON, current flow in the inductor increases at the rate of: IL t() = VIN R© 1− e −R©t L (Equation 3) where L is in Henrys and R' is the sum of the switch equivalent resistance (typically 0.8 Ω at +25°C) and the dc resistance of the inductor. In most applications, the voltage drop across the switch is small compared to V IN so a simpler equation can be used: IL t() = VIN L t (Equation 4) Replacing ‘t’ in the above equation with the ON time of the ADP1111 (7 µs, typical) will define the peak current for a given inductor value and input voltage. At this point, the inductor energy can be calculated as follows: EL = 1
2 L • I2 PEAK (Equation 5)
As previously mentioned, EL must be greater than P L/fOSC so that the ADP1111 can deliver the necessary power to the load. For best efficiency, peak current should be limited to 1 A or less. Higher switch currents will reduce efficiency because of increased saturation voltage in the switch. High peak current also increases output ripple. As a general rule, keep peak current as low as possible to minimize losses in the switch, inductor and diode. In practice, the inductor value is easily selected using the equations above. For example, consider a supply that will generate 12 V at 40 mA from a 9 V battery, assuming a 6 V end-of-life voltage. The inductor power required is, from Equation 1: PL = 12V + 0.5V − 6V() • 40 mA() = 260 mW On each switching cycle, the inductor must supply: PL fOSC = 260 mW 72 kHz = 3.6µJ Since the required inductor power is fairly low in this example, the peak current can also be low. Assuming a peak current of 500 mA as a starting point, Equation 4 can be rearranged to recommend an inductor value: L = VIN IL(MAX ) t = 6V 500 mA 7µs = 84 µH Substituting a standard inductor value of 68 µH with 0.2 Ω dc resistance will produce a peak switch current of: IPEAK = 6V 1.0Ω 1− e −1.0Ω• 7 µs 68 µH = 587 mA Once the peak current is known, the inductor energy can be calculated from Equation 5: EL = 1 2 68 µH() • 587 mA() = 11.7µJ Since the inductor energy of 11.7 µJ is greater than the P L/fOSC requirement of 3.6 µJ, the 68 µH inductor will work in this application. By substituting other inductor values into the same equations, the optimum inductor value can be selected. When selecting an inductor, the peak current must not exceed the maximum switch current of 1.5 A. If the equations shown above result in peak currents > 1.5 A, the ADP1110 should be considered. Since this device has a 70% duty cycle, more energy is stored in the inductor on each cycle. This results is greater output power. The peak current must be evaluated for both minimum and maximum values of input voltage. If the switch current is high when V IN is at its minimum, the 1.5 A limit may be exceeded at the maximum value of V IN. In this case, the ADP1111’s current limit feature can be used to limit switch current. Simply select a resistor (using Figure 6) that will limit the maximum switch current to the IPEAK value calculated for the minimum value of VIN. This will improve efficiency by producing a constant I PEAK as VIN increases. See the “Limiting the Switch Current” section of this data sheet for more information. Note that the switch current limit feature does not protect the circuit if the output is shorted to ground. In this case, current is only limited by the dc resistance of the inductor and the forward voltage of the diode. INDUCTOR SELECTION–STEP-DOWN CONVERTER The step-down mode of operation is shown in Figure 19. Unlike the step-up mode, the ADP1111’s power switch does not saturate when operating in the step-down mode; therefore, switch current should be limited to 650 mA in this mode. If the input voltage will vary over a wide range, the I LIM pin can be used to limit the maximum switch current. Higher switch current is possible by adding an external switching transistor as shown in Figure 21. The first step in selecting the step-down inductor is to calculate the peak switch current as follows: IPEAK = 2 IOUT DC VOUT + VD VIN − VSW + VD (Equation 6) where DC = duty cycle (0.5 for the ADP1111) VSW = voltage drop across the switch VD = diode drop (0.5 V for a 1N5818) IOUT = output current VOUT = the output voltage VIN = the minimum input voltage
–8– REV. 0 As previously mentioned, the switch voltage is higher in step- down mode than in step-up mode. V SW is a function of switch current and is therefore a function of V IN, L, time and VOUT. For most applications, a V SW value of 1.5 V is recommended. The inductor value can now be calculated: L = VIN MIN() − VSW − VOUT IPEAK
- tON (Equation 7) where tON = switch ON time (7 µs). If the input voltage will vary (such as an application that must operate from a 9 V, 12 V or 15 V source), an R LIM resistor should be selected from Figure 6. The R LIM resistor will keep switch current constant as the input voltage rises. Note that there are separate R LIM values for step-up and step-down modes of operation. For example, assume that +5 V at 300 mA is required from a +12 V to +24 V source. Deriving the peak current from Equation 6 yields: IPEAK = 2•300 mA 0.5 5 + 0.5 12 − 1.5+ 0.5 = 600 mA Then, the peak current can be inserted into Equation 7 to calculate the inductor value: L = 12 − 1.5− 5 600 mA •7µs = 64 µH Since 64 µH is not a standard value, the next lower standard value of 56 µH would be specified. To avoid exceeding the maximum switch current when the input voltage is at +24 V, an R LIM resistor should be specified. Using the step-down curve of Figure 6, a value of 560 Ω will limit the switch current to 600 mA. INDUCTOR SELECTION–POSITIVE-TO-NEGATIVE CONVERTER The configuration for a positive-to-negative converter using the ADP1111 is shown in Figure 22. As with the step-up converter, all of the output power for the inverting circuit must be supplied by the inductor. The required inductor power is derived from the formula: P = IL OUT VVOUT D+() • () (Equation 8) The ADP1111 power switch does not saturate in positive-to- negative mode. The voltage drop across the switch can be modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω resistor. When the switch turns on, inductor current will rise at a rate determined by: IL t() = VL R© 1− e − R©t L (Equation 9) where: R' = 0.65 Ω + RL(DC) VL = VIN – 0.75 V For example, assume that a –5 V output at 50 mA is to be generated from a +4.5 V to +5.5 V source. The power in the inductor is calculated from Equation 8: During each switching cycle, the inductor must supply the following energy: PL fOSC = 275 mW 72 kHz = 3.8µJ Using a standard inductor value of 56 µH with 0.2 Ω dc resistance will produce a peak switch current of: IPEAK = 4.5V − 0.75V 0.65Ω+ 0.2Ω 1− e − 0.85Ω• 7 µs 56 µH = 445 mA Once the peak current is known, the inductor energy can be calculated from (Equation 9): EL = 1 2 56 µH() • 445 mA() = 5.54µJ Since the inductor energy of 5.54 µJ is greater than the P L/fOSC requirement of 3.82 µJ, the 56 µH inductor will work in this application. The input voltage only varies between 4.5 V and 5.5 V in this application. Therefore, the peak current will not change enough to require an R LIM resistor and the ILIM pin can be connected directly to VIN. Care should be taken, of course, to ensure that the peak current does not exceed 650 mA. CAPACITOR SELECTION For optimum performance, the ADP1111’s output capacitor must be selected carefully. Choosing an inappropriate capacitor can result in low efficiency and/or high output ripple. Ordinary aluminum electrolytic capacitors are inexpensive but often have poor Equivalent Series Resistance (ESR) and Equivalent Series Inductance (ESL). Low ESR aluminum capacitors, specifically designed for switch mode converter applications, are also available, and these are a better choice than general purpose devices. Even better performance can be achieved with tantalum capacitors, although their cost is higher. Very low values of ESR can be achieved by using OS-CON capacitors (Sanyo Corporation, San Diego, CA). These devices are fairly small, available with tape-and-reel packaging and have very low ESR. The effects of capacitor selection on output ripple are demon- strated in Figures 15, 16 and 17. These figures show the output of the same ADP1111 converter that was evaluated with three different output capacitors. In each case, the peak switch current is 500 mA, and the capacitor value is 100 µF. Figure 15 shows a Panasonic HF-series 16-volt radial cap. When the switch turns off, the output voltage jumps by about 90 mV and then decays as the inductor discharges into the capacitor. The rise in voltage indicates an ESR of about 0.18 Ω . In Figure 16, the aluminum electrolytic has been replaced by a Sprague 293D series, a 6 V tantalum device. In this case the output jumps about 30 mV, which indicates an ESR of 0.06 Ω . Figure 17 shows an OS-CON 16–volt capacitor in the same circuit, and ESR is only 0.02 Ω .
RHYS, with a value of 1 M Ω to 10 MΩ , provides the hysteresis. resistor, and RHYS creates the hysteresis. Figure 28. All Surface Mount +3 V to +5 V Step-Up Converter
9 V to 5 V Step-Down Converter
Figure 29. 9 V to 5 V Step-Down Converter
20 V to 5 V Step-Down Converter
Figure 30. 20 V to 5 V Step-Down Converter Figure 31. +5 V to –5 V Converter
–15–REV. 0 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 8-Lead Plastic DIP (N-8) 0.430 (10.92) 0.348 (8.84) 0.280 (7.11) 0.240 (6.10) PIN 1 SEATING PLANE0.022 (0.558) 0.014 (0.356) 0.060 (1.52) 0.015 (0.38) 0.210 (5.33) MAX 0.130 (3.30) MIN 0.070 (1.77) 0.045 (1.15) 0.100 (2.54) BSC 0.160 (4.06) 0.115 (2.93) 0.325 (8.25) 0.300 (7.62) 0.015 (0.381) 0.008 (0.204) 0.195 (4.95) 0.115 (2.93) 8-Lead SOIC (SO-8) 0.1968 (5.00) 0.1890 (4.80) 8 5 0.2440 (6.20) 0.2284 (5.80) PIN 1 0.1574 (4.00) 0.1497 (3.80) 0.0688 (1.75) 0.0532 (1.35)SEATING PLANE 0.0098 (0.25) 0.0040 (0.10) 0.0192 (0.49) 0.0138 (0.35) 0.0500 (1.27) BSC 0.0098 (0.25) 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41) 0.0196 (0.50) 0.0099 (0.25)x 45°
–16– C2213–12–10/96PRINTED IN U.S.A.