AAT2552IRN-CAE-T1

AAT2552 ANALOGICTECH | Alldatasheet

Document overview

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• PDF pages: 33

Technical content

Features

• Battery Charger: — Input Voltage Range: 4V to 6.5V — Programmable Charging Current up to 500mA — Highly Integrated Battery Charger
• Charging Device
• Reverse Blocking Diode
• Current Sensing
• Step-Down Converter: — Input Voltage Range: 2.7V to 5.5V — Output Voltage Range: 0.6V to V IN — 300mA Output Current — Up to 96% Efficiency — 45µA Quiescent Current — 1.5MHz Switching Frequency — 120µs Start-Up Time
• Linear Regulator: — 300mA Output Current — Low Dropout: 400mV at 300mA — Fast Line and Load Transient Response — High Accuracy: ±1.5% — 85µA Quiescent Current
• Short-Circuit, Over-Temperature, and Current Limit Protection
• TDFN34-16 Package
• -40°C to +85°C Temperature Range

Applications

• Bluetooth ® Headsets
• Cellular Phones
• GPS
• Handheld Instruments
• MP3 and Portable Music Players
• PDAs and Handheld Computers
• Portable Media Players AAT2552 Total Power Solution for Portable Applications Typical Application BATT- ADP GND BAT MODE ISET INB INA ENB ENA BATT+ AAT2552 Adapter/USB Input STAT EN_BATEnable RSET C Battery Pack OUT SystemL1 FBB LX RFBB2 RFBB1 RFBA2 RFBA1 COUTB 4.7μF VOUTB OUTA FBACOUTA VOUTA 2552.2007.04.1.0 1 SystemPower™

(Top View) AGND FBB ENB EN_BAT ISET MODE ENA FBA BAT PGND LX STAT ADP INB INA OUTA Pin # Symbol Function 1 EN_BAT Enable pin for the battery charger. When connected to logic low, the battery charger is dis- abled and consumes less than 1µA of current. When connected to logic high, the charger operates normally (pulled down internally). 2 ISET Charge current set point. Connect a resistor from this pin to ground. Refer to typical charac- teristics curves for resistor selection. 3 AGND Analog ground. 4 FBB Feedback input for the step-down converter. This pin must be connected directly to an exter- nal resistor divider. Nominal voltage is 0.6V. 5 ENB Enable pin for the step-down converter. When connected to logic low, the step-down convert- er is disabled and consumes less than 1µA of current. When connected to logic high, the con- verter operates normally (pulled up internally). 6 MODE Pulled down internally for automatic PWM/LL operation. Connect to logic high for forced PWM. Drive with external clock signal to synchronize step-down converter to external clock in PWM mode. 7 ENA Enable pin for the linear regulator. When connected to logic low, the regulator is disabled and consumes less than 1µA of current. When connected to logic high, the LDO operates normal- ly (pulled up internally). 8 FBA Feedback input for the LDO. This pin must be connected directly to an external resistor divider. Nominal voltage is 1.24V. 9 OUTA Linear regulator output. Connect a 2.2µF capacitor from this pin to ground. 10 INA Linear regulator input voltage. Connect a 1µF or greater capacitor from this pin to ground. 11 INB Input voltage for the step-down converter. 12 LX Output of the step-down converter. Connect the inductor to this pin. Internally, it is connected to the drain of both high- and low-side MOSFETs. 13 PGND Power ground. 14 BAT Battery charging and sensing. Connect to positive terminal of Lithium-ion/polymer battery. 15 ADP Input from USB port or AC wall adapter. 16 STAT Open drain status pin for charger. EP Exposed paddle (bottom): connect to ground directly beneath the package. AAT2552 Total Power Solution for Portable Applications 2 2552.2007.04.1.0

Symbol Description Value Units PD Maximum Power Dissipation 2.0 W θJA Thermal Resistance2 50 °C/W Symbol Description Value Units VINA, VINB Input Voltage to GND 6.0 V VADP Adapter Voltage to GND -0.3 to 7.5 V VLX LX to GND -0.3 to V IN + 0.3 V VFB FB to GND -0.3 to V IN + 0.3 V VEN ENA, ENB, EN_BAT to GND -0.3 to 6.0 V VX BAT, ISET, STAT -0.3 to V ADP + 0.3 V TJ Operating Junction Temperature Range -40 to 150 °C TLEAD Maximum Soldering Temperature (at leads, 10 sec) 300 °C AAT2552 Total Power Solution for Portable Applications 2552.2007.04.1.0 3 1. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at condi- tions other than the operating conditions specified is not implied. Only one Absolute Maximum Rating should be applied at any one time. 2. Mounted on an FR4 board.

Electrical Characteristics1 VINB = 3.6V; TA = -40°C to +85°C, unless otherwise noted. Typical values are T A = 25°C. Symbol Description Conditions Min Typ Max Units Step-Down Converter VIN Input Voltage 2.7 5.5 V VINB Rising 2.6 VVUVLO UVLO Threshold Hysteresis 250 mV VOUT Output Voltage Tolerance2 IOUTB = 0 to 300mA, -3.0 3.0 %VINB = 2.7V to 5.5V VOUT Output Voltage Range 0.6 V INB V IQ Quiescent Current No Load 45 90 µA ISHDN Shutdown Current V ENB = GND 1.0 µA ILIM P-Channel Current Limit 300 mA RDS(ON)H High-Side Switch On Resistance 0.3 Ω RDS(ON)L Low-Side Switch On Resistance 0.5 Ω ILXLEAK LX Leakage Current V INB = 5.5V, VLX = 0 to VINB 1.0 µA ΔVOUT/ΔVOUT Load Regulation I OUTB = 0mA to 300mA 0.4 % ΔVLinereg/ΔVIN Line Regulation V INB = 2.7V to 5.5V 0.1 %/V VFB Feedback Threshold Voltage Accuracy V INB = 3.6V 0.591 0.6 0.609 V IFB FB Leakage Current V OUTB = 1.0V 0.2 µA FOSC Oscillator Frequency 1.5 MHz TS Startup Time From Enable to Output 120 µsRegulation TSD Over-Temperature Shutdown Threshold 140 °C THYS Over-Temperature Shutdown Hysteresis 15 °C VEN(L) Enable Threshold Low 0.6 V VEN(H) Enable Threshold High 1.4 V IEN Input Low Current V INB = VENB = 5.5V -1.0 1.0 µA AAT2552 Total Power Solution for Portable Applications 4 2552.2007.04.1.0 1. The AAT2552 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assu red by design, characterization, and correlation with statistical process controls. 2. Output voltage tolerance is independent of feedback resistor network accuracy.

Electrical Characteristics1 VINA = VOUT(NOM) + 1V. IOUT = 1mA, COUT = 2.2µF, TA = -40°C to +85°C, unless otherwise noted. Typical val- ues are TA = 25°C. Symbol Description Conditions Min Typ Max Units Linear Regulator VOUT Output Voltage Tolerance IOUTA = 1mA TA = 25°C -1.5 1.5 %to 300mA T A = -40°C to +85°C -2.5 2.5 VOUT Output Voltage Range 1.2 3.3 V VFB Feedback Voltage Accuracy 1.22 1.24 1.26 V VIN Input Voltage VOUT + 5.5 VVDO VDO Dropout Voltage3 IOUTA = 300mA; VOUT = 3.3V 400 650 mV ΔVOUT/ Line Regulation V INA = VOUTA + 1 to 5.0V 0.09 %/VVOUT*ΔVIN IOUT Output Current V OUTA > 2.0V 300 mA ISC Short-Circuit Current V OUTA < 0.4V 400 mA IQ Quiescent Current V INA = 5V; VENA = VIN 85 150 µA ISHDN Shutdown Current V INA = 5V; VENA = 0V 1.0 µA 1kHz 70 PSRR Power Supply Rejection IOUTA =10mA 10kHz 50 dBRatio 1MHz 30 TSD Over-Temperature 140 °CShutdown Threshold THYS Over-Temperature 15 °CShutdown Hysteresis eN Output Noise e NBW = 100Hz to 100kHz 95 µVRMS/ √Hz TC Output Voltage 8 ppm/°CTemperature Coefficient VEN(L) Enable Threshold Low 0.6 V VEN(H) Enable Threshold High 1.4 V IEN Enable Input Current V INA = VENA = 5.5V 1.0 µA AAT2552 Total Power Solution for Portable Applications 2552.2007.04.1.0 5 1. The AAT2552 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assu red by design, characterization, and correlation with statistical process controls. 2. VDO is defined as VIN - VOUT when VOUT is 98% of nominal. 3. For VOUT <2.3V, VDO = 2.5V - VOUT.

Total Power Solution for Portable Applications 6 2552.2007.04.1.0 Electrical Characteristics1 VADP = 5V; TA = -40°C to +85°C, unless otherwise noted. Typical values are T A = 25°C. Symbol Description Conditions Min Typ Max Units Battery Charger Operation VADP Adapter Voltage Range 4.0 6.5 V VUVLO Under-Voltage Lockout (UVLO) Rising Edge 3 4 V UVLO Hysteresis 150 mV IOP Operating Current Charge Current = 200mA 0.5 1 mA ISHUTDOWN Shutdown Current V BAT = 4.25V, VEN_BAT = GND 0.3 1 µA ILEAKAGE Reverse Leakage Current from BAT Pin VBAT = 4V, ADP Pin Open 0.4 2 µA Voltage Regulation VBAT_EOC End of Charge Accuracy 4.158 4.20 4.242 V VMIN Preconditioning Voltage Threshold 2.8 3.0 3.2 V VRCH Battery Recharge Voltage Threshold Measured from V BAT_EOC -0.1 V Current Regulation ICH Charge Current Programmable Range 30 500 mA ΔICH/ICH Charge Current Regulation Tolerance I CHARGE = 200mA -10 10 % VSET ISET Pin Voltage 2 V KI_A Current Set Factor: ICH/ISET 800 Charging Devices RDS(ON) Charging Transistor On Resistance V ADP = 5.5V 0.5 0.8 Ω Logic Control/Protection VEN(H) Enable Threshold High 1.6 V VEN(L) Enable Threshold Low 0.4 V VSTAT Output Low Voltage STAT Pin Sinks 4mA 0.4 V ISTAT STAT Pin Current Sink Capability 8 mA VOVP Over-Voltage Protection Threshold 4.4 V ITK/ICHG Pre-Charge Current I CH = 100mA 10 % ITERM/ICHG Charge Termination Threshold Current 10 % 1. The AAT2552 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assu red by design, characterization, and correlation with statistical process controls.

Typical Characteristics–Battery Charger Battery Charging Current vs. Battery Voltage VBAT (V) ICH (mA) 100 200 300 400 500 600 RSET = 3.24K RSET = 5.62K RSET = 8.06K RSET = 16.2K RSET = 31.6K Constant Charging Current vs. Temperature (RSET = 8.06kΩΩ) Temperature (°C) ICH (mA) 190 193 195 198 200 203 205 208 210 -50 -25 0 25 50 75 100 Operating Current vs. Temperature (VIN = 5.0V; RSET = 8.06kΩΩ) Temperature (°C) IOP (µA) 440 460 480 500 520 540 -50 -25 0 25 50 75 100 Sleep Mode Current vs. Input Voltage (RSET = 8.06kΩΩ) Input Voltage (V) ISLEEP (nA) 100 200 300 400 500 600 700 800 85°C 25°C -40°C Constant Charging Current vs. Set Resistors (VIN = 5.0V) RSET (kΩΩ) ICH (mA) 100 1000 10000 1 10 100 1000 Operating Supply Current vs. RSET (VIN = 5.0V) RSET (kΩΩ) IOP (µA) 100 1000 1 10 100 Constant Current Mode Preconditioning Mode AAT2552 Total Power Solution for Portable Applications 2552.2007.04.1.0 7

Typical Characteristics–Battery Charger Preconditioning Charge Current vs. Temperature (RSET = 8.06kΩΩ) Temperature (°C) ITK (mA) 19.2 19.6 20.0 20.4 20.8 -40 -15 10 35 60 85 Preconditioning Voltage Threshold vs. Temperature (RSET = 8.06kΩΩ) Temperature (°C) VMIN (V) 2.97 2.98 2.99 3.00 3.01 3.02 3.03 -40 -15 10 35 60 85 Recharging Threshold Voltage vs. Temperature (RSET = 8.06kΩΩ) Temperature (°C) VRCH (V) 4.04 4.06 4.08 4.10 4.12 4.14 4.16 -40 -15 10 35 60 85 Constant Charging Current vs. Input Voltage (VIN = 5.62V) VIN (V) ICH (mA) 285 290 295 300 305 310 4 4.5 5 5.5 6 6.5 VIN = 3.6V VIN = 4V VIN = 3.3V End of Charge Voltage Regulation vs. Temperature (VIN = 5V; RSET = 8.06kΩΩ) Temperature (°C) VBAT_EOC (V) 4.185 4.190 4.195 4.200 4.205 4.210 4.215 -40 -15 10 35 60 85 End of Charge Battery Voltage vs. Input Voltage VIN (V) VBAT_EOC (V) 4.194 4.196 4.198 4.200 4.202 4.204 4.206 4.5 5 5.5 6 6.5 RSET = 8.06kΩ RSET = 31.6kΩ AAT2552 Total Power Solution for Portable Applications 8 2552.2007.04.1.0

Typical Characteristics–Battery Charger Enable Threshold High vs. Input Voltage (RSET = 8.06kΩΩ) VIN (V) VEN(H) (V) 0.7 0.8 0.9 1.0 1.1 1.2 -40°C 85°C 25°C Enable Threshold Low vs. Input Voltage (RSET = 8.06kΩΩ) VIN (V) VEN(L) (V) 0.6 0.7 0.8 0.9 1.0 1.1 -40°C 25°C 85°C AAT2552 Total Power Solution for Portable Applications 2552.2007.04.1.0 9

Typical Characteristics–Step-Down Converter Efficiency vs. Load (VOUT = 1.2V; L = 1.5µH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 VIN = 2.7V VIN = 3.6V VIN = 5.0V VIN = 4.2V DC Regulation (VOUT = 1.2V; L = 1.5μH) Output Current (mA) Output Error (%) -1.0 -0.5 0.0 0.5 1.0 0.1 1 10 100 1000 VIN = 5.0V VIN = 2.7VVIN = 4.2V VIN = 3.6V Efficiency vs. Load (VOUT = 1.8V; L = 3.3µH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 VIN = 2.7V VIN = 3.6V VIN = 5.0V VIN = 4.2V DC Regulation (VOUT = 1.8V; L = 3.3µH) Output Current (mA) Output Error (%) -1.0 -0.5 0.0 0.5 1.0 0.1 1 10 100 1000 VIN = 4.2V VIN = 5.0V VIN = 3.6V VIN = 2.7V Efficiency vs. Load (VOUT = 3.3V; L = 5.6µH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 VIN = 5.0V VIN = 4.2V VIN = 3.6V DC Regulation (VOUT = 3.3V; L = 5.6µH) Output Current (mA) Output Error (%) -1.0 -0.5 0.0 0.5 1.0 0.1 1 10 100 1000 VIN = 5.0V VIN = 4.2V VIN = 3.6V AAT2552 Total Power Solution for Portable Applications 10 2552.2007.04.1.0

Typical Characteristics–Step-Down Converter N-Channel RDS(ON) vs. Input Voltage VIN (V) RDS(ON)L (mΩΩ) 300 400 500 600 700 800 900 1000 2.5 3 3.5 4 4.5 5 5.5 6 25°C 85°C 100°C 120°C P-Channel RDS(ON) vs. Input Voltage VIN (V) RDS(ON)H (mΩΩ) 100 200 300 400 500 600 2.5 3 3.5 4 4.5 5 5.5 6 25°C 85°C 100°C 120°C Output Voltage Accuracy vs. Temperature (VIN = 3.6V; VO = 1.8V; IOUT = 150mA) Temperature (°°C) Output Accuracy (%) -2.0 -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 2.0 -40 -15 10 35 60 85 No Load Quiescent Current vs. Input Voltage Input Voltage (V) IQ (mA) 85°C 25°C -40°C Line Regulation (VOUT = 1.8V) Input Voltage (V) Accuracy (%) -0.4 -0.3 -0.2 -0.1 0.1 0.2 IOUT = 10mA IOUT = 50mA IOUT = 150mA Soft Start (VIN = 3.6V; VOUT = 1.8V; IOUT = 150mA) Time (100µs/div) Enable and Output Voltage (top) (V) Inductor Current (bottom) (A) 0.0 0.1 0.2 0.3 VEN VOUT IL AAT2552 Total Power Solution for Portable Applications 2552.2007.04.1.0 11

Typical Characteristics–Step-Down Converter Output Voltage Ripple (VIN = 3.6V; VOUT = 1.8V; IOUT = 300mA) Time (0.2µs/div) Output Voltage (AC coupled) (top) (V) Inductor Current (bottom) (A) 1.79 1.80 1.81 0.1 0.2 0.3 0.4 Output Voltage Ripple (VIN = 3.6V; VOUT = 1.8V; IOUT = 1mA) Time (5µs/div) Output Voltage (AC coupled) (top) (V) Inductor Current (bottom) (A) -20 -0.10 -0.05 0.00 0.05 Load Transient Response (10mA to 300mA; VIN = 3.6V; VOUT = 1.8V; COUT = 4.7µF; C = 100pF) Time (20µs/div) Output Voltage (top) (V) Load and Inductor Current (bottom) (A) 1.6 1.7 1.8 1.9 2.0 -0.2 0.0

0.2 VOUT

(VOUT = 1.8V @ 150mA, CFF = 100pF) Time (25µs/div) Output Voltage (top) (V) Input Voltage (bottom) (V) 1.75 1.80 1.85 1.90 3.1 3.6 4.1 4.6 AAT2552 Total Power Solution for Portable Applications 12 2552.2007.04.1.0

Typical Characteristics–LDO Regulator Output Voltage vs. Temperature (VIN = 3.6V; VO = 1.8V; IOUT = 150mA) Temperature (°°C) Output Voltage (V) 3.296 3.297 3.298 3.299 3.300 3.301 -40 -15 10 35 60 85 Enable Threshold Voltage vs. Input Voltage Input Voltage (V) VENABLE (V) 0.82 0.84 0.86 0.88 0.9 0.92 0.94 0.96 VEN(H) VEN(L) Dropout Characteristics Input Voltage (V) Output Voltage (V) 2.4 2.6 2.8 3.2 3.4 3.6 3.8 3 3.2 3.4 3.6 3.8 4 IOUT = 0mA IOUT = 300mA IOUT = 100mA IOUT = 50mA Dropout Voltage vs. Output Current Output Current (mA) Dropout Voltage (V) 0.0 0.1 0.2 0.3 0.4 0.5 0 50 100 150 200 250 300 -40°C 85°C 25°C Quiescent Current vs. Temperature (VIN = 5V) Temperature (°°C) IQ (µA) 100 110 120 -40 -15 10 35 60 85 Dropout Voltage vs. Temperature Temperature (°°C) Dropout Voltage (V) 0.0 0.1 0.2 0.3 0.4 0.5 -40 -20 0 20 40 60 80 100 120 IL = 300mA IL = 200mA IL = 100mA IL = 50mA AAT2552 Total Power Solution for Portable Applications 2552.2007.04.1.0 13

Total Power Solution for Portable Applications 14 2552.2007.04.1.0 Typical Characteristics–LDO Regulator LDO Output Noise (COUT = 4.7µF; IOUT = 10mA; RLOAD = 330; 98.33µVrms) Frequency (kHz) nVrms/sqrt (Hz) 100 1000 10000 0.01 0.1 1 10 100 1000 Turn-Off Response Time (VIN = 4.2V; IOUT = 300mA) Time (50µs/div) Enable and Output Voltage VEN = 2V/div VOUT = 1V/div Turn-On Time From Enable (VIN = 4.2V; IOUT = 300mA) Time (100µs/div) Enable and Output Voltage VEN = 2V/div VOUT = 1V/div Line Transient Response (IOUT = 300mA) Time (100µs/div) Output Voltage (top) (V) Input Voltage (bottom) (V) 3.30 3.35 3.40 4.0 4.5 5.0 VOUT VIN Load Transient Response (1mA to 300mA; VIN = 5.0V; VOUT = 3.3V) Time (100µs/div) Output Voltage (top) (V) Output Current (bottom) (A) 3.2 3.4 3.6 -0.2 0.0 0.2 0.4 VOUT IL

Total Power Solution for Portable Applications 2552.2007.04.1.0 15 Functional Block Diagram Charge Control Current Compare Reverse Blocking ADP ISET UVLO Err. Amp STAT EN_BAT Logic DH DL Voltage Reference Input Voltage Reference INA OUTA AGND Fast Start Control Active Feedback Control BAT INB FBB LX PGND ENB ENA MODE FBA From Charger Section Constant Current Charge Status CV/Pre-Charge Over-Temperature Protection Over-Current Protection Err. Amp Functional Description The AAT2552 is a high performance power man- agement IC comprised of a lithium-ion/polymer battery charger, a step-down converter, and a lin- ear regulator. The linear regulator is designed for high-speed turn-on and fast transient response, and good power supply ripple rejection. The step- down converter operates in both fixed and variable frequency modes for high efficiency performance. The switching frequency is 1.5MHz, minimizing the size of the inductor. In light load conditions, the device enters power-saving mode; the switch- ing frequency is reduced and the converter con- sumes 45µA of current, making it ideal for battery- operated applications. Battery Charger The battery charger is designed for single-cell lithi- um-ion/polymer batteries using a constant current and constant voltage algorithm. The battery charg- er operates from the adapter/USB input voltage range from 4V to 6.5V. The adapter/USB charging current level can be programmed up to 500mA for rapid charging applications. A status monitor out- put pin is provided to indicate the battery charge state by directly driving one external LED. Internal device temperature and charging state are fully monitored for fault conditions. In the event of an over-voltage or over-temperature failure, the device will automatically shut down, protecting the charging device, control system, and the battery under charge. Other features include an integrat- ed reverse blocking diode and sense resistor.

Total Power Solution for Portable Applications 16 2552.2007.04.1.0 Switch-Mode Step-Down Converter The step-down converter operates with an input voltage of 2.7V to 5.5V. The switching frequency is 1.5MHz, minimizing the size of the inductor. Under light load conditions, the device enters power-sav- ing mode; the switching frequency is reduced, and the converter consumes 45µA of current, making it ideal for battery-operated applications. The output voltage is programmable from V IN to as low as 0.6V. Power devices are sized for 300mA current capability while maintaining over 96% efficiency at full load. Light load efficiency is maintained at greater than 80% down to 1mA of load current. A high-DC gain error amplifier with internal compen- sation controls the output. It provides excellent transient response and load/line regulation. The AAT2552 synchronous step-down converter can be synchronized to an external clock signal applied to the MODE pin. Linear Regulator The advanced circuit design of the linear regulator has been specifically optimized for very fast start- up. This proprietary CMOS LDO has also been tai- lored for superior transient response characteris- tics. These traits are particularly important for appli- cations that require fast power supply timing. The high-speed turn-on capability is enabled through implementation of a fast-start control cir- cuit which accelerates the power-up behavior of fundamental control and feedback circuits within the LDO regulator. The LDO regulator output has been specifically optimized to function with low- cost, low-ESR ceramic capacitors; however, the design will allow for operation over a wide range of capacitor types. The regulator comes with complete short-circuit and thermal protection. The combination of these two internal protection circuits gives a comprehen- sive safety system to guard against extreme adverse operating conditions. The regulator features an enable/disable function. This pin (ENA) is active high and is compatible with CMOS logic. The LDO regulator will go into the dis- able shutdown mode when the voltage on the ENA pin falls below 0.6V. If the enable function is not needed in a specific application, it may be tied to INA to keep the LDO regulator in a continuously on state. Under-Voltage Lockout The AAT2552 has internal circuits for UVLO and power on reset features. If the ADP supply voltage drops below the UVLO threshold, the battery charger will suspend charging and shut down. When power is reapplied to the ADP pin or the UVLO condition recovers, the system charge con- trol will automatically resume charging in the appropriate mode for the condition of the battery. If the input voltage of the step-down converter drops below UVLO, the internal circuit will shut down. Protection Circuitry Over-Voltage Protection An over-voltage protection event is defined as a condition where the voltage on the BAT pin exceeds the over-voltage protection threshold OVP). If this over-voltage condition occurs, the charger control circuitry will shut down the device. The charger will resume normal charging operation after the over-voltage condition is removed. Current Limit, Over-Temperature Protection For overload conditions, the peak input current is lim- ited at the step-down converter. As load impedance decreases and the output voltage falls closer to zero, more power is dissipated internally, which causes the internal die temperature to rise. In this case, the ther- mal protection circuit completely disables switching, which protects the device from damage. The battery charger has a thermal protection circuit which will shut down charging functions when the internal die temperature exceeds the preset ther- mal limit threshold. Once the internal die tempera- ture falls below the thermal limit, normal charging operation will resume. Control Loop The AAT2552 contains a compact, current mode step-down DC/DC controller. The current through the P-channel MOSFET (high side) is sensed for current loop control, as well as short-circuit and overload protection. A fixed slope compensation signal is added to the sensed current to maintain stability for duty cycles greater than 50%. The peak current mode loop appears as a voltage-pro- grammed current source in parallel with the output capacitor. The output of the voltage error amplifier programs the current mode loop for the necessary

Total Power Solution for Portable Applications 2552.2007.04.1.0 17 peak switch current to force a constant output volt- age for all load and line conditions. Internal loop compensation terminates the transconductance voltage error amplifier output. The error amplifier reference is fixed at 0.6V. Battery Charging Operation Battery charging commences only after checking several conditions in order to maintain a safe charg- ing environment. The input supply (ADP) must be above the minimum operating voltage (UVLO) and the enable pin must be high (internally pulled down). When the battery is connected to the BAT pin, the charger checks the condition of the battery and determines which charging mode to apply. If the bat- tery voltage is below V MIN, the charger begins bat- tery pre-conditioning by charging at 10% of the pro- grammed constant current; e.g., if the programmed current is 150mA, then the pre-conditioning current (trickle charge) is 15mA. Pre-conditioning is purely a safety precaution for a deeply discharged cell and will also reduce the power dissipation in the internal series pass MOSFET when the input-output voltage differential is at its highest. Pre-conditioning continues until the battery voltage reaches V MIN (see Figure 1). At this point, the charger begins constant-current charging. The cur- rent level for this mode is programmed using a sin- gle resistor from the ISET pin to ground. Programmed current can be set from a minimum 15mA up to a maximum of 500mA. Constant cur- rent charging will continue until the battery voltage reaches the voltage regulation point, V BAT. When the battery voltage reaches VBAT, the battery charg- er begins constant voltage mode. The regulation voltage is factory programmed to a nominal 4.2V (±0.5%) and will continue charging until the charg- ing current has reduced to 10% of the programmed current. After the charge cycle is complete, the pass device turns off and the device automatically goes into a power-saving sleep mode. During this time, the series pass device will block current in both direc- tions, preventing the battery from discharging through the IC. The battery charger will remain in sleep mode, even if the charger source is disconnected, until one of the following events occurs: the battery ter- minal voltage drops below the V RCH threshold; the charger EN pin is recycled; or the charging source is reconnected. In all cases, the charger will mon- itor all parameters and resume charging in the most appropriate mode. Figure 1: Current vs. Voltage Profile During Charging Phases. Constant Current Charge Phase Constant Voltage Charge Phase Preconditioning Trickle Charge Phase Charge Complete Voltage Constant Current Mode Voltage Threshold Regulated Current Trickle Charge and Termination Threshold I = CC / 10 I = Max CC

Total Power Solution for Portable Applications 18 2552.2007.04.1.0 Battery Charging System Operation Flow Chart Power On Reset Power Input Voltage VADP > VUVLO Fault Conditions Monitoring OV, OT Preconditioning Test VMIN > VBAT Current Phase Test VBAT_EOC > VBAT Voltage Phase Test IBAT > ITERM No No Yes No Preconditioning (Trickle Charge) Constant Current Charge Mode Constant Voltage Charge Mode Yes Yes Yes Charge Completed Charge Control No Recharge Test VRCH > VBAT Yes No Shut Down Yes Enable Yes No

Total Power Solution for Portable Applications 2552.2007.04.1.0 19

Application Information

The EN_BAT pin is internally pulled down. When pulled to a logic high level, the battery charger is enabled. When left open or pulled to a logic low level, the battery charger is shut down and forced into the sleep state. Charging will be halted regardless of the battery voltage or charging state. When it is re- enabled, the charge control circuit will automatically reset and resume charging functions with the appro- priate charging mode based on the battery charge state and measured cell voltage from the BAT pin. Separate ENA and ENB inputs are provided to independently enable and disable the LDO and step-down converter, respectively. This allows sequencing of the LDO and step-down outputs dur- ing startup. The LDO is enabled when the ENA pin is pulled high. The control and feedback circuits have been optimized for high-speed, monotonic turn-on char- acteristics. The step-down converter is enabled when the ENB pin is pulled high. Soft start increases the inductor current limit point in discrete steps when the input voltage or ENB input is applied. It limits the current surge seen at the input and eliminates output voltage overshoot. When pulled low, the ENB input forces the AAT2552 into a low-power, non-switching state. The step-down converter input current during shutdown is less than 1µA. Adapter or USB Power Input Constant current charge levels up to 500mA may be programmed by the user when powered from a sufficient input power source. The battery charger will operate from the adapter input over a 4.0V to 6.5V range. The constant current fast charge cur- rent for the adapter input is set by the R SET resistor connected between ISET and ground. Refer to Table 1 for recommended R SET values for a desired constant current charge level. Programming Charge Current The fast charge constant current charge level is user programmed with a set resistor placed between the ISET pin and ground. The accuracy of the fast charge, as well as the preconditioning trick- le charge current, is dominated by the tolerance of the set resistor used. For this reason, a 1% toler- ance metal film resistor is recommended for the set resistor function. Fast charge constant current lev- els from 30mA to 500mA may be set by selecting the appropriate resistor value from Table 1. Table 1: R SET Values. Figure 2: Constant Charging Current vs. Set Resistor Values. Charge Status Output The AAT2552 provides battery charge status via a status pin. This pin is internally connected to an N- channel open drain MOSFET, which can be used to drive an external LED. The status pin can indicate several conditions, as shown in Table 2. RSET (kΩΩ) ICH (mA) 100 1000 1 10 100 1000 Normal Set Resistor ICHARGE (mA) Value R1 (k Ω) 500 3.24 400 4.12 300 5.36 250 6.49 200 8.06 150 10.7 100 16.2 50 31.6 40 38.3 30 53.6 20 78.7 15 105

Total Power Solution for Portable Applications 2552.2007.04.1.0 21 By substitution, we can derive the maximum charge current before reaching the thermal limit condition (thermal cycling). The maximum charge current is the key factor when designing battery charger applications. In general, the worst condition is the greatest volt- age drop across the IC, when battery voltage is charged up to the preconditioning voltage thresh- old. Figure 4 shows the maximum charge current in different ambient temperatures. Figure 4: Maximum Charging Current Before Thermal Cycling Becomes Active. There are three types of losses associated with the step-down converter: switching losses, conduction losses, and quiescent current losses. Conduction losses are associated with the R DS(ON) characteris- tics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assuming continuous conduction mode (CCM), a simplified form of the losses is given by: I Q is the step-down converter quiescent current. The term tsw is used to estimate the full load step- down converter switching losses. For the condition where the step-down converter is in dropout at 100% duty cycle, the total device dis- sipation reduces to: Since R DS(ON), quiescent current, and switching losses all vary with input voltage, the total losses should be investigated over the complete input voltage range. Given the total losses, the maximum junction tem- perature can be derived from the θ JA for the TDFN34-16 package which is 50°C/W. Capacitor Selection Linear Regulator Input Capacitor (C6) An input capacitor greater than 1µF will offer supe- rior input line transient response and maximize power supply ripple rejection. Ceramic, tantalum, or aluminum electrolytic capacitors may be select- ed for C IN. There is no specific capacitor ESR requirement for CIN. However, for 300mA LDO reg- ulator output operation, ceramic capacitors are rec- ommended for C IN due to their inherent capability over tantalum capacitors to withstand input current surges from low impedance sources such as bat- teries in portable devices. Battery Charger Input Capacitor (C1) In general, it is good design practice to place a decoupling capacitor between the ADP pin and GND. An input capacitor in the range of 1µF to 22µF is recommended. If the source supply is unregulated, it may be necessary to increase the capacitance to keep the input voltage above the under-voltage lockout threshold during device enable and when battery charging is initiated. If the adapter input is to be used in a system with an external power supply source, such as a typical AC-to-DC wall adapter, then a C IN capacitor in the range of 10µF should be used. A larger input TJ(MAX) = PTOTAL · ΘJA + TAMB PTOTAL = IO 2 · RDSON(H) + IQ · VIN PTOTAL IO 2 · (RDSON(H) · VO + RDSON(L) · [VIN - VO]) VIN + (tsw · FS · IO + IQ) · VIN VIN (V) ICH(MAX) (mA) 100 150 200 250 300 350 400 450 500 TA = 60°C TA = 85°C TA = 45°C (TJ(MAX) - TA) θJA VIN - VBAT ICH(MAX) = - VIN · IOP (PD(MAX) - VIN · IOP) VIN - VBAT ICH(MAX) =

Total Power Solution for Portable Applications 22 2552.2007.04.1.0 capacitor in this application will minimize switching or power transient effects when the power supply is "hot plugged" in. Step-Down Converter Input Capacitor (C6) Select a 4.7µF to 10µF X7R or X5R ceramic capac- itor for the input. To estimate the required input capacitor size, determine the acceptable input rip- ple level (V PP) and solve for C IN. The calculated value varies with input voltage and is a maximum when V IN is double the output voltage. Always examine the ceramic capacitor DC voltage coefficient characteristics when selecting the prop- er value. For example, the capacitance of a 10µF, 6.3V, X5R ceramic capacitor with 5.0V DC applied is actually about 6µF. The maximum input capacitor RMS current is: The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current. for V IN = 2 · VO The term appears in both the input voltage ripple and input capacitor RMS current equations and is a maximum when V O is twice VIN. This is why the input voltage ripple and the input capacitor RMS current ripple are a maximum at 50% duty cycle. The input capacitor provides a low impedance loop for the edges of pulsed current drawn by the step- down converter. Low ESR/ESL X7R and X5R ceramic capacitors are ideal for this function. To minimize stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C6) can be seen in the evaluation board layout in Figure 7. A laboratory test set-up typically consists of two long wires running from the bench power supply to the evaluation board input voltage pins. The induc- tance of these wires, along with the low-ESR ceramic input capacitor, can create a high Q net- work that may affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage during load transients. Errors in the loop phase and gain meas- urements can also result. Since the inductance of a short PCB trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. In applications where the input power source lead inductance cannot be reduced to a level that does not affect the converter performance, a high ESR tantalum or aluminum electrolytic capacitor should be placed in parallel with the low ESR, ESL bypass ceramic capacitor. This dampens the high Q net- work and stabilizes the system. The linear regula- tor and the step-down convertor share the same input capacitor on the evaluation board. VO VIN VO VIN IO RMS(MAX)I 2= VO VIN VO VIN VO VIN VO VIN CIN(MIN) = 1 VPP IO VO VIN VO VIN VO VIN CIN = VO VIN VPP IO

Total Power Solution for Portable Applications 2552.2007.04.1.0 23 Linear Regulator Output Capacitor (C5) For proper load voltage regulation and operational stability, a capacitor is required between OUT and GND. The C OUT capacitor connection to the LDO regulator ground pin should be made as directly as practically possible for maximum device perform- ance. Since the regulator has been designed to function with very low ESR capacitors, ceramic capacitors in the 1.0µF to 10µF range are recom- mended for best performance. Applications utilizing the exceptionally low output noise and optimum power supply ripple rejection should use 2.2µF or greater for C OUT. In low output current applications, where output load is less than 10mA, the minimum value for C OUT can be as low as 0.47µF. Battery Charger Output Capacitor (C2) The battery charger of the AAT2552 only requires a 1µF ceramic capacitor on the BAT pin to maintain circuit stability. This value should be increased to 10µF or more if the battery connection is made any distance from the charger output. If the AAT2552 is to be used in applications where the battery can be removed from the charger, such as with desktop charging cradles, an output capacitor greater than 10µF may be required to prevent the device from cycling on and off when no battery is present. Step-Down Converter Output Capacitor (C3) The output capacitor limits the output ripple and provides holdup during large load transitions. A 4.7µF to 10µF X5R or X7R ceramic capacitor typi- cally provides sufficient bulk capacitance to stabi- lize the output during large load transitions and has the ESR and ESL characteristics necessary for low output ripple. For enhanced transient response and low temperature operation applications, a 10µF (X5R, X7R) ceramic capacitor is recommended to stabilize extreme pulsed load conditions. The output voltage droop due to a load transient is dominated by the capacitance of the ceramic out- put capacitor. During a step increase in load cur- rent, the ceramic output capacitor alone supplies the load current until the loop responds. Within two or three switching cycles, the loop responds and the inductor current increases to match the load current demand. The relationship of the output volt- age droop during the three switching cycles to the output capacitance can be estimated by: Once the average inductor current increases to the DC load level, the output voltage recovers. The above equation establishes a limit on the minimum value for the output capacitor with respect to load transients. The internal voltage loop compensation also limits the minimum output capacitor value to 4.7µF. This is due to its effect on the loop crossover frequency (bandwidth), phase margin, and gain margin. Increased output capacitance will reduce the crossover frequency with greater phase margin. The maximum output capacitor RMS ripple current is given by: Dissipation due to the RMS current in the ceram- ic output capacitor ESR is typically minimal, resulting in less than a few degrees rise in hot- spot temperature. Inductor Selection The step-down converter uses peak current mode control with slope compensation to maintain stabil- ity for duty cycles greater than 50%. The output inductor value must be selected so the inductor current down slope meets the internal slope com- pensation requirements. The internal slope com- pensation for the AAT2552 is 0.45A/µsec. This equates to a slope compensation that is 75% of the inductor current down slope for a 1.8V output and 3.0µH inductor. VOUT · (VIN(MAX) - VOUT) RMS(MAX)I L · FS · VIN(MAX) COUT = 3 · ΔILOAD VDROOP · FS

Total Power Solution for Portable Applications 24 2552.2007.04.1.0 For most designs, the step-down converter operates with inductor values from 1µH to 4.7µH. Table 6 dis- plays inductor values for the AAT2552 for various output voltages. Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation characteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive loss- es due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. The 3.0µH CDRH2D09 series inductor selected from Sumida has a 150mΩ DCR and a 470mA DC current rating. At full load, the inductor DC loss is 9.375mW which gives a 2.08% loss in efficiency for a 250mA, 1.8V output. Adjustable Output Voltage for the Step- down Converter Resistors R2 and R3 of Figure 5 program the out- put of the step down converter and regulate at a voltage higher than 0.6V. To limit the bias current required for the external feedback resistor string while maintaining good noise immunity, the sug- gested value for R3 is 59k Ω. Decreased resistor values are necessary to maintain noise immunity on the FBB pin, resulting in increased quiescent current. Table 3 summarizes the resistor values for various output voltages. With enhanced transient response for extreme pulsed load application, an external feed-forward capacitor (C8 in Figure 5) can be added. Table 3: Adjustable Resistor Values For Step-Down Converter. Adjustable Output Voltage for the LDO The output voltage for the LDO can be pro- grammed by an external resistor divider network. As shown below, the selection of R4 and R5 is a straightforward matter. R5 is chosen by considering the tradeoff between the feedback network bias cur- rent and resistor value. Higher resistor values allow stray capacitance to become a larger factor in circuit performance whereas lower resistor values increase bias current and decrease efficiency. To select appro- priate resistor values, first choose R5 such that the feedback network bias current is reasonable. Then, according to the desired V OUT, calculate R4 according to the equation below. An example calculation follows. An R5 value of 59kΩ is chosen, resulting in a small feedback network bias current of 1.24V/59kΩ ≈ 21µA. The desired output voltage is 1.8V. From this information, R4 is calculated from the equation below. The result is R4 = 26.64kΩ. Since 26.64kΩ is not a standard 1%-value, 26.7kΩ is selected. From this example calculation, for V OUT = 1.8V, use R5 = 59kΩ and R4 = 26.7kΩ. Example output voltages and cor- responding resistor values are provided in Table 4. VOUT VREF R3 = 59kΩ R3 = 221kΩ VOUT (V) R2 (k Ω) R2 (k Ω) 0.8 19.6 75 0.9 29.4 113 1.0 39.2 150 1.1 49.9 187 1.2 59.0 221 1.3 68.1 261 1.4 78.7 301 1.5 88.7 332 1.8 118 442 1.85 124 464 2.0 137 523 2.5 187 715 3.3 267 1000 VOUT VREF 3.3V 0.6V 0.75 ⋅ VO L = = ≈ 1.67 ⋅ VOm 0.75 ⋅ VO 0.45A µsec AA µsec 0.75 ⋅ VO m = = = 0.45 L 0.75 ⋅ 1.8V 3.0µH A µsec

Total Power Solution for Portable Applications 2552.2007.04.1.0 25 Table 4: Adjustable Resistor Values for the LDO. Printed Circuit Board Layout Considerations For the best results, it is recommended to physi- cally place the battery pack as close as possible to the AAT2552 BAT pin. To minimize voltage drops on the PCB, keep the high current carrying traces adequately wide. Refer to the AAT2552 evaluation board for a good layout example (see Figures 6 and 7). The following guidelines should be used to help ensure a proper layout. 1. The input capacitors (C1, C6) should connect as closely as possible to ADP, INA, and INB. It is pos- sible to use two input capacitors for INA and INB. 2. C4 and L1 should be connected as closely as possible. The connection of L1 to the LX pin should be as short as possible. Do not make the node small by using narrow trace. The trace should be kept wide, direct, and short. 3. The feedback pin should be separate from any power trace and connect as closely as possible to the load point. Sensing along a high-current load trace will degrade DC load regulation. Feedback resistors should be placed as closely as possible to the FBB pin to minimize the length of the high impedance feedback trace. If possi- ble, they should also be placed away from the LX (switching node) and inductor to improve noise immunity. 4. The resistance of the trace from PGND should be kept to a minimum. This will help to minimize any error in DC regulation due to differences in the potential of the internal signal ground and the power ground. 5. A high density, small footprint layout can be achieved using an inexpensive, miniature, non- shielded, high DCR inductor. R4 Standard 1% Values VOUT (V) (R5 = 59kΩ) R4 (kΩ) 3.3 97.6 2.8 75.0 2.5 60.4 2.0 36.5 1.8 26.7 1.5 12.4

Total Power Solution for Portable Applications 26 2552.2007.04.1.0 Figure 5: AAT2552 Evaluation Board Schematic. Figure 6: AAT2552 Evaluation Board Figure 7: AAT2552 Evaluation Board Top Side Layout. Bottom Side Layout. ADP RED LED 1.5K 10μF 8.06K 10μF 59k 4.7μF 59k 100pF (Optional) 4.7μF VoA VoB

123 Power Selection

12 EN_BAT

EN_LDO JP2 EN_BUCK JP3 ENB 5EN_BAT1 MODE6 LX12 ISET 2 AGND3 INB 11 FBB4 PGND13 BAT 14 ADP15 STAT16 INA 10 OUTA 9 ENA 7 FBA 8 VOUTB (V) VOUTA (V)R2 ( Ω) R4 ( Ω) 13 9.2K 1.8 118K 3.0 237K 1.2 59K 2.5 187K 3.0μH (CDRH2D09/HP; DCR 150mΩ; 470mA @ 20°C) 1.5μH (CDRH2D09/HP; DCR 88mΩ; 730mA @ 20°C) 4.7μH (CDRH2D09/HP; DCR 230mΩ; 410mA @ 20°C) 2.2μH (CDRH2D09/HP; DCR 115mΩ; 600mA @ 20°C) 3.9μH (CDRH2D09/HP; DCR 180mΩ; 450mA @ 20°C) 10μF (at bottom layer)

1.24 R4 short, R5 open

1.8 26.7K 2.0 36.5K 2.8 75.0K 1.5 12.4K 2.5 60.4K

0.6 R2 short, R3 open

3.0 97.6K3.3 267K 5.6μH (CDRH2D09/HP; DCR 260m Ω; 370mA @ 20°C)

Total Power Solution for Portable Applications 2552.2007.04.1.0 27 Table 5: AAT2552 Evaluation Board Component Listing. Component Part Number Description Manufacturer U1 AAT2552IRN Total Power Solution for Portable Applications AnalogicTech C1, C2 ECJ-1VB0J106M CER 10μF 6.3V X5R 0603 Panansonic C3, C5 GRM188R60J475KE19 CER 4.7μF 6.3V X5R 0603 Murata C6 GRM319R61A106KE19 CER 10μF 10V X5R 1206 Murata C4 GRM1886R1H101JZ01J CER 100pF 50V 5% R2H 0603 Murata L1 CDRH2D09 Shielded SMD, 3x3x1mm Sumida R6 Chip Resistor 1.5KΩ, 5%, 1/4W 0603 Vishay R1 Chip Resistor 8.06KΩ, 1%, 1/4W 0603 Vishay R2 Chip Resistor 118KΩ, 1%, 1/4W 0603 Vishay R3, R5 Chip Resistor 59KΩ, 1%, 1/4W 0603 Vishay R4 Chip Resistor 60.4KΩ, 1%, 1/4W 0603 Vishay JP1, JP2, PRPN401PAEN Conn. Header, 2mm zip Sullins Electronics JP3, JP4 D1 CMD15-21SRC/TR8 Red LED 1206 Chicago Miniature Lamp

Total Power Solution for Portable Applications 28 2552.2007.04.1.0 Step-Down Converter Design Example (to be updated) Specifications VO = 1.8V @ 250mA, Pulsed Load ΔILOAD = 200mA VIN = 2.7V to 4.2V (3.6V nominal) FS = 1.5MHz TAMB = 85°C 1.8V Output Inductor (use 3.0µH; see Table 3) For Sumida inductor CDRH2D09-3R0, 3.0µH, DCR = 150m Ω. 1.8V Output Capacitor VDROOP = 0.1V 3.0µH · 1.5MHz · 4.2V 23 RMSI L1 · FS · VIN(MAX) = · 3 · ΔILOAD VDROOP · FS 3 · 0.2A 0.1V · 1.5MHz COUT = = = 4µF (use 4.7µF)

• = 66mArms (VO) · (VIN(MAX) - VO) = Pesr = esr · IRMS 2 = 5mΩ · (66mA)2 = 21.8µW IPKL1 = IO + ΔIL1 = 250mA + 114mA = 364mA2 PL1 = IO 2 ⋅ DCR = 250mA2 ⋅ 150mΩ = 9.375mW L1 = 1.67 ⋅ VO2 = 1.67 ⋅ 1.8V = 3µHµsec A µsec A

Input Ripple VPP = 25mV AAT2552 Losses TJ(MAX) = TAMB + ΘJA · PLOSS = 85°C + (50°C/W) · 26.14mW = 86.3°C PTOTAL + (tsw · FS · IO + IQ) · VIN IO 2 · (RDSON(H) · VO + RDSON(L) · [VIN -VO]) VIN 4.2V IO RMSI P = esr · IRMS 2= = 0.1Arms CIN = = = 1.38µF (use 4.7µF )1 VPP IO 25mV 0.2A AAT2552 Total Power Solution for Portable Applications 2552.2007.04.1.0 29

Total Power Solution for Portable Applications 30 2552.2007.04.1.0 Table 6: Step-Down Converter Component Values. Table 7: Suggested Inductors and Suppliers. Inductance Max DC DCR Size (mm) Manufacturer Part Number (µH) Current (mA) (m Ω) LxWxH Type Sumida CDRH2D09-1R5 1.5 730 110 3.0x3.0x1.0 Shielded Sumida CDRH2D09-2R2 2.2 600 144 3.0x3.0x1.0 Shielded Sumida CDRH2D09-2R5 2.5 530 150 3.0x3.0x1.0 Shielded Sumida CDRH2D09-3R0 3.0 470 194 3.0x3.0x1.0 Shielded Sumida CDRH2D09-3R9 3.9 450 225 3.0x3.0x1.0 Shielded Sumida CDRH2D09-4R7 4.7 410 287 3.0x3.0x1.0 Shielded Sumida CDRH2D09-5R6 5.6 370 325 3.0x3.0x1.0 Shielded Sumida CDRH2D11-1R5 1.5 900 68 3.2x3.2x1.2 Shielded Sumida CDRH2D11-2R2 2.2 780 98 3.2x3.2x1.2 Shielded Sumida CDRH2D11-3R3 3.3 600 123 3.2x3.2x1.2 Shielded Sumida CDRH2D11-4R7 4.7 500 170 3.2x3.2x1.2 Shielded Taiyo Yuden NR3010T1R5N 1.5 1200 80 3.0x3.0x1.0 Shielded Taiyo Yuden NR3010T2R2M 2.2 1100 95 3.0x3.0x1.0 Shielded Taiyo Yuden NR3010T3R3M 3.3 870 140 3.0x3.0x1.0 Shielded Taiyo Yuden NR3010T4R7M 4.7 750 190 3.0x3.0x1.0 Shielded FDK MIPWT3226D-1R5 1.5 1200 90 3.2x2.6x0.8 Chip shielded FDK MIPWT3226D-2R2 2.2 1100 100 3.2x2.6x0.8 Chip shielded FDK MIPWT3226D-3R0 3.0 1000 120 3.2x2.6x0.8 Chip shielded FDK MIPWT3226D-4R2 4.2 900 140 3.2x2.6x0.8 Chip shielded 1. For reduced quiescent current, R3 = 221k Ω. Output Voltage R3 = 59kΩ R3 = 221kΩ L1 (µH)VOUTB (V) R3 (kΩ) R1 (kΩ) 0.6 R2 short, R3 open R2 short, R3 open 1.5 0.8 19.6 75 1.5 0.9 29.4 113 1.5 1.0 39.2 150 1.5 1.1 49.9 187 1.5 1.2 59.0 221 1.5 1.3 68.1 261 1.5 1.4 78.7 301 2.2 1.5 88.7 332 2.7 1.8 118 442 3.0/3.3 1.85 124 464 3.0/3.3 2.0 137 523 3.0/3.3 2.5 187 715 3.9/4.2 3 237 887 4.9 3.3 267 1000 5.6

Total Power Solution for Portable Applications 2552.2007.04.1.0 31 Table 8: Surface Mount Capacitors. Value Voltage Temp. Case Manufacturer Part Number (µF) Rating Co. Size Murata GRM21BR61A106KE19 10 10 X5R 0805 Murata GRM188R60J475KE19 4.7 6.3 X5R 0603 Murata GRM188R61A225KE34 2.2 10 X5R 0603 Murata GRM188R60J225KE19 2.2 6.3 X5R 0603 Murata GRM188R61A105KA61 1.0 10 X5R 0603 Murata GRM185R60J105KE26 1.0 6.3 X5R 0603

Total Power Solution for Portable Applications 32 2552.2007.04.1.0

Ordering Information

All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more information, please visit our website at http://www.analogictech.com/pbfree. Package Marking 1 Part Number (Tape and Reel)2 TDFN34-16 UVXYY AAT2552IRN-CAE-T1 1. XYY = assembly and date code. 2. Sample stock is generally held on part numbers listed in BOLD. Legend Voltage Code Adjustable A(0.6) 0.9 B Adjustable (1.2) E 1.5 G 1.8 I 1.9 Y 2.5 N 2.6 O 2.7 P 2.8 Q 2.85 R 2.9 S 3.0 T 3.3 W 4.2 C

Total Power Solution for Portable Applications 2552.2007.04.1.0 33 © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech pr oduct. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service with- out notice. Except as provided in AnalogicTech’s terms and conditions of sale, AnalogicTech assumes no liability whatsoever, an d AnalogicTech disclaims any express or implied war- ranty relating to the sale and/or use of AnalogicTech products including liability or warranties relating to fitness for a part icular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. In order to minimize risks associated with the customer’s applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. Testing and other quality control techniques are utilized to the extent An alogicTech deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holder s. Advanced Analogic Technologies, Inc. 830 E. Arques Avenue, Sunnyvale, CA 94085 Phone (408) 737-4600 Fax (408) 737-4611 Package Information1 TDFN34-16 All dimensions in millimeters. 3.000 ± 0.050 1.600 ± 0.050 0.050 ± 0.050 0.229 ± 0.051 (4x)

0.850 MAX

4.000 ± 0.050 3.300 ± 0.050 Index Area Detail "A" Top View Bottom View Side View 0.350 ± 0.100 0.230 ± 0.0500.450 ± 0.050 Detail "A" Pin 1 Indicator (optional) C0.3 1. The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the lead terminals due to the manufacturing process. A solder fillet at the exposed copper edge cannot be guaranteed and is not re quired to ensure a proper bottom solder connection.