AAT2514

AAT2514_08 ANALOGICTECH | Alldatasheet

Document overview

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Technical content

Features

• V IN Range:2.5V to 5.5V
• Up to 600mA Output Current
• High Efficiency: Up to 96%
• 1.5MHz Constant Frequency Operation
• 100% Duty Cycle Dropout Operation
• Low R DS(ON) Internal Switches: 0.35Ω
• Current Mode Operation for Excellent Line and Load Transient Response
• Adaptive Slope Compensation
• Soft Start
• Short-Circuit and Thermal Fault Protection
• <1 μA Shutdown Current
• Power-On Reset Output
• Small, Thermally Enhanced TDFN33 -10 Package
• -40°C to +85°C Temperature Range

Applications

• Cellular Telephones
• Digital Still Cameras
• PDAs
• Portable Media Players
• Wireless and DSL Modems Typical Application EN1 LX1 GND AAT2514 L1 2.2μH VOUT1 1.8V 10μF 10μF VIN 2.5V to 5.5V IN EN2 LX 2 FB2 FB1 VOUT2 2.5V 100kΩ RESET L2 2.2μH 316kΩ 634kΩR3 316kΩ 1MΩ 10μF POR

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT2514 Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Pin Descriptions Pin # Symbol Function 1 FB1 Feedback input for channel 1. Connect FB1 to the center point of an external resistor divider. The feed- back threshold voltage is 0.6V. 2 EN1 Channel 1 enable pin. Active high. In shutdown, all functions are disabled drawing <1μA supply current. Do not leave EN1 fl oating. 3 IN Power supply input pin. Must be closely decoupled to GND with a 2.2 μF or greater ceramic capacitor. 4 LX1 Channel 1 switching node pin. Connect the output inductor to this pin.

5 GND Ground

6 N/C No connection

8 POR Power-on reset, active low. Open drain. External resistor (100k Ω) is required. - 9 EN2 Channel 2 enable pin. Active high. In shutdown, all functions are disabled drawing <1μA supply cur- rent. Do not leave EN2 fl oating. 10 FB2 Feedback input for channel 2. Connect FB2 to the center point of an external resistor divider. The feed- back threshold voltage is 0.6V. EP Exposed paddle. The exposed paddle should be connected to board ground plane and GND. The ground plane should include a large exposed copper pad under the package for thermal dissipation (see pack- age outline). Pin Configuration AAT2514-IDE TDFN33-10 (Top View) EXPOSED PAD FB1 EN1 IN LX1 GND FB2 EN2 POR LX2 NC 10-Lead (3mm X 3mm) Plastic Thin DFN Exposed Pad is PGND Must be connected to GND.

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT2514 Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Absolute Maximum Ratings1 Symbol Description Value Units VIN Input Supply Voltage -0.3 to +6.0 V VEN1, VEN2 EN1, EN2 Voltages -0.3 to V IN + 0.3 V VFB1, VFB2 FB1, FB2 Voltages -0.3 to V IN + 0.3 V VLX1, VLX2 LX1, LX2 Voltages -0.3 to V IN + 0.3 V VPOR POR Voltage -0.3 to +6.0 V TA Operating Temperature Range2 -40 to +85 °C TJ Junction Temperature2 +125 °C TSTORAGE Storage Temperature Range -65 to +150 °C TLEAD Lead Temperature (Soldering, 10s) +300 °C Recommended Operating Conditions Symbol Description Value Units θJA Thermal Resistance3 45 °C/W PD Maximum Power Dissipation at TA = 25°C 2.2 W 1. Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. 2. TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: T J = TA + PD · θJA. 3. Thermal resistance is specified with approximately 1 square inch of 1 oz copper.

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT2514 Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET

Electrical Characteristics

VIN = VEN = 3.6V, TA = 25°C, unless otherwise noted. Symbol Description Conditions Min Typ Max Units Step-Down Converter VIN Input Voltage Range 2.5 5.5 V IQ Input DC Supply Current Active Mode, VFB = 0.5V 500 800 μAShutdown Mode, EN1 = EN2 = 0V, VIN = 4.2V 0.3 2.0 VFB Regulated Feedback Voltage TA = 25°C, Channel 1 or 2 0.5880 0.6000 0.6120 VTA = 0°C ≤ TA ≤ +85°C, Channel 1 or 2 0.5865 0.6000 0.6135 TA = -40°C ≤ TA ≤ +85°C, Channel 1 or 2 (See Note 2) 0.5850 0.6000 0.6150 IFB FB Input Bias Current -30 30 nA ΔVOUT/ VOUT/ΔVIN Output Voltage Line Regulation VIN = 2.5V to 5.5V, IOUT = 10mA 0.11 0.40 %/V ΔVOUT/ VOUT/ΔIOUT Output Voltage Load Regulation IOUT = 10mA to 600mA 0.0015 %/mA ILIM Maximum Output Current V IN = 3.0V 600 mA TS Startup Time From Enable to Output Regulation 100 μs TSD Over-Temperature Shutdown Threshold 140 °C THYS Over-Temperature Shutdown Hysteresis 15 °C FOSC Oscillator Frequency V FB = 0.6V 1.2 1.5 1.8 MHz RDS(ON) P-Channel MOSFET I LX = 300mA 0.35 0.45 ΩN-Channel MOSFET I LX = 300mA 0.28 0.45 Peak Inductor Current V IN = 3V, VFB = 0.5V; Duty Cycle <35% 1.20 A VEN(L) Enable Threshold Low 0.3 V VEN(H) Enable Threshold High 1.5 V IEN EN Input Current -1.0 1.0 μA Power-On Reset Threshold (POR) VFB Ramping Up 8.5 %VFB Ramping Down -8.5 Power-On Reset Delay 175 ms Power-On Reset On-Resistance 100 Ω 1. Specifications over the temperature range are guaranteed by design and characterization. 2. The regulated feedback voltage is tested in an internal test mode that connects V FB to the output of the error amplifier.

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT2514 Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Typical Characteristics Efficiency vs. Load Current (VOUT = 2.5V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 4.2V VIN = 3.3V VIN = 2.7V Efficiency vs. Load Current (VOUT = 1.8V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.3V VIN = 4.2V Efficiency vs. Load Current (VOUT = 1.5V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.3V VIN = 4.2V Efficiency vs. Load Current (VOUT = 1.2V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.3V VIN = 4.2V Efficiency vs. Input Voltage (VOUT = 1.8V; TA = 25ºC) Input Voltage (V) Efficiency (%) 100 ILOAD = 100mA ILOAD = 600mA Load Regulation (VIN = 3.6V; VOUT = 1.8V; L = 2.2μH) Load Current (mA) Output Voltage Error (%) -1.0 -0.8 -0.6 -0.4 -0.2 0.2 0.4 0.6 0.8 1.0 1 10 100 1000

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT2514 Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Typical Characteristics Frequency vs. Input Voltage (VIN = 3.6V; VOUT = 1.8V; ILOAD = 150mA; L = 2.2μH) Input Voltage (V) Switching Frequency (MHz)1.46 1.48 1.50 1.52 1.54 1.56 1.58 Frequency vs. Temperature (VIN = 3.6V; VOUT = 1.8V; ILOAD = 150mA; L = 2.2μH) Temperature (°C) Switching Frequency (MHz) -40 -10 20 50 65 1.2 1.3 1.4 1.5 1.6 1.7 1.8 80-25 5 35 RDS(ON) vs. Input Voltage (TA = 25ºC) Input Voltage (V) RDS(ON) (Ω) 0.20 0.25 0.30 0.35 0.40 0.45 Main switch Synchronous switch VFB vs. Temperature (VIN = 3.6V; VOUT = 1.8V; ILOAD = 0mA) Temperature (°°C) Voltage (V) -45 -30 -15 0 15 30 45 60 75 9 0 0.588 0.591 0.594 0.597 0.600 0.603 0.606 0.609 0.612 Load Transient Response (Light Load Mode to PWM Mode; L = 2.2µH; CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V) Time (20µs/div) VSW (2V/div) VOUT (200mV/div) IOUT (400mA/div) Load Transient Response (PWM Mode Only; ILOAD = 180mA to 400mA; L = 2.2µH; CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V) Time (20µs/div) VSW (2V/div) VOUT (200mV/div) IOUT (500mA/div)

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT2514 Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Functional Description The AAT2514 is a dual high performance 600mA, 1.5MHz fixed frequency monolithic switch-mode step-down con- verter which uses current mode architecture with an adaptive slope compensation scheme. It minimizes exter- nal component size and optimizes efficiency over the com- plete load range. The adaptive slope compensation allows the device to remain stable over a wider range of inductor values so that smaller values (1μH to 4.7μH) with associ- ated lower DCR can be used to achieve higher efficiency. Apart from the small bypass input capacitor, only a small L-C filter is required at each output. The adjustable out- puts can be programmed with external feedback to any voltage, ranging from very low output voltages to the input voltage and by using an internal reference of 0.6V. The part uses internal MOSFETs for each channel to achieve high efficiency. At dropout, the converter duty cycle increases to 100% and the output voltages track the input voltage minus the low R DS(ON) drop of the P-channel high-side MOSFETs. The converter efficiency has been optimized for all load conditions, ranging from no load to 600mA at V IN = 3V with an input voltage range from 2.5V to 5.5V. The internal error amplifier and com- pensation provides excellent transient response, load, and line regulation. Internal soft start eliminates any output voltage overshoot when the enable or the input voltage is applied. Functional Block Diagram EA ICOMP - PWM Logic R Q S _Q POR Counter ISENSE AMP 10 7REGULATOR 2 (Same as Regulator 1) FB1 EN1 EN2 FB2 IN LX1 POR LX2 Bandgap Reference OSC Slope Comp OVDET -650mV Regulator 1 600mV IZERO COMP Overtemperature and Shortcircuit Protection Non- Overlap Control

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Current Mode PWM Control Slope compensated current mode PWM control provides stable switching and cycle-by-cycle current limit for excellent load and line response and protection of the internal main switch (P-channel MOSFET) and synchro- nous rectifier (N-channel MOSFET). During normal opera- tion, the internal P-channel MOSFET is turned on for a specified time to ramp the inductor current at each rising edge of the internal oscillator, and is switched off when the peak inductor current is above the error voltage. The current comparator, I COMP, limits the peak inductor cur- rent. When the main switch is off, the synchronous recti- fier turns on immediately and stays on until either the inductor current starts to reverse, as indicated by the current reversal comparator, I ZERO, or the beginning of the next clock cycle. The OVDET comparator controls output transient overshoot by turning the main switch off and keeping it off until the fault is no longer present. Control Loop The AAT2514 is a peak current mode step-down converter. The current through the P-channel MOSFET (high side) is sensed for current loop control, as well as short circuit and overload protection. An adaptive slope compensation sig- nal is added to the sensed current to maintain stability for duty cycles greater than 50%. The peak current mode loop appears as a voltage-programmed current source in parallel with the output capacitor. The output of the volt- age error amplifier programs the current mode loop for the necessary peak switch current to force a constant output voltage for all load and line conditions. Internal loop compensation terminates the transconductance volt- age error amplifier output. For fixed voltage versions, the error amplifier reference voltage is internally set to pro- gram the converter output voltage. For the adjustable output, the error amplifier reference is fixed at 0.6V. Enable The enable pins are active high. When pulled low, the enable input forces the AAT2514 into a low-power, non- switching state. The total input current during shutdown is less than 2μA. Current Limit and Over-Temperature Protection For overload conditions, the peak input current is limit- ed. To minimize power dissipation and stresses under current limit and short-circuit conditions, switching is terminated after entering current limit for a series of pulses. Switching is terminated for seven consecutive clock cycles after a current limit has been sensed for a series of four consecutive clock cycles. Thermal protec- tion completely disables switching when internal dissipa- tion becomes excessive. The junction over-temperature threshold is 140°C with 15°C of hysteresis. Once an over-temperature or over-current fault conditions is removed, the output voltage automatically recovers. Dropout Operation When the input voltage decreases toward the value of the output voltage, the AAT2514 allows the main switch to remain on for more than one switching cycle and increases the duty cycle until it reaches 100%. The duty cycle D of a step-down converter is defined as: D = TON · FOSC · 100% ≈ VOUT VIN

• 100% Where TON is the main switch on time and F OSC is the oscillator frequency (1.5MHz). The output voltage then is the input voltage minus the voltage drop across the main switch and the inductor. At low input supply voltage, the R DS(ON) of the P-channel MOSFET increases and the efficiency of the converter decreases. Caution must be exercised to ensure the heat dissipated does not exceed the maximum junction tem- perature of the IC. Maximum Load Current The AAT2514 will operate with an input supply voltage as low as 2.5V; however, the maximum load current decreases at lower input due to the large IR drop on the main switch and synchronous rectifier. The slope com- pensation signal reduces the peak inductor current as a function of the duty cycle to prevent sub-harmonic oscil- lations at duty cycles greater than 50%. Conversely, the current limit increases as the duty cycle decreases.

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET For output voltages above 2.0V, when light-load efficiency is important, the minimum recommended inductor size is 2.2μH. For optimum voltage-positioning load transients, choose an inductor with DC series resistance in the 50mΩ to 150m Ω range. For higher efficiency at heavy loads (above 200mA), or minimal load regulation (with some transient overshoot), the resistance should be kept below 100mΩ. The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation (600mA + 105mA). Table 2 lists some typical surface mount induc- tors that meet target applications for the AAT2514. Manufacturer's specifications list both the inductor DC cur- rent rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation char- acteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive losses due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. For exam- ple, the 2.2μH CR43 series inductor selected from Sumida has a 71.2mΩ DCR and a 1.75ADC current rating. At full load, the inductor DC loss is 25mW, which gives a 2.8% loss in efficiency for a 600mA, 1.5V output. Slope Compensation The AAT2514 step-down converter uses peak current mode control with a unique adaptive slope compensation scheme to maintain stability with lower value inductors for duty cycles greater than 50%. Using lower value inductors provides better overall efficiency and also makes it easier to standardize on one inductor for different required out- put voltage levels. In order to do this and keep the step- down converter stable when the duty cycle is greater than 50%, the AAT2514 separates the slope compensation into 2 phases. The required slope compensation is automati- cally detected by an internal circuit using the feedback voltage V FB before the error amp comparison to VREF. Error AmpVREF VFB When below 50% duty cycle, the slope compensation is 0.284A/μs; but when above 50% duty cycle, the slope compensation is set to 1.136A/ μs. The output inductor value must be selected so the inductor current down slope meets the internal slope compensation requirements. Below 50% duty cycle, the slope compensation require- ment is: 1.25 2 · Lm = = 0.284A/µs Therefore: 0.625 mL = = 2.2µH Above 50% duty cycle, 2 · Lm = = 1.136A/µs Part L ( μH) Max DCR (m Ω) Rated DC Current (A) Size WxLxH (mm) CDRH2D11/HP 1.5 80 1.35 3.3 173 0.9 4.7 238 0.75 Sumida CDRH4D18 1.0 45 1.72 3.3 110 1.04 4.7 162 0.84 Toko D312C 1.5 120 1.29 3.3 180 0.98 4.7 240 0.79 Table 2: Typical Surface Mount Inductors.

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Therefore: 2.5 mL = = 2.2µH With these adaptive settings, a 2.2 μH inductor can be used for all output voltages from 0.6V to 5V. Input Capacitor Selection The input capacitor reduces the surge current drawn from the input and switching noise from the device. The input capacitor impedance at the switching frequency shall be less than the input source impedance to prevent high frequency switching current passing to the input. The calculated value varies with input voltage and is a maximum when V IN is double the output voltage. VO VIN CIN = VO VIN VPP IO VO VIN VO VIN CIN(MIN) = 1 VPP IO A low ESR input capacitor sized for maximum RMS cur- rent must be used. Ceramic capacitors with X5R or X7R dielectrics are highly recommended because of their low ESR and small temperature coefficients. A 22μF ceramic capacitor for most applications is sufficient. A large value may be used for improved input voltage filtering. The maximum input capacitor RMS current is: VO VIN VO VIN The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current VO VIN VO VIN IO RMS(MAX)I 2= To minimize stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C1) can be seen in the evaluation board layout in Figure 3. A laboratory test set- up typically consists of two long wires running from the bench power supply to the evaluation board input voltage pins. The inductance of these wires, along with the low- ESR ceramic input capacitor, can create a high Q net-work that may affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage during load transients. Errors in the loop phase and gain measurements can also result. Since the inductance of a short PCB trace feeding the input volt- age is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. In applications where the input power source lead inductance cannot be reduced to a level that does not affect the converter performance, a high ESR tanta- lum or aluminum electrolytic should be placed in parallel with the low ESR, ESL bypass ceramic. This dampens the high Q network and stabilizes the system. Output Capacitor Selection The function of output capacitance is to store energy to attempt to maintain a constant voltage. The energy is stored in the capacitor’s electric field due to the voltage applied. The value of output capacitance is generally selected to limit output voltage ripple to the level required by the specification. Since the ripple current in the output induc- tor is usually determined by L, V OUT, and V IN, the series impedance of the capacitor primarily determines the out- put voltage ripple. The three elements of the capacitor that contribute to its impedance (and output voltage ripple) are equivalent series resistance (ESR), equivalent series inductance (ESL), and capacitance (C). The out- put voltage droop due to a load transient is dominated by the capacitance of the ceramic output capacitor. During a step increase in load current, the ceramic output capaci- tor alone supplies the load current until the loop responds. Within two or three switching cycles, the loop responds and the inductor current increases to match the load current demand. The relationship of the output volt- age droop during the three switching cycles to the output capacitance can be estimated by: COUT = 3 · ΔILOAD VDROOP · FS

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET In many practical designs, to get the required ESR, a capacitor with much more capacitance than is needed must be selected. For both continuous or discontinuous inductor current mode operation, the ESR of the C OUT needed to limit the ripple to ∆VO, V peak-to-peak is: ΔVO ΔIL ESR ≤ Ripple current flowing through a capacitor’s ESR causes power dissipation in the capacitor. This power dissipation causes a temperature increase internal to the capacitor. Excessive temperature can seriously shorten the expect- ed life of a capacitor. Capacitors have ripple current rat- ings that are dependent on ambient temperature and should not be exceeded. The output capacitor ripple cur- rent is the inductor current, I L, minus the output current, IO. The RMS value of the ripple current flowing in the output capacitance (continuous inductor current mode operation) is given by: 6IRMS = ΔIL · ΔIL · 0.289 ESL can be a problem by causing ringing in the low megahertz region but can be controlled by choosing low ESL capacitors, limiting lead length (PCB and capacitor), and replacing one large device with several smaller ones connected in parallel. In conclusion, in order to meet the requirement of out- put voltage ripple small and regulation loop stability, ceramic capacitors with X5R or X7R dielectrics are rec- ommended due to their low ESR and high ripple current ratings. The output ripple V OUT is determined by: VOUT · (VIN - VOUT) VIN · fOSC · L 8 · fOSC · C3 A 10μF ceramic capacitor can satisfy most applications. Thermal Calculations There are three types of losses associated with the AAT2514 step-down converter: switching losses, conduc- tion losses, and quiescent current losses. Conduction losses are associated with the R DS(ON) characteristics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switch- ing devices. At full load, assuming continuous conduction mode(CCM), a simplified form of the losses is given by: PTOTAL IO 2 · (RDSON(HS) · VO + RDSON(LS) · [VIN - VO]) VIN + (tsw · F · IO + IQ) · VIN IQ is the step-down converter quiescent current. The term tsw is used to estimate the full load step-down con- verter switching losses. For the condition where the step-down converter is in dropout at 100% duty cycle, the total device dissipation reduces to: PTOTAL = IO 2 · RDSON(HS) + IQ · VIN Since RDS(ON), quiescent current, and switching losses all vary with input voltage, the total losses should be inves- tigated over the complete input voltage range. Given the total losses, the maximum junction temperature can be derived from the θ JA for the MSOP-10 or DFN-10 pack- ages, which is 45°C/W. TJ(MAX) = PTOTAL · ΘJA + TAMB Manufacturer Part Number Value Voltage (V) Temp. Co. Case Murata GRM219R60J106KE19 10 μF 6.3 X5R 0805 Murata GRM21BR60J226ME39 22 μF 6.3 X5R 0805 Murata GRM1551X1E220JZ01B 22pF 25 JIS 0402 Table 3: Typical Surface Mount Capacitors.

Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT2514 Dual Channel 600mA Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Advanced Analogic Technologies, Inc.

3230 Scott Boulevard, Santa Clara, CA 95054

Phone (408) 737-4600 Fax (408) 737-4611 © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech pr oduct. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifi cations or to discontinue any product or service without notice. Except as provided in AnalogicTech’s terms and conditions of sale, AnalogicTech assumes no liability whatsoever, and AnalogicTech disclaims any express or implied warranty relating to the sale and/or use of AnalogicTech products including liability or warranties relating to fi tness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. In order to minimize risks associated with the customer’s applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. Testing and other quality control techniques are utilized to the extent AnalogicTech deems necessary to support this warranty. Specifi c testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holders.

Ordering Information

Output Voltage1 Package Marking 2 Part Number (Tape & Reel)3 Adj. 0.6V to VIN TDFN33-10 ZBXYY AAT2514IDE-AA-T1 All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more information, please visit our website at http://www.analogictech.com/about/quality.aspx. Package Information4 TDFN33-10 Top View 3.00 ± 0.05 3.00 ± 0.05 Pin 1 dot by marking Pin 1 identification R0.200 Bottom View 1.70 ± 0.05 0.500 BSC0.23 ± 0.05 0.40 ± 0.05 2.40 ± 0.05 Side View 0.05 ± 0.05

0.203 REF

0.75 ± 0.05 All dimensions in millimeters. 1. Please contact Sales for other voltage options. 2. XYY = assembly and date code. 3. Sample stock is generally held on part numbers listed in BOLD. 4. The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the lead terminals due to the manufacturing process. A solder fillet at the exposed copper edge cannot be guaranteed and is not required to ensure a proper bottom solder connection.