AAT2514

AAT2514 ANALOGICTECH | Alldatasheet

Document overview

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Technical content

Features

• V IN Range:2.5Vto5.5V
• Upto600mAOutputCurrent
• HighEfficiency:Upto96%
• 1.5MHzConstantFrequencyOperation
• 100%DutyCycleDropoutOperation
• LowR DS(ON) InternalSwitches:0.35Ω
• CurrentModeOperationforExcellentLine andLoadTransientResponse
• AdaptiveSlopeCompensation
• SoftStart
• Short-CircuitandThermalFaultProtection
• <1µAShutdownCurrent
• Power-OnResetOutput
• Small,ThermallyEnhancedTDFN33-10 Package
• -40°Cto+85°CTemperatureRange

Applications

• CellularTelephones
• DigitalStillCameras
• PDAs
• PortableMediaPlayers
• WirelessandDSLModems TypicalApplication EN1 LX1 GND AAT2514 L1 2.2µH VOUT1 1.8V 10µF 10µF VIN 2.5V to 5.5V IN EN2 LX 2 FB2 FB1 VOUT2 2.5V 100kΩ RESET L2 2.2µH 316kΩ 634kΩR3 316kΩ 1MΩ 10µF POR

Dual Channel 600mAStep-Down Converter 2 2514.2007.06.1.0 Pin Descriptions Pin Configuration AAT2514-IDE TDFN33-10 (Top View) 10-Lead (3mm X 3mm) Plastic Thin DFN Exposed Pad is PGND Must be connected to GND. EXPOSED PAD FB1 EN1 IN LX1 GND FB2 EN2 POR LX2 NC Pin # Symbol Function 1 FB1 Feedbackinputforchannel1.ConnectFB1tothecenterpointofanexternalresistordivider. Thefeedbackthresholdvoltageis0.6V. 2 EN1 Channel1enablepin.Activehigh.Inshutdown,allfunctionsaredisableddrawing<1µAsup- plycurrent.DonotleaveEN1floating. 3 IN Powersupplyinputpin. MustbecloselydecoupledtoGNDwitha2.2µForgreaterceramic capacitor. 4 LX1 Channel1switchingnodepin.Connecttheoutputinductortothispin.

5 GND Ground

6 N/C Noconnection

8 POR Power-onreset,activelow.Opendrain.Externalresistor(100kΩ)isrequired. 9 EN2 Channel2enablepin.Activehigh. Inshutdown,allfunctionsaredisableddrawing<1µA supplycurrent.DonotleaveEN2floating. 10 FB2 Feedbackinputforchannel2.ConnectFB2tothecenterpointofanexternalresistordivider. Thefeedbackthresholdvoltageis0.6V. EP Exposedpaddle.TheexposedpaddleshouldbeconnectedtoboardgroundplaneandGND. Thegroundplaneshouldincludealargeexposedcopperpadunderthepackageforthermal dissipation(seepackageoutline).

Dual Channel 600mAStep-Down Converter 2514.2007.06.1.0 3 Absolute Maximum Ratings1 Recommended Operating Conditions Symbol Description Value Units θJA ThermalResistance3 45 °C/W PD MaximumPowerDissipationatT A =25°C 2.2 W Symbol Description Value Units VIN InputSupplyVoltage -0.3to+6.0 V VEN1,VEN2 EN1,EN2Voltages -0.3toV IN +0.3 V VFB1,VFB2 FB1,FB2Voltages -0.3toV IN +0.3 V VLX1,VLX2 LX1,LX2Voltages -0.3toV IN +0.3 V VPOR PORVoltage -0.3to+6.0 V TA OperatingTemperatureRange 2 -40to+85 °C TJ JunctionTemperature2 +125 °C TSTORAGE StorageTemperatureRange -65to+150 °C TLEAD LeadTemperature(Soldering,10s) +300 °C 1.AbsoluteMaximumRatingsarethosevaluesbeyondwhichthelifeofadevicemaybeimpaired. 2.TJ iscalculatedfromtheambienttemperatureT A andpowerdissipationP D accordingtothefollowingformula:T J =TA +PD xθJA. 3.Thermalresistanceisspecifiedwithapproximately1squareinchof1ozcopper.

Dual Channel 600mAStep-Down Converter 4 2514.2007.06.1.0

Electrical Characteristics

VIN =VEN =3.6V,T A =25°C,unlessotherwisenoted. Symbol Description Conditions Min Typ Max Units Step-Down Converter VIN InputVoltageRange 2.5 5.5 V IQ InputDCSupplyCurrent ActiveMode,V FB =0.5V 500 800 µAShutdownMode,EN1=EN2=0V,V IN =4.2V 0.3 2.0 TA =25°C,Channel1or2 0.5880 0.6000 0.6120 VFB IFB FBInputBiasCurrent -30 30 nA ∆VOUT/ OutputVoltageLine VIN =2.5Vto5.5V,I OUT =10mA 0.11 0.40 %/VVOUT/∆VIN Regulation ∆VOUT/ OutputVoltageLoad IOUT =10mAto600mA 0.0015 %/mAVOUT/∆IOUT Regulation ILIM MaximumOutputCurrent V IN =3.0V 600 mA TS StartupTime FromEnabletoOutput 100 µsRegulation TSD Over-Temperature 140 °CShutdownThreshold THYS Over-Temperature 15 °CShutdownHysteresis FOSC OscillatorFrequency V FB =0.6V 1.2 1.5 1.8 MHz RDS(ON) P-ChannelMOSFET I LX =300mA 0.35 0.45 ΩN-ChannelMOSFET I LX =300mA 0.28 0.45 PeakInductorCurrent V IN =3V,V FB =0.5V;DutyCycle<35% 1.20 A VEN(L) EnableThresholdLow 0.3 V VEN(H) EnableThresholdHigh 1.5 V IEN ENInputCurrent -1.0 1.0 µA VFB RampingUp 8.5 % Power-OnReset VFB RampingDown -8.5 % Threshold(POR) Power-OnResetDelay 175 ms Power-OnResetOn-Resistance 100 Ω 1.Specificationsoverthetemperaturerangeareguaranteedbydesignandcharacterization. 2.TheregulatedfeedbackvoltageistestedinaninternaltestmodethatconnectsV FB totheoutputoftheerroramplifier.

Dual Channel 600mAStep-Down Converter 2514.2007.06.1.0 5 Typical Characteristics Efficiency vs. Input Voltage (VOUT = 1.8V; TA = 25ºC) Input Voltage (V) Efficiency (%) 100 ILOAD = 600mA Load Regulation (VIN = 3.6V; VOUT = 1.8V; L = 2.2µH) Load Current (mA) Output Voltage Error (%) -1.0 -0.8 -0.6 -0.4 -0.2 0.2 0.4 0.6 0.8 1.0 1 10 100 1000 Efficiency vs. Load Current (VOUT = 1.5V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.3V VIN = 4.2V Efficiency vs. Load Current (VOUT = 1.2V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.3V VIN = 4.2V Efficiency vs. Load Current (VOUT = 2.5V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 4.2V VIN = 3.3V VIN = 2.7V Efficiency vs. Load Current (VOUT = 1.8V; TA = 25°C) Load Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.3V VIN = 4.2V

Dual Channel 600mAStep-Down Converter 6 2514.2007.06.1.0 Typical Characteristics Load Transient Response (Light Load Mode to PWM Mode; L = 2.2µH; CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V) Time (20µs/div) VSW (2V/div) VOUT (200mV/div) IOUT (400mA/div) Load Transient Response (PWM Mode Only; ILOAD = 180mA to 400mA; L = 2.2µH; CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V) Time (20µs/div) VSW (2V/div) VOUT (200mV/div) IOUT (500mA/div) RDS(ON) vs. Input Voltage (TA = 25ºC) Input Voltage (V) RDS(ON) (Ω) 0.20 0.25 0.30 0.35 0.40 0.45 Main switch Synchronous switch VFB vs. Temperature (VIN = 3.6V; VOUT = 1.8V; ILOAD = 0mA) Temperature (°°C) Voltage (V) -45 -30 -15 0 15 30 45 60 75 90 0.588 0.591 0.594 0.597 0.600 0.603 0.606 0.609 0.612 Frequency vs. Input Voltage (VIN = 3.6V; VOUT = 1.8V; ILOAD = 150mA; L = 2.2µH) Input Voltage (V) Switching Frequency (MHz)1.46 1.48 1.50 1.52 1.54 1.56 1.58 Frequency vs. Temperature (VIN = 3.6V; VOUT = 1.8V; ILOAD = 150mA; L = 2.2µH) Temperature (°C) Switching Frequency (MHz) -40 -10 20 50 65 1.2 1.3 1.4 1.5 1.6 1.7 1.8 80-25 5 35

Dual Channel 600mAStep-Down Converter 2514.2007.06.1.0 7 Functional Block Diagram EA ICOMP - PWM Logic R Q S _Q POR Counter ISENSE AMP 10 7REGULATOR 2 (Same as Regulator 1) FB1 EN1 EN2 FB2 IN LX1 POR LX2 Bandgap Reference OSC Slope Comp OVDET -650mV Regulator 1 600mV IZERO COMP Overtemperature and Shortcircuit Protection Non- Overlap Control Functional Description The AAT2514 is a dual high performance 600mA, 1.5MHz fixed frequency monolithic switch-mode step-down converter which uses current mode architecture with an adaptive slope compensation scheme.Itminimizesexternalcomponentsizeand optimizes efficiency over the complete load range. Theadaptiveslopecompensationallowsthedevice toremainstableoverawiderrangeofinductorval- uessothatsmallervalues(1µHto4.7µH)withasso- ciated lower DCR can be used to achieve higher efficiency. Apartfromthesmallbypassinputcapacitor,onlya small L-C filter is required at each output. The adjustable outputs can be programmed with exter- nalfeedbacktoanyvoltage,rangingfromverylow outputvoltagestotheinputvoltageandbyusingan internal reference of 0.6V. The part uses internal MOSFETsforeachchanneltoachievehighefficien- cy.Atdropout,theconverterdutycycleincreasesto 100%andtheoutputvoltagestracktheinputvoltage minus the low R DS(ON) drop of the P-channel high- side MOSFETs. The converter efficiency has been optimized for all load conditions, ranging from no load to 600mA at V IN = 3V with an input voltage rangefrom2.5Vto5.5V.Theinternalerroramplifier and compensation provides excellent transient response,load,andlineregulation.Internalsoftstart eliminates any output voltage overshoot when the enableortheinputvoltageisapplied.

Dual Channel 600mAStep-Down Converter 8 2514.2007.06.1.0 Current Mode PWM Control Slope compensated current mode PWM control provides stable switching and cycle-by-cycle cur- rent limit for excellent load and line response and protection of the internal main switch (P-channel MOSFET) and synchronous rectifier (N-channel MOSFET).Duringnormaloperation,theinternalP- channel MOSFETis turned on for a specified time toramptheinductorcurrentateachrisingedgeof theinternaloscillator,andisswitchedoffwhenthe peak inductor current is above the error voltage. The current comparator, I COMP, limits the peak inductor current. When the main switch is off, the synchronous rectifier turns on immediately and stays on until either the inductor current starts to reverse, as indicated by the current reversal com- parator, I ZERO, or the beginning of the next clock cycle.TheOVDETcomparatorcontrolsoutputtran- sient overshoot by turning the main switch off and keepingitoffuntilthefaultisnolongerpresent. Control Loop The AAT2514 is a peak current mode step-down converter. The current through the P-channel MOSFET (high side) is sensed for current loop control, as well as short circuit and overload pro- tection.An adaptive slope compensation signal is addedtothesensedcurrenttomaintainstabilityfor duty cycles greater than 50%. The peak current modeloopappearsasavoltage-programmedcur- rent source in parallel with the output capacitor. The output of the voltage error amplifier programs the current mode loop for the necessary peak switchcurrenttoforceaconstantoutputvoltagefor all load and line conditions. Internal loop compen- sation terminates the transconductance voltage error amplifier output. For fixed voltage versions, theerroramplifierreferencevoltageisinternallyset to program the converter output voltage. For the adjustable output, the error amplifier reference is fixedat0.6V. Enable The enable pins are active high. When pulled low, the enable input forces the AAT2514 into a low- power, non-switching state. The total input current duringshutdownislessthan2µA. Current Limit and Over-Temperature Protection. For overload conditions, the peak input current is limited.Tominimizepowerdissipationandstresses under current limit and short-circuit conditions, switching is terminated after entering current limit for a series of pulses. Switching is terminated for sevenconsecutiveclockcyclesafteracurrentlimit has been sensed for a series of four consecutive clock cycles. Thermal protection completely dis- ables switching when internal dissipation becomes excessive.Thejunctionover-temperaturethreshold is 140°C with 15°C of hysteresis. Once an over- temperature or over-current fault conditions is removed,theoutputvoltageautomaticallyrecovers. Dropout Operation Whentheinputvoltagedecreasestowardthevalue of the output voltage, theAAT2514 allows the main switch to remain on for more than one switching cycle and increases the duty cycle until it reaches 100%. The duty cycle D of a step-down converter is definedas: WhereT ON isthemainswitchontimeandF OSC is theoscillatorfrequency(1.5MHz). The output voltage then is the input voltage minus the voltage drop across the main switch and the inductor.Atlowinputsupplyvoltage,theR DS(ON) of theP-channelMOSFETincreasesandtheefficien- cy of the converter decreases. Caution must be exercised to ensure the heat dissipated does not exceedthemaximumjunctiontemperatureoftheIC. Maximum Load Current TheAAT2514willoperatewithaninputsupplyvolt- age as low as 2.5V; however, the maximum load currentdecreasesatlowerinputduetothelargeIR droponthemainswitchandsynchronousrectifier. The slope compensation signal reduces the peak inductor current as a function of the duty cycle to prevent sub-harmonic oscillations at duty cycles greater than 50%. Conversely, the current limit increasesasthedutycycledecreases. D = TON · FOSC · 100% ≈ VOUT VIN

• 100%

Dual Channel 600mAStep-Down Converter 2514.2007.06.1.0 9 Applications Information Setting the Output Voltage Figure1showsthebasicapplicationcircuitforthe AAT2514. Resistors R1 and R3 and R2 and R4 program the output to regulate at a voltage higher than0.6V.To limitthe biascurrent requiredforthe external feedback resistor string while maintaining goodnoiseimmunity,theminimumsuggestedvalue forR1andR3is59kΩ.Althoughalargervaluewill further reduce quiescent current, it will also increasetheimpedanceofthefeedbacknode,mak- ing it more sensitive to external noise and interfer- ence. Table 1 summarizes the resistor values for variousoutputvoltageswithR1andR3settoeither 59kΩforgoodnoiseimmunityor316kΩforreduced noloadinputcurrent. The adjustable feedback resistors, combined with a external feed forward capacitors (C4 and C5 in Figure1),deliverenhancedtransientresponsefor extreme pulsed load applications. The addition of thefeedforwardcapacitortypicallyrequiresalarg- er output capacitor C2 and C3 for stability. The external resistor sets the output voltage according tothefollowingequation: Table 1: Resistor Selection for Output Voltage Setting; Standard 1% Resistor Values Substituted Closest to the Calculated Values. R1, R3 = 59kΩΩ R1 , R3 = 316k ΩΩ VOUT (V) R2, R4 (kΩΩ) R 2, R4 (k ΩΩ) 0.8 19.6 105 0.9 29.4 158 1.0 39.2 210 1.1 49.9 261 1.2 59.0 316 1.3 68.1 365 1.4 78.7 422 1.5 88.7 475 1.8 118 634 1.85 124 655 2.0 137 732 2.5 187 1000 3.3 267 1430 = 1 21 + 6 . 0R RV ·V or OUT 

• R1 - 1VOUT VREF R2 = Figure 1: AAT2514 Typical Application Circuit. EN1 LX1 GND AAT2514 L1 2.2µH VOUT1 1.8V 10µF 10µF VIN 2.5V to 5.5V IN EN2 LX2 FB2 FB1 VOUT2 2.5V C5 22pF C4 22pF 100kΩ RESET L2 2.2µH 316kΩ 634kΩR3 316kΩ 1MΩ 10µF POR

Dual Channel 600mA Step-Down Converter 10 2514.2007.06.1.0 Table 2: Typical Surface Mount Inductors. Part L (µH) Max DCR (m ΩΩ) R ated DC Current (A) Size WxLxH (mm) 1.5 80 1.35 4.7 238 0.75 1.0 45 1.72 4.7 162 0.84 1.5 120 1.29 4.7 240 0.79 Inductor Selection For most designs, the AAT2514 operates with inductor values of 1µH to 4.7µH. Low inductance values are physically smaller, but require faster switching, which results in some efficiency loss. The inductor value can be derived from the follow- ing equation: Where ∆I L is inductor ripple current. Large value inductors lower ripple current and small value inductors result in high ripple currents. Choose inductor ripple current approximately 35% of the maximum load current 600mA, or ∆I L = 210mA. For output voltages above 2.0V, when light-load efficiency is important, the minimum recommended inductor size is 2.2µH. For optimum voltage-posi- tioning load transients, choose an inductor with DC series resistance in the 50mΩ to 150mΩ range. For higher efficiency at heavy loads (above 200mA), or minimal load regulation (with some transient over- shoot), the resistance should be kept below 100mΩ. The DC current rating of the inductor should be at least equal to the maximum load cur- rent plus half the ripple current to prevent core sat- uration (600mA + 105mA). Table 2 lists some typi- cal surface mount inductors that meet target appli- cations for the AAT2514. Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation characteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive loss- es due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. For example, the 2.2µH CR43 series inductor selected from Sumida has a 71.2mΩ DCR and a 1.75ADC current rating. At full load, the inductor DC loss is 25mW which gives a 2.8% loss in efficiency for a 600mA, 1.5V output. Slope Compensation The AAT2514 step-down converter uses peak cur- rent mode control with a unique adaptive slope compensation scheme to maintain stability with lower value inductors for duty cycles greater than 50%. Using lower value inductors provides better overall efficiency and also makes it easier to stan- dardize on one inductor for different required output voltage levels. In order to do this and keep the step-down converter stable when the duty cycle is greater than 50%, the AAT2514 separates the slope compensation into 2 phases. The required slope compensation is automatically detected by an internal circuit using the feedback voltage V FB before the error amp comparison to VREF. L = VOUT · (VIN - VOUT) VIN · ∆IL · fOSC

Dual Channel 600mA Step-Down Converter 2514.2007.06.1.0 11 When below 50% duty cycle, the slope compensa- tion is 0.284A/µs; but when above 50% duty cycle, the slope compensation is set to 1.136A/µs. The output inductor value must be selected so the inductor current down slope meets the internal slope compensation requirements. Below 50% duty cycle, the slope compensation requirement is: Therefore: Above 50% duty cycle, Therefore: With these adaptive settings, a 2.2µH inductor can be used for all output voltages from 0.6V to 5V. Input Capacitor Selection The input capacitor reduces the surge current drawn from the input and switching noise from the device. The input capacitor impedance at the switching fre- quency shall be less than the input source imped- ance to prevent high frequency switching current passing to the input. The calculated value varies with input voltage and is a maximum when V IN is double the output voltage. A low ESR input capacitor sized for maximum RMS current must be used. Ceramic capacitors with X5R or X7R dielectrics are highly recommended because of their low ESR and small temperature coefficients. A 22µF ceramic capacitor for most applications is sufficient. A large value may be used for improved input voltage filtering. The maximum input capacitor RMS current is: The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current To minimize stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C1) can be seen in the evaluation board layout in Figure 3. A laboratory test set-up typically consists of two long wires running from the bench power supply to the evaluation board input voltage pins. The inductance of these wires, along with the low-ESR ceramic input capacitor, can create a high Q net-work that may IO RMS(MAX)I 2= VO VIN VO VIN VO VIN VO VIN CIN(MIN) = 1 VPP IO VO VIN VO VIN VO VIN CIN = VO VIN VPP IO 2.5 mL = = 2.2µH 2 · Lm = = 1.136A/µs 0.625 mL = = 2.2µH 1.25 2 · Lm = = 0.284A/µs Error AmpVREF VFB

Dual Channel 600mA Step-Down Converter 12 2514.2007.06.1.0 affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage during load transients. Errors in the loop phase and gain measurements can also result. Since the inductance of a short PCB trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. In applica- tions where the input power source lead inductance cannot be reduced to a level that does not affect the converter performance, a high ESR tantalum or alu- minum electrolytic should be placed in parallel with the low ESR, ESL bypass ceramic. This dampens the high Q network and stabilizes the system. Output Capacitor Selection The function of output capacitance is to store ener- gy to attempt to maintain a constant voltage. The energy is stored in the capacitor's electric field due to the voltage applied. The value of output capacitance is generally select- ed to limit output voltage ripple to the level required by the specification. Since the ripple current in the output inductor is usually determined by L, V OUT, and VIN, the series impedance of the capacitor pri- marily determines the output voltage ripple. The three elements of the capacitor that contribute to its impedance (and output voltage ripple) are equiva- lent series resistance (ESR), equivalent series inductance (ESL), and capacitance (C). The output voltage droop due to a load transient is dominated by the capacitance of the ceramic output capacitor. During a step increase in load current, the ceramic output capacitor alone supplies the load current until the loop responds. Within two or three switch- ing cycles, the loop responds and the inductor cur- rent increases to match the load current demand. The relationship of the output voltage droop during the three switching cycles to the output capaci- tance can be estimated by: In many practical designs, to get the required ESR, a capacitor with much more capacitance than is needed must be selected. For both continuous or discontinuous inductor current mode operation, the ESR of the C OUT needed to limit the ripple to ∆VO, V peak-to-peak is: Ripple current flowing through a capacitor's ESR causes power dissipation in the capacitor. This power dissipation causes a temperature increase internal to the capacitor. Excessive temperature can seriously shorten the expected life of a capacitor. Capacitors have ripple current ratings that are dependent on ambient temperature and should not be exceeded. The output capacitor ripple current is the inductor current, I L, minus the output current, IO. The RMS value of the ripple current flowing in the output capacitance (continuous inductor current mode operation) is given by: ESL can be a problem by causing ringing in the low megahertz region but can be controlled by choosing low ESL capacitors, limiting lead length (PCB and capacitor), and replacing one large device with sev- eral smaller ones connected in parallel. In conclusion, in order to meet the requirement of output voltage ripple small and regulation loop stabil- ity, ceramic capacitors with X5R or X7R dielectrics are recommended due to their low ESR and high rip- ple current ratings. The output ripple V OUT is deter- mined by: A 10µF ceramic capacitor can satisfy most applica- tions. ∆VOUT ≤ · ESR + VPP · (VIN - VOUT) VIN · fOSC · L 8 · fOSC · COUT 6IRMS = ∆IL · ∆IL · 0.289 ∆VO ∆IL ESR ≤ COUT = 3 · ∆ILOAD VDROOP · FS

a: Top Layer b: Bottom Layer Figure 2: AAT2514 Typical Application Circuit Layout. AAT2514 Dual Channel 600mA Step-Down Converter 14 2514.2007.06.1.0

Dual Channel 600mA Step-Down Converter 2514.2007.06.1.0 15

Ordering Information

3.00 ±± 0.05 3.00 ±± 0.05 Pin 1 dot by marking Pin 1 identification R0.200 Bottom View 1.70 ±± 0.05 0.500 BSC0.23 ±± 0.05 0.40 ±± 0.05 2.40 ±± 0.05 Side View 0.05 ±± 0.05

0.203 REF

0.75 ±± 0.05 All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” m eans semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more information, please visit our website at http://www.analogictech.com/pbfree. Output Voltage1 Package Marking 2 Part Number (Tape & Reel)3 Adj. 0.6V to VIN TDFN33-10 ZBXYY AAT2514IDE-AA-T1 1. Please contact Sales for other voltage options. 2. XYY = assembly and date code. 3. Sample stock is generally held on part numbers listed in BOLD. 4. The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the lead terminals due to the manufacturing process. A solder fillet at the exposed copper edge cannot be guaranteed and is not required to ensure a proper bottom solder connection.

Dual Channel 600mA Step-Down Converter 16 2514.2007.06.1.0 Advanced Analogic Technologies, Inc. 830 E. Arques Avenue, Sunnyvale, CA 94085 Phone (408) 737-4600 Fax (408) 737-4611 © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service with- out notice. Except as provided in AnalogicTech’s terms and conditions of sale, AnalogicTech assumes no liability whatsoever, and AnalogicTech disclaims any express or implied war- ranty relating to the sale and/or use of AnalogicTech products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. In order to minimize risks associated with the customer’s applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. Testing and other quality control techniques are utilized to the extent AnalogicTech deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holders.