AAT2511_0605 ANALOGICTECH | Alldatasheet

Document overview

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Technical content

Features

  • Up to 97% Efficiency
  • 25µA Quiescent Current Per Channel IN Range: 2.7V to 5.5V
  • V OUT Range: 0.6V to VIN
  • Output Current: 600mA
  • Low R DS(ON) 0.4Ω Integrated Power Switches
  • Low Drop Out 100% Duty Cycle
  • 1.0MHz Switching Frequency
  • Shutdown Current <1µA
  • Current Mode Operation
  • Internal Reference Soft Start
  • Short-Circuit Protection
  • Fast Turn-On (100µs Typical)
  • Over-Temperature Protection
  • 3mm x 3mm, < 1mm high
  • TDFN33-12 Package
  • -40°C to +85°C Temperature Range

Applications

  • Cellular Phones
  • Digital Cameras
  • Handheld Instruments
  • Microprocessor/DSP Core/IO Power
  • PDAs and Handheld Computers
  • Portable Media Players Typical Application AAT2511 Efficiency (VIN = 3.3V) Load Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 2.5V 1.8V 4.7μH 10μF 1.8V 118k 59.0k 10μH 10μF 2.5V 59.0k 10μF VIN 187k FB1 EN1 LX1 GND2 LX2 GND1 SGND1 VIN1 VIN2 SGND2 FB2 EN2 AAT2511 U1 0.1μF

(Top View) EN1 FB1 SGND1 EN2 FB2 SGND2 VIN1 LX1 GND1 VIN2 LX2 GND2 Pin # Symbol Function 1, 4 EN1, EN2 Converter enable input. A logic high enables the converter channel. A logic low forces the channel into shutdown mode, reducing the channel supply current to less than 1µA. This pin should not be left floating. When not actively controlled, this pin can be tied directly to the source voltage (VIN1, VIN2). 2, 5 FB1, FB2 Feedback input pin. An external resistive divider ties to this point and programs the output voltage to the desired value. 3, 6 SGND1, SGND2 Signal ground. Return the feedback resistive divider to this ground. See section on PCB layout guidelines and evaluation board layout diagram. 7, 10 GND2, GND1 Main power ground return. Connect to the input and output capacitor return. See sec- tion on PCB layout guidelines and evaluation board layout diagram. 8, 11 LX2, LX1 Output switching node that connects to the respective output inductor. 9, 12 VIN2, VIN1 Input supply voltage. Must be closely decoupled to the respective power gnd. EP Exposed paddle (bottom). Use properly sized vias for thermal coupling to the ground plane. See section on PCB layout guidelines. AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2 2511.2006.05.1.4

Symbol Description Value Units PD Maximum Power Dissipation 2 W θJA Thermal Resistance2 50 °C/W Symbol Description Value Units VIN VIN1, VIN2 to SGND1, SGND2, GND1, and GND2 6.0 V VLX LX1, LX2 to GND1, GND2 -0.3 to V P + 0.3 V VFB FB1 and FB2 to SGND1, SGND2, GND1, and GND2 -0.3 to V P + 0.3 V VEN EN1 and EN2 to SGND1, SGND2, GND1, and GND2 -0.3 to 6.0 V TJ Operating Junction Temperature Range -40 to 150 °C TLEAD Maximum Soldering Temperature (at leads, 10 sec) 300 °C AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 3 1. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at condi- tions other than the operating conditions specified is not implied. Only one Absolute Maximum Rating should be applied at any one time. 2. Mounted on an FR4 board with exposed paddle connected to ground plane.

Electrical Characteristics1 TA = -40°C to 85°C, unless otherwise noted. Typical values are T A = 25°C, VIN = 3.6V. Symbol Description Conditions Min Typ Max Units Step-Down Converter Channels VIN Input Voltage 2.7 5.5 V VIN Rising 2.6 V VUVLO UVLO Threshold Hysteresis 100 mV VIN Falling 1.8 V VOUT Output Voltage Tolerance I OUT = 0 to 400mA, VIN = 2.7 - 5.5V -3.0 +3.0 % VOUT Output Voltage Range 0.6 V IN V IQ Quiescent Current No Load, 0.6V Adjustable Version, 25 50 µAPer Channel ISHDN Shutdown Current EN = SGND = GND 1.0 µA ILIM P-Channel Current Limit 600 mA RDS(ON)H High Side Switch On Resistance 0.45 Ω RDS(ON)L Low Side Switch On Resistance 0.4 Ω ILXLK LX Leakage Current VIN = 5.5V, VLX = 0 to VIN, 1µ AEN = SGND = GND ΔVLinereg Line Regulation V IN = 2.7V to 5.5V 0.2 %/V VFB FB Threshold Voltage Accuracy 0.6V Output, No Load, T A = 25°C 591 600 609 mV TS Start-Up Time From Enable to Output Regulation 100 µs FOSC Oscillator Frequency T A = 25°C 0.7 1.0 1.5 MHz TSD Over-Temperature Shutdown 140 °CThreshold THYS Over-Temperature Shutdown 15 °CHysteresis EN VEN(L) Enable Threshold Low 0.6 V VEN(H) Enable Threshold High 1.4 V IEN Input Low Current V IN = VFB = 5.5V -1.0 1.0 µA AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 4 2511.2006.05.1.4 1. The AAT2511 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assu red by design, characterization, and correlation with statistical process controls. 2. For adjustable version with higher than 2.5V output, please consult your AnalogicTech representative.

Typical Channel Characteristics DC Regulation (VOUT = 1.5V; L = 4.7μμH) Output Current (mA) Output Error (%) -3.0 -2.0 -1.0 0.0 1.0 2.0 3.0 0.1 1 10 100 1000 VIN = 3.6V VIN = 4.2V VIN = 2.7V Efficiency vs. Load (VOUT = 1.5V; L = 4.7μμH) Output Current (mA) Efficiency (%) 6070 100 0.1 1 10 100 1000 VIN = 3.6V VIN = 4.2V VIN = 2.7V DC Regulation (VOUT = 2.5V; L = 10μμH) Output Current (mA) Output Error (%) -3.0 -2.0-1.0 0.0 1.0 2.03.0 0.1 1 10 100 1000 VIN = 3.6V VIN = 3.3V VIN = 3.0V Efficiency vs. Load (VOUT = 2.5V; L = 10μμH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 VIN = 3.0V VIN = 3.6V VIN = 3.3V DC Regulation (VOUT = 3.3V; L = 10μμH) Output Current (mA) Output Error (%) -3.0 -2.0 -1.0 0.0 1.0 2.0 3.0 0.1 1 10 100 1000 VIN = 4.2V VIN = 3.9V Efficiency vs. Load (VOUT = 3.3V; L = 10μμH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 VIN = 3.9V VIN = 4.2V AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 5

Typical Channel Characteristics N-Channel RDS(ON) vs. Input Voltage Input Voltage (V) RDS(ON) (mΩΩ) 300 350400 450 500 550600 650 700750 25°C 120°C 100°C 85°C P-Channel RDS(ON) vs. Input Voltage Input Voltage (V) RDS(ON) (mΩΩ) 300 350 400 450 500 550 600 650700750 25°C 120°C 100°C 85°C Quiescent Current vs. Input Voltage (VO = 1.8V) Input Voltage (V) Supply Current (μμA) 85°C 25°C -40°C Switching Frequency vs. Temperature (VIN = 3.6V; VO = 1.5V) Temperature (°°C) Variation (%) -0.20 -0.10 0.00 0.100.20 -40 -20 0 20 40 60 80 100 Output Voltage Error vs. Temperature (VIN = 3.6V; VO = 2.5V) Temperature (°°C) Output Error (%) -2.0 -1.5 -1.0-0.5 0.0 0.5 1.0 2.0 1.5 -40 -20 0 20 40 60 80 100 Frequency vs. Input Voltage (VOUT = 1.8V) Input Voltage (V) Frequency Variation (%) -2.0 -1.5 -1.0 -0.5 0.00.5 1.0 AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 6 2511.2006.05.1.4

Typical Channel Characteristics Soft Start (VIN = 3.6V; VOUT = 1.5V; L = 4.7μμH) Enable and Output Voltage (top) (V) Inductor Current (bottom) (A) Time (50μs/div) -4.0 -3.0 -2.0 -1.0 0.0 1.0 2.0 3.0 4.0 -0.5 0.0 0.5 1.01.5 2.0 2.5 3.0 3.5 Output Ripple (VIN = 3.6V; VOUT = 1.8V; 400mA) Output Voltage (AC Coupled) (top) (mV) Inductor Current (bottom) (A) Time (250ns/div) -120 -100 -80 -60 -40 -20 2040 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.80.9 Line Regulation (VOUT = 1.5V) Input Voltage (V) Accuracy (%) -0.5 0.5 1.5 2.5 3 3.5 4 4.5 5 5.5 6 IOUT = 600mA IOUT = 100mA IOUT = 10mA Line Transient (VOUT = 2.5V @ 500mA) Output Voltage (top) (V) Input Voltage (bottom) (V) Time (25μμs/div) 2.15 2.20 2.25 2.302.35 2.40 2.452.50 2.55 2.60 3.0 3.5 4.0 4.5 5.0 5.56.06.57.0 Load Transient Response (30mA - 300mA; VIN = 3.6V; VOUT = 2.5V; C1 = 22μμF) Output Voltage (top) (V) Load and Inductor Current (200mA/div) (bottom) Time (25μs/div) 2.05 2.15 2.25 2.35 2.45 2.55 2.65 -0.1 0.1 0.30.5 0.70.9 1.11.31.5 30mA 300mA Load Transient Response (30mA - 300mA; VIN = 3.6V; VOUT = 1.5V; C1 = 22μμF) Output Voltage (top) (V) Load and Inductor Current (200mA/div) (bottom) Time (25μs/div) 1.00 1.05 1.10 1.15 1.20 1.25 1.301.351.40 1.45 1.501.55 1.60 1.65 -0.1 0.1 0.3 0.5 0.7 0.9 1.1 1.3 1.5 30mA 300mA AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 7

Dual 600mA, 1MHz Step-Down DC/DC Converter 8 2511.2006.05.1.4 Functional Block Diagram Note: Internal resistor divider included for ≥1.2V versions. For low voltage versions, the feedback pin is tied directly to the error amplifier input. EN1 LX1 Err. Amp. DH DL GND1 VIN1FB1 SGND2 Voltage Reference Control Logic EN2 LX2 Err. Amp. DH DL GND2 Comp. Logic Logic Control Logic VIN2 FB2 SGND1 Voltage Reference Comp. See Note See Note Operation Device Summary The AAT2511 is a constant frequency peak current mode PWM converter with internal compensation. Each channel has independent input, enable, feed- back, and ground pins with non-synchronized 1MHz clocks. Both converters are designed to operate with an input voltage range of 2.7V to 5.5V. The output voltage ranges from 0.6V to the input voltage. The 0.6V fixed model shown in Figure 1 is also the adjustable version and is externally programmable with a resistive divider as shown in Figure 2. The converter MOSFET power stage is sized for 600mA load capability with up to 97% efficiency. Light load efficiency exceeds 80% at a 500µA load.

Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 9 Soft Start The AAT2511 soft-start control prevents output volt- age overshoot and limits inrush current when either the input power or the enable input is applied. When pulled low, the enable input forces the con- verter into a low-power, non-switching state with a bias current of less than 1µA. Low Dropout Operation For conditions where the input voltage drops to the output voltage level, the converter duty cycle increases to 100%. As 100% duty cycle is approached, the minimum off-time initially forces the high side on-time to exceed the 1MHz clock cycle and reduce the effective switching frequency. Once the input drops below the level where the out- put can be regulated, the high side P-channel MOSFET is turned on continuously for 100% duty cycle. At 100% duty cycle, the output voltage tracks the input voltage minus the I*R drop of the high side P-channel MOSFET R DS(ON). Low Supply The under-voltage lockout (UVLO) feature guaran- tees sufficient V IN bias and proper operation of all internal circuitry prior to activation. Fault Protection For overload conditions, the peak inductor current is limited. Thermal protection disables switching when the internal dissipation or ambient temperature becomes excessive. The junction over-temperature threshold is 140°C with 15°C of hysteresis. Figure 1: AAT2511 Adjustable Output with Enhanced Transient Response. 4.7μH 10μF 1.8V 118k 59.0k 100pF C4 10μH 10μF 2.5V 59.0k 100pF 10μF VIN 187k FB1 EN1 LX1 GND2 LX2 GND1 SGND1 VIN1 VIN2 SGND2 FB2 EN2 AAT2511 U1 0.1μF

Dual 600mA, 1MHz Step-Down DC/DC Converter 10 2511.2006.05.1.4 Applications Information Inductor Selection The step-down converter uses peak current mode control with slope compensation to maintain stabil- ity for duty cycles greater than 50%. The output inductor value must be selected so the inductor current down slope meets the internal slope com- pensation requirements. The internal slope com- pensation for the adjustable and low-voltage fixed versions of the AAT2511 is 0.24A/µsec. This equates to a slope compensation that is 75% of the inductor current down slope for a 1.5V output and 4.7µH inductor. This is the internal slope compensation for the adjustable (0.6V) version or low-voltage fixed ver- sion. When externally programming the 0.6V ver- sion to a 2.5V output, the calculated inductance would be 7.5µH. In this case, a standard 10µH value is selected. Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation characteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive loss- es due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. The 4.7µH CDRH3D16 series inductor selected from Sumida has a 105mΩ DCR and a 900mA DC current rating. At full load, the inductor DC loss is 38mW for a 600mA 1.5V output. Input Capacitor Select a 4.7µF to 10µF X7R or X5R ceramic capac- itor for the input. To estimate the required input capacitor size, determine the acceptable input rip- ple level (V PP) and solve for C. The calculated value varies with input voltage and is a maximum when V IN is double the output voltage. This equation provides an estimate for the input capacitor required for a single channel. VO VIN CIN = VO VIN VPP IO 0.75 ⋅ VO L = = ≈ 3 ⋅ VO = 3 ⋅ 2.5V = 7.5μH m 0.75V 0.24A /μsec μsec A μsec A 0.75 ⋅ VO m = = = 0.24 L 0.75 ⋅ 1.5V 4.7μH A μsec Table 1: Inductor Values. Configuration Output Voltage Inductor Slope Compensation External Resistive Divider 2.5V 10µH 0.24A/µsec

The equation below solves for input capacitor size for both channels. It makes the worst-case assumptions that both converters are operating at 50% duty cycle and are synchronized. Because the AAT2511 channels will generally oper- ate at different duty cycles and are not synchro- nized, the actual ripple will vary and be less than the ripple (V PP) used to solve for the input capaci- tor in the equation above. Always examine the ceramic capacitor DC voltage coefficient characteristics when selecting the prop- er value. For example, the capacitance of a 10µF 6.3V X5R ceramic capacitor with 5V DC applied is actually about 6µF. The maximum input capacitor RMS current is: The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current of both converters combined. This equation also makes the worst-case assump- tion that both converters are operating at 50% duty cycle and are synchronized. Since the converters are not synchronized and are not both operating at 50% duty cycle, the actual RMS current will always be less than this. Losses associated with the input ceramic capacitor are typically minimal. The term appears in both the input voltage ripple and input capacitor RMS current equations. It is a maximum when V O is twice VIN. This is why the input voltage ripple and the input capacitor RMS current ripple are a maximum at 50% duty cycle. The input capacitor provides a low impedance loop for the edges of pulsed current drawn by the AAT2511. Low ESR/ESL X7R and X5R ceramic capacitors are ideal for this function. To minimize the stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C3 and C8) can be seen in the evaluation board layout in Figure 4. Since decoupling must be as close to the input pins as possible, it is necessary to use two decoupling capacitors. C3 provides the bulk capacitance required for both converters, while C8 is a high frequency bypass capacitor for the second channel (see C3 and C8 placement in Figure 4). A laboratory test set-up typically consists of two long wires running from the bench power supply to the evaluation board input voltage pins. The induc- tance of these wires, along with the low ESR ceramic input capacitor, can create a high Q net- work that may affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage during load transients. Errors in the loop phase and gain measurements can also result. Since the inductance of a short printed circuit board trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. In applications where the input power source lead inductance cannot be reduced to a level that does not affect converter performance, a high ESR tan- talum or aluminum electrolytic capacitor should be placed in parallel with the low ESR, ESL bypass ceramic capacitor. This dampens the high Q net- work and stabilizes the system. Output Capacitor The output capacitor limits the output ripple and provides holdup during large load transitions. A 4.7µF to 10µF X5R or X7R ceramic capacitor typi- cally provides sufficient bulk capacitance to stabi- VO VIN VO VIN IO1(MAX) + IO2(MAX) RMS(MAX)I 2= VO1 VIN VO1 VIN VO2 VIN VO2 VIN CIN = 1 VPP IO1 + IO2 AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 11

Dual 600mA, 1MHz Step-Down DC/DC Converter 12 2511.2006.05.1.4 lize the output during large load transitions and has the ESR and ESL characteristics necessary for low output ripple. The output voltage droop due to a load transient is dominated by the capacitance of the ceramic out- put capacitor. During a step increase in load cur- rent the ceramic output capacitor alone supplies the load current until the loop responds. As the loop responds, the inductor current increases to match the load current demand. This typically takes two to three switching cycles and can be estimated by: Once the average inductor current increases to the DC load level, the output voltage recovers. The above equation establishes a limit on the minimum value for the output capacitor with respect to load transients. The internal voltage loop compensation also limits the minimum output capacitor value to 4.7µF. This is due to its effect on the loop crossover frequency (bandwidth), phase margin, and gain margin. Increased output capacitance will reduce the crossover frequency with greater phase margin. The maximum output capacitor RMS ripple current is given by: Dissipation due to the RMS current in the ceramic output capacitor ESR is typically minimal, resulting in less than a few degrees rise in hot spot temperature. Adjustable Output Resistor Selection For applications requiring an adjustable output volt- age, the 0.6V version can be programmed exter- nally. Resistors R1 through R4 of Figure 1 program the output to regulate at a voltage higher than 0.6V. To limit the bias current required for the external feedback resistor string, the minimum suggested value for R2 and R4 is 59k Ω. Although a larger value will reduce the quiescent current, it will also increase the impedance of the feedback node, making it more sensitive to external noise and interference. Table 2 summarizes the resistor val- ues for various output voltages with R2 and R4 set to either 59k Ω for good noise immunity or 221k Ω for reduced no load input current. The adjustable version of the AAT2511 in combina- tion with an external feedforward capacitor (C4 and C5 of Figure 1) delivers enhanced transient response for extreme pulsed load applications. The addition of the feedforward capacitor typically requires a larger output capacitor (C1 and C2) for stability. Table 2: Adjustable Resistor Values For Use With 0.6V Version. R2, R4 = 59kΩΩ R2, R4 = 221kΩΩ VOUT (V) R1, R3 (k ΩΩ) R1, R3 (k ΩΩ) 0.8 19.6 75 0.9 29.4 113 1.0 39.2 150 1.1 49.9 187 1.2 59.0 221 1.3 68.1 261 1.4 78.7 301 1.5 88.7 332 1.8 118 442 1.85 124 464 2.0 137 523 2.5 187 715 VOUT VREF 1.5V 0.6V VOUT · (VIN(MAX) - VOUT) RMS(MAX)I L · F · VIN(MAX) COUT = 3 · ΔILOAD VDROOP · FS

There are three types of losses associated with the AAT2511 converter: switching losses, conduction losses, and quiescent current losses. Conduction losses are associated with the R DS(ON) characteristics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assum- ing continuous conduction mode (CCM), a simplified form of the dual converter losses is given by: I Q is the AAT2511 quiescent current for one chan- nel and tsw is used to estimate the full load switch- ing losses. For the condition where channel one is in dropout at 100% duty cycle, the total device dissipation reduces to: Since R DS(ON), quiescent current, and switching losses all vary with input voltage, the total losses should be investigated over the complete input voltage range. Given the total losses, the maximum junction tem- perature can be derived from the θ JA for the TDFN33-12 package which is 50°C/W. PCB Layout The following guidelines should be used to insure a proper layout. 1. Due to the pin placement of V IN for both con- verters, proper decoupling is not possible with just one input capacitor. The large input capaci- tor C3 should connect as closely as possible to V P and GND, as shown in Figure 3. The addi- tional input bypass capacitor C8 is necessary for proper high frequency decoupling of the second converter. 2. The output capacitor and inductor should be connected as closely as possible. The connec- tion of the inductor to the LX pin should also be as short as possible. 3. The feedback trace should be separate from any power trace and connect as closely as possible to the load point. Sensing along a high-current load trace will degrade DC load regulation. If external feedback resistors are used, they should be placed as closely as possible to the FB pin. This prevents noise from being coupled into the high impedance feedback node. 4. The resistance of the trace from the load return to GND should be kept to a minimum. This will help to minimize any error in DC regulation due to differences in the potential of the internal sig- nal ground and the power ground. 5. For good thermal coupling, PCB vias are required from the pad for the TDFN paddle to the ground plane. The via diameter should be 0.3mm to 0.33mm and positioned on a 1.2 mm grid. TJ(MAX) = PTOTAL · ΘJA + TAMB PTOTAL = IO1 2 · RDSON(HS) + (tsw · F · IO2 + 2 · IQ) · VIN IO2 2 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2]) VIN PTOTAL IO1 2 · (RDSON(HS) · VO1 + RDSON(LS) · [VIN -VO1]) VIN + (tsw · F · [IO1 + IO2] + 2 · IQ) · VIN IO2 2 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2]) VIN AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 13

VO1 = 2.5V @ 600mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA VO2 = 1.8V @ 600mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA VIN = 2.7V to 4.2V (3.6V nominal) FS = 1.0 MHz TAMB = 85°C 2.5V VO1 Output Inductor (see Table 1) For Sumida inductor CDRH3D16, 10µH, DCR = 210m Ω. 1.8V VO2 Output Inductor (see Table 1) For Sumida inductor CDRH3D16, 4.7µH, DCR = 105m Ω. IPK2 = IO2 + ΔI2 = 0.6A + 0.11A = 0.71A2 PL2 = IO2 2 ⋅ DCR = (0.6A)2 ⋅ 105mΩ = 38mW L2 = 3 ⋅ VO2 = 3 ⋅ 1.8V = 5.4μHμsec A μsec A VO VO1 2.5V 2.5V ΔI1 = ⋅ 1 - = ⋅ 1 - = 100mAL1 ⋅ F VIN 10μH ⋅ 1.0MHz 4.2V IPK1 = IO1 + ΔI1 = 0.6A + 0.05A = 0.65A2 PL1 = IO1 2 ⋅ DCR = (0.4A)2 ⋅ 210mΩ = 76mW L1 = 3 ⋅ VO1 = 3 ⋅ 2.5V = 7.5μHμsec A μsec A AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 14 2511.2006.05.1.4

2.5V Output Capacitor 1.8V Output Capacitor Input Capacitor Input Ripple VPP = 25mV. IO1 + IO2 RMS(MAX)I P = esr · IRMS 2= = 6Arms CIN = = = 15.78 μF1 VPP IO1 + IO2 25mV 1.2A 4.7μH · 1.0MHz · 4.2V 23 RMS(MAX)I L · F · VIN(MAX) 3 · ΔILOAD VDROOP · FS 3 · 0.6A 0.2V · 1MHz COUT = = = 9 μF

  • = 63mArms (VOUT) · (VIN(MAX) - VOUT) = Pesr = esr · IRMS 2 = 5mΩ · (63mA)2 = 20μW 10μH · 1MHz · 4.2V 23 RMS(MAX)I L · F · VIN(MAX) 3 · ΔILOAD VDROOP · FS 3 · 0.6A 0.2V · 1MHz COUT = = = 9 μF
  • = 29mArms (VOUT) · (VIN(MAX) - VOUT) = Pesr = esr · IRMS 2 = 5mΩ · (29mA)2 = 4.2μW AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 15

The maximum dissipation occurs at dropout where V IN = 2.7V. All values assume an ambient temperature of 85°C and a junction temperature of 120°C. Figure 2: AAT2511 Evaluation Board Schematic. see Table 3 4.7μF C21 VO1 GND see Table 3 59.0k FB1 EN1 LX1 GND2 LX2 GND1 SGND1 VIN1 VIN2 SGND2 FB2 EN2 AAT2511U1 10μF see Table 3 VO2 GND LX2 see Table 3 59.0k C51 4.7μF C11 LX1 123 Output 1 Enable 321 Output 2 Enable VIN 0.01μF 0.01μF 0.1μF TJ(MAX) = TAMB + ΘJA · PLOSS = 85°C + (50°C/W) · 527mW = 111°C PTOTAL + (tsw · F · IO2 + 2 · IQ) · VIN IO1 2 · (RDSON(HS) · VO1 + RDSON(LS) · (VIN -VO1)) + IO2 2 · (RDSON(HS) · VO2 + RDSON(LS) · (VIN -VO2)) VIN + 5ns · 1MHz · 0.6A + 60μA) · 2.7V = 527mW 2.7V AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 16 2511.2006.05.1.4 1. For enhanced transient configuration C5, C4 = 100pF and C1, C2 = 10µF.

Table 3: Evaluation Board Component Values. Figure 3: AAT2511 Evaluation Board Top Side. Figure 4: AAT2511 Evaluation Board Bottom Side. Adjustable Version R2, R4 = 59kΩΩ R2, R4 = 221kΩΩ1 (0.6V device) VOUT (V) R1, R3 (k ΩΩ) R1, R3 (k ΩΩ) L1, L2 (µH) 0.8 19.6 75.0 4.7 0.9 29.4 113 4.7 1.0 39.2 150 4.7 1.1 49.9 187 4.7 1.2 59.0 221 4.7 1.3 68.1 261 4.7 1.4 78.7 301 4.7 1.5 88.7 332 4.7 1.8 118 442 4.7 1.85 124 464 4.7 2.0 137 523 4.7 or 6.8 2.5 187 715 10 3.3 267 1000 10 AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 17 1. For reduced quiescent current, R2 and R4 = 221k Ω.

Table 4: Typical Surface Mount Inductors. Table 5: Surface Mount Capacitors. Manufacturer Part Number Value Voltage Temp. Co. Case MuRata GRM219R61A475KE19 4.7µF 10V X5R 0805 MuRata GRM21BR60J106KE19 10uF 6.3V X5R 0805 MuRata GRM21BR60J226ME39 22uF 6.3V X5R 0805 Inductance Max DC DCR Size (mm) Manufacturer Part Number (µH) Current (A) ( ΩΩ) LxWxH Type AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 18 2511.2006.05.1.4

Ordering Information

All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means Voltage Package Channel 1 Channel 2 Marking1 Part Number (Tape and Reel)2 TDFN33-12 0.6V 0.6V QRXYY AAT2511IWP-AA-T1 AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 2511.2006.05.1.4 19 1. XYY = assembly and date code. 2. Sample stock is generally held on part numbers listed in BOLD. Legend Voltage Code Adjustable A(0.6V) 0.9 B 1.2 E 1.5 G 1.8 I 1.9 Y 2.5 N 2.6 O 2.7 P 2.8 Q 2.85 R 2.9 S 3.0 T 3.3 W 4.2 C

Package Information

All dimensions in millimeters. Top View Bottom View Detail "B" Detail "A"Side View 3.00 ± 0.05 Index Area (D/2 x E/2) Detail "A" Detail "B" 1.70 ± 0.05 3.00 ± 0.05 0.05 ± 0.05 0.229 ± 0.051 7.5° ± 7.5° 2.40 ± 0.05 0.16 Pin 1 Indicator (optional) 0.075 ± 0.075

0.1 REF

0.8 + 0.05 -0.20 Option A: C0.30 (4x) max Chamfered corner Option B: R0.30 (4x) max Round corner AAT2511 Dual 600mA, 1MHz Step-Down DC/DC Converter 20 2511.2006.05.1.4 Advanced Analogic Technologies, Inc. 830 E. Arques Avenue, Sunnyvale, CA 94085 Phone (408) 737-4600 Fax (408) 737-4611 © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service without notice. Customers are advised to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. All products are sold sub- ject to the terms and conditions of sale supplied at the time of order acknowledgement, including those pertaining to warranty, patent infringement, and limitation of liability. AnalogicTech warrants performance of its semiconductor products to the specifications applicable at the time of sale in accordance with AnalogicTech’s standard warranty. Testing and other quality con- trol techniques are utilized to the extent AnalogicTech deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and produ ct names appearing in this document are regis- tered trademarks or trademarks of their respective holders.