AAT1145 ANALOGICTECH | Alldatasheet

Document overview

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Technical content

Features

  • Input Voltage Range: 2.5V to 5.5V
  • Output Voltages from 0.6V to V IN
  • 1.2A Output Current
  • High Efficiency: Up to 95%
  • 1.5MHz Constant Switching Frequency
  • Low R DS(ON) Internal Switches: 0.15Ω
  • Allows Use of Ceramic Capacitors
  • Current Mode Operation for Excellent Line and Load Transient Response
  • Short-Circuit and Thermal Fault Protection
  • Soft Start
  • Low Dropout Operation: 100% Duty Cycle
  • Low Shutdown Current: I SHUTDOWN < 1μA
  • TDFN33-10 Package
  • -40°C to +85°C Temperature Range

Applications

  • Cellular Phones
  • Digital Cameras
  • DSP Core Supplies
  • PDAs
  • Portable Instruments
  • Smart Phones Typical Application PGND PGND LX LXEN IN AIN AGND OUTAAT1145-1.8 10μF VIN 2.5V-5.5V 2.2μH 22μF VOUT 1.8V,1.2A AGND6

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Pin Descriptions Pin # Symbol Function 1E N Enable pin. Active high. In shutdown, all functions are disabled drawing <1μA supply current. Do not leave EN fl oating. 2 IN Power supply input pin. Must be closely decoupled to AGND with a 2.2 μF or greater ceramic capacitor. 3 AIN Analog supply input pin. Provides bias for internal circuitry. 4, 6 AGND Analog ground pin

5 FB/OUT

FB pin (AAT1145IDE-0.6): Adjustable version feedback input. Connect FB to the center point of the external resistor divider. The feedback threshold voltage is 0.6V. OUT pin (AAT1145IDE-1.8): Fixed version feedback input. Connect OUT to the output voltage, VOUT. 7, 8 LX Switching node pin. Connect the output inductor to this pin. 9, 10 PGND Power ground pin EP Power ground exposed pad. Must be connected to bare copper ground plane. Pin Configuration TDFN-10 (Top View) AIN FB/OUT1 AGND EN IN LX AGND LX PGND PGND 1. FB pin for the adjustable voltage version (AAT1145IDE-0.6), OUT pin for the fixed voltage version (AAT1145IDE-1.8).

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Absolute Maximum Ratings1 Symbol Description Value Units IN, AIN Input Supply Voltages -0.3 to 6.0 V VFB, VLX FB, LX Voltages -0.3 to VIN + 0.3 V VEN EN Voltage -0.3 to VIN + 0.3 V PGND, AGND Ground Voltages -0.3 to 6.0 V TA Operating Temperature Range -40 to +85 °C TSTORAGE Storage Temperature -65 to 150 °C TLEAD Lead Temperature (Soldering, 10s) 300 °C Thermal Information3 Symbol Description Value Units θJA Thermal Resistance2 45 °C/W PD Maximum Thermal Dissipation at TA = 25°C 2.2 W 1. Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. 2. TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: T J = TA + PD · θJA. 3. Thermal Resistance is specified with approximately 1 square inch of 1 oz. copper.

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Electrical Characteristics1 VIN = 3.6V, TA = -40°C to +85°C unless otherwise noted; typical values are T A = 25°C. Symbol Description Conditions Min Typ Max Units VIN Input Voltage Range2 2.5 5.5 V VOUT Output Voltage Range 0.6 V IN V IQ Input DC Supply Current Active Mode V FB = 0.5V 300 500 μA Shutdown Mode V EN = 0V, VAIN = 5.5V 0.1 1 μA IFB Feedback Input Bias Current V FB = 0.65V 30 nA VFB Regulated Feedback Voltage3 TA = 25°C 0.5880 0.6000 0.6120 VTA = 0°C ≤ TA ≤ 85°C 0.5865 0.6000 0.6135 TA = -40°C ≤ TA ≤ 85°C 0.5850 0.6000 0.6150 ΔVLINEREG/ ΔVIN Line Regulation V IN = 2.5V to 5.5V, IOUT = 10mA 0.10 0.20 %/V ΔVLOADREG/ ΔIOUT Load Regulation I OUT = 10mA to 1200mA 0.50 %/A VFB VOUT Output Voltage Accuracy V IN = 2.5 to 4.2V, IOUT = 10 to 1200mA -3 +3 % FOSC Oscillator Frequency V FB = 0.6V 1.2 1.5 1.8 MHz TS Startup Time From Enable to Output Regulation 1.3 ms TSD Over-Temperature Shutdown Threshold 170 °C THYS Over-Temperature Shutdown Hysteresis 10 °C ILIM Peak Switch Current 2.5 A RDS(ON) P-CH MOSFET V IN = 3.6V 135 200 mΩN-CH MOSFET V IN = 3.6V 95 150 VEN(L) Enable Threshold Low 0.3 V VEN(H) Enable Threshold High 1.5 V IEN Input Low Current V IN = VEN = 5.5V -1.0 1.0 μA 1. The AAT1145 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assu red by design, characterization, and correla- tion with statistical process controls. 2. VIN should be not less than V OUT + VDROPOUT, where VDROPOUT = IOUT · (RDS(ON)PMOS + ESRINDUCTOR), typically VDROPOUT = 0.3V. 3. The regulated feedback voltage is tested in an internal test mode that connects V FB to the output of the error amplifier.

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Typical Characteristics Efficiency vs. Output Current (VOUT = 3.3V; TA = 25°°C; L = 2.2µH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 10000 VIN = 3.6V VIN = 4.2V VIN = 5.0V Efficiency vs. Output Current (VOUT = 1.8V; TA = 25°°C; L = 2.2µH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 10000 VIN = 2.5V VIN = 3.6V VIN = 5.0V VIN = 4.2V Efficiency vs. Output Current (VOUT = 1.5V; TA = 25°°C; L = 2.2µH) Output Current (mA) Efficiency (%) 0.1 1 10 100 1000 10000 100 VIN = 4.2V VIN = 5.0V VIN = 3.6V VIN = 2.5V Efficiency vs. Output Current (VOUT = 1.2V; TA = 25°°C; L = 2.2µH) Output Current (mA) Efficiency (%) 100 0.1 1 10 100 1000 10000 VIN = 2.5V VIN = 3.6V VIN = 4.2V VIN = 5.0V Output Voltage vs. Output Current (VOUT = 3.3V; TA = 25°°C; L = 2.2µH) Output Current (mA) Output Voltage (V) 3.200 3.220 3.240 3.260 3.280 3.300 3.320 3.340 3.360 3.380 3.400 0 200 400 600 800 1000 1200 VIN = 5.0V VIN = 4.2V VIN = 3.6V Output Voltage vs. Output Current (VOUT = 1.8V; TA = 25°°C; L = 2.2µH) Output Current (mA) Output Voltage (V) 1.780 1.785 1.790 1.795 1.800 1.805 1.810 1.815 1.820 0 200 400 600 800 1000 1200 VIN = 5.0V VIN = 4.2V VIN = 2.5V VIN = 3.6V

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Typical Characteristics Output Voltage vs. Output Current (VOUT = 1.5V; TA = 25°°C; L = 2.2µH) Output Current (mA) Output Voltage (V) 0 200 400 600 800 1000 1200 1.45 1.46 1.47 1.48 1.49 1.5 1.51 1.52 1.53 1.54 1.55 VIN = 2.5V VIN = 3.6V VIN = 4.2VVIN = 5.0V Output Voltage vs. Output Current (VOUT = 1.2V; TA = 25°°C; L = 2.2µH) Output Current (mA) Output Voltage (V) 1.160 1.170 1.180 1.190 1.200 1.210 1.220 1.230 0 200 400 600 800 1000 1200 VIN = 5.0V VIN = 3.6V VIN = 4.2VVIN = 2.5V Input Current vs. Input Voltage (VOUT = 3.3V; L = 2.2µH) Input Voltage (V) Input Current (mA) 0.5 1.5 2.5 3.5 4.5 Load Transient Response (VOUT = 1.8V, AC Coupled; VIN = 3.6V; 150mA to 1.2A; L = 2.2µH) Time (40µs/div) VOUT (100mV/div) IOUT (500mA/div) 1.2A 150mA Start-Up Response (VOUT = 1.8V; VIN = 3.6V; No Load; CIN = 10µF; COUT = 22µF; L = 2.2µH) Time (1ms/div) VENABLE (2V/div) VOUT (1V/div) Start-Up Response (VOUT = 1.8V; VIN = 3.6V; ILOAD = 1.2A; CIN = 10µF; COUT = 22µF; L = 2.2µH) Time (1ms/div) VENABLE (2V/div) VOUT (1V/div) IIN (500mA/div)

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Functional Description The AAT1145 is a high output current monolithic switch- mode step-down DC-DC converter. The device operates at a fixed 1.5MHz switching frequency, and uses a slope compensated current mode architecture. This step-down DC-DC converter can supply up to 1200mA output cur- rent at V IN = 3V and has an input voltage range from 2.5V to 5.5V. It minimizes external component size and optimizes efficiency at the heavy load range. The slope compensation allows the device to remain stable over a wider range of inductor values so that smaller values (1μH to 4.7μH) with lower DCR can be used to achieve higher efficiency. Apart from the small bypass input capacitor, only a small L-C filter is required at the output. The fixed output version requires only three external power components (C IN, COUT, and L). The adjustable ver- sion can be programmed with external feedback to any voltage, ranging from 0.6V to near the input voltage. It uses internal MOSFETs to achieve high efficiency and can generate very low output voltages by using an internal reference of 0.6V. At dropout, the converter duty cycle increases to100% and the output voltage tracks the input voltage minus the low R DS(ON) drop of the P-channel high-side MOSFET and the inductor DCR. The internal error amplifier and compensation provides excellent transient response, load and line regulation. Internal soft start eliminates any output voltage overshoot when the enable or the input voltage is applied. Current Mode PWM Control Slope compensated current mode PWM control provides stable switching and cycle-by-cycle current limit for excellent load and line response with protection of the internal main switch (P-channel MOSFET) and synchro- nous rectifier (N-channel MOSFET). During normal operation, the internal P-channel MOSFET is turned on for a specified time to ramp the inductor current at each rising edge of the internal oscillator, and switched off when the peak inductor current is above the error volt- age. The current comparator, I COMP, limits the peak inductor current. When the main switch is off, the syn- chronous rectifier turns on immediately and stays on Functional Block Diagram R1* R2* R1* R2* PWM LOGIC IN VIN 2.5V to 5.5V VOUT VIN COUT LX PGND OVDET OSC SLOPE COMP Softstart REF 0.65V 0.6V 0.6V AIN AGND ICOMP SET RESET SHUTDOWN EN OUT Over-Temperature and Short-Circuit Protection *The resistor divider R1 + R2 is internally set for the fixed output versions, and is externally set for the adjustable output versions. NON-OVERLAP CONTROL IZERO COMP ISENSE AMP

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET until either the inductor current starts to reverse, as indicated by the current reversal comparator, I ZERO, or the beginning of the next clock cycle. Control Loop The AAT1145 is a peak current mode step-down con- verter. The current through the P-channel MOSFET (high side) is sensed for current loop control, as well as short circuit and overload protection. A slope compensation signal is added to the sensed current to maintain stabil- ity for duty cycles greater than 50%. The peak current mode loop appears as a voltage-programmed current source in parallel with the output capacitor. The output of the voltage error amplifier programs the current mode loop for the necessary peak switch current to force a constant output voltage for all load and line conditions. Internal loop compensation terminates the transconduc- tance voltage error amplifier output. For fixed voltage versions, the error amplifier reference voltage is inter- nally set to program the converter output voltage. For the adjustable output, the error amplifier reference is fixed at 0.6V. Soft Start / Enable Soft start limits the current surge seen at the input and eliminates output voltage overshoot. The enable pin is active high. When pulled low, the enable input (EN) forces the AAT1145 into a low-power, non-switching state. The total input current during shutdown is less than 1μA. Current Limit and Over-Temperature Protection For overload conditions, the peak input current is limited to 2.5A. To minimize power dissipation and stresses under current limit and short-circuit conditions, switch- ing is terminated after entering current limit for a series of pulses. The termination lasts for seven consecutive clock cycles after a current limit has been sensed during a series of four consecutive clock cycles. Thermal protection completely disables switching when internal dissipation becomes excessive. The junction over-temperature threshold is 170°C with 10°C of hys- teresis. Once an over-temperature or over-current fault conditions is removed, the output voltage automatically recovers. Dropout Operation When the battery input voltage decreases near the value of the output voltage, the AAT1145 allows the main switch to remain on for more than one switching cycle and increases the duty cycle until it reaches 100%. The duty cycle D of a step-down converter is defined as:

  • 100%D = TON · FOSC · 100% ≈ VOUT VIN Where TON is the main switch on time and F OSC is the oscillator frequency. The output voltage then is the input voltage minus the voltage drop across the main switch and the inductor. At low input supply voltage, the R DS(ON) of the P-channel MOSFET increases, and the efficiency of the converter decreases. Caution must be exercised to ensure the heat dissipated does not exceed the maxi- mum junction temperature of the IC. Maximum Load Current The AAT1145 will operate with an input supply voltage as low as 2.5V, however, the maximum load current decreases at lower input voltages due to a large IR drop on the main switch and synchronous rectifier. The slope compensation signal reduces the peak inductor current as a function of the duty cycle to prevent sub-harmonic oscillations at duty cycles greater than 50%. Conversely the current limit increases as the duty cycle decreases.

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Where ΔIL is inductor ripple current. Large value induc- tors lower ripple current and small value inductors result in high ripple currents. Choose inductor ripple current approximately 30% of the maximum load current 1200mA, or ΔIL = 360mA For output voltages above 2.0V, when light-load effi- ciency is important, the minimum recommended induc- tor is 2.2μH. Manufacturer’s specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the satura- tion characteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive losses due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. For optimum voltage-positioning load tran- sients, choose an inductor with DC series resistance in the 20m Ω to 100m Ω range. For higher efficiency at heavy loads (above 200mA), or minimal load regulation (but some transient overshoot), the resistance should be kept below 100mΩ. The DC current rating of the induc- tor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation (1200mA + 360mA). Table 2 lists some typical surface mount inductors that meet target applications for the AAT1145. For example, the 2.2μH SD3118-2R2-R inductor selected from Coiltronics has a 74mΩ DCR and a 2.00ADC current rating. At full load, the inductor DC loss is 106mW which gives a 5% loss in efficiency for a 1200mA, 1.8V output. Slope Compensation The AAT1145 step-down converter uses peak current mode control with slope compensation for stability when duty cycles are greater than 50%. The slope compensa- tion is set to maintain stability with lower value inductors which provide better overall efficiency. The output induc- tor value must be selected so the inductor current down slope meets the internal slope compensation require- ments. As an example, the value of the slope compensa- tion is set to 1A/ μs which is large enough to guarantee stability when using a 2.2μH inductor for all output volt- age levels from 0.6V to 3.3V. The worst case external current slope (m) using the 2.2μH inductor is when V OUT = 3.3V and is: VOUT Lm = = = 1.5A/µs 3.3 2.2 To keep the power supply stable when the duty cycle is above 50%, the internal slope compensation (mA) should be: 2ma ≥ · m = 0.75A/µs Therefore, to guarantee current loop stability, the slope of the compensation ramp must be greater than one-half of the down slope of the current waveform. So the inter- nal slope compensated value of 1A/ μs will guarantee stability using a 2.2μH inductor value for all output volt- ages from 0.6V to 3.3V. Input Capacitor Selection The input capacitor reduces the surge current drawn from the input and switching noise from the device. The input capacitor impedance at the switching frequency should be less than the input source impedance to pre- vent high frequency switching current passing to the input. The calculated value varies with input voltage and is a maximum when V IN is double the output voltage. VO VIN CIN = VO VIN VPP IO CIN(MIN) = 1 VPP IO A low ESR input capacitor sized for maximum RMS cur- rent must be used. Ceramic capacitors with X5R or X7R dielectrics are highly recommended because of their low ESR and small temperature coefficients. A 22μF ceram- ic capacitor for most applications is sufficient. A large value may be used for improved input voltage filtering. The maximum input capacitor RMS current is: VO VIN VO VIN

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current. RMS(MAX)I· I O To minimize stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C1) can be seen in the evaluation board layout in Figures 3 and 4. A laboratory test set-up typically consists of two long wires running from the bench power supply to the eval- uation board input voltage pins. The inductance of these wires, along with the low-ESR ceramic input capacitor, can create a high Q network that may affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage dur- ing load transients. Errors in the loop phase and gain measurements can also result. Since the inductance of a short PCB trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. In applications where the input power source lead induc- tance cannot be reduced to a level that does not affect the converter performance, a high ESR tantalum or alu- minum electrolytic should be placed in parallel with the low ESR, ESL bypass ceramic. This dampens the high Q network and stabilizes the system. Output Capacitor Selection The function of output capacitance is to store energy to attempt to maintain a constant voltage. The energy is stored in the capacitor’s electric field due to the voltage applied. The value of output capacitance is generally selected to limit output voltage ripple to the level required by the specification. Since the ripple current in the output induc- tor is usually determined by L, V OUT and V IN, the series impedance of the capacitor primarily determines the out- put voltage ripple. The three elements of the capacitor that contribute to its impedance (and output voltage ripple) are equivalent series resistance (ESR), equivalent series inductance (ESL), and capacitance (C). The output voltage droop due to a load transient is dominated by the capacitance of the ceramic output capacitor. During a step increase in load current, the ceramic output capacitor alone supplies the load current until the loop responds. Within two switching cycles, the loop responds and the inductor current increases to match the load current demand. The relationship of the output voltage droop during the two switching cycles to the output capacitance can be estimated by: COUT = 2 · ΔILOAD VDROOP · fS In many practical designs, to get the required ESR, a capacitor with much more capacitance than is needed must be selected. For both continuous or discontinuous inductor current mode operation, the ESR of the C OUT needed to limit the ripple to ∆VO, V peak-to-peak is: ESR ≤ ΔVO ΔIL Ripple current flowing through a capacitor’s ESR causes power dissipation in the capacitor. This power dissipation causes a temperature increase internal to the capacitor. Excessive temperature can seriously shorten the expect- ed life of a capacitor. Capacitors have ripple current rat- ings that are dependent on ambient temperature and should not be exceeded. The output capacitor ripple cur- rent is the inductor current, I L, minus the output current, IO. The RMS value of the ripple current flowing in the output capacitance (continuous inductor current mode operation) is given by: RMSI= ΔIL · = ΔIL · 0.289 ESL can be a problem by causing ringing in the low megahertz region but can be controlled by choosing low ESL capacitors, limiting lead length (PCB and capacitor), and replacing one large device with several smaller ones connected in parallel.

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET In conclusion, in order to meet the requirement of out- put voltage ripple small and regulation loop stability, ceramic capacitors with X5R or X7R dielectrics are rec- ommended due to their low ESR and high ripple current ratings. The output ripple V OUT is determined by: VOUT · (VIN - VOUT) VIN · fOSC · L 8 · fOSC · COUT ΔVOUT ≤ · ESR +⎛ A 22μF ceramic capacitor can satisfy most applications. Thermal Calculations There are three types of losses associated with the AAT1145 step-down converter: switching losses, con- duction losses, and quiescent current losses. Conduction losses are associated with the R DS(ON) characteristics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assuming continuous con- duction mode (CCM), a simplified form of the losses is given by: PTOTAL IO 2 · (RDSON(HS) · VO + RDSON(LS) · [VIN - VO]) VIN + (tsw · F · IO + IQ) · VIN IQ is the step-down converter quiescent current. The term tsw is used to estimate the full load step-down converter switching losses. For the condition where the step-down converter is in dropout at 100% duty cycle, the total device dissipation reduces to: PTOTAL = IO 2 · RDSON(HS) + IQ · VIN Since RDS(ON), quiescent current, and switching losses all vary with input voltage, the total losses should be inves- tigated over the complete input voltage range. Given the total losses, the maximum junction temperature can be derived from the θ JA for the DFN-10 package which is 45°C/W. TJ(MAX) = PTOTAL · ΘJA + TAMB Layout Guidance When laying out the PC board, the following layout guideline should be followed to ensure proper operation of the AAT1145: 1. The exposed pad (EP) must be reliably soldered to the GND plane. A PGND pad below EP is strongly recommended. 2. The power traces, including the GND trace, the LX trace and the IN trace should be kept short, direct and wide to allow large current flow. The L1 connec- tion to the LX pins should be as short as possible. Use several VIA pads when routing between layers. 3. The input capacitor (C1) should connect as closely as possible to IN (Pin 2) and AGND (Pins 4 and 6) to get good power filtering. 4. Keep the switching node, LX (Pins 7 and 8), away from the sensitive FB/OUT node. 5. The feedback trace or OUT pin (Pin 2) should be separate from any power trace and connect as close- ly as possible to the load point. Sensing along a high-current load trace will degrade DC load regula- tion. If external feedback resistors are used, they should be placed as closely as possible to the FB pin (Pin 5) to minimize the length of the high impedance feedback trace. 6. The output capacitor C2 and L1 should be connected as closely as possible. The connection of L1 to the LX pin should be as short as possible and there should not be any signal lines under the inductor. 7. The resistance of the trace from the load return to PGND should be kept to a minimum. This will help to minimize any error in DC regulation due to differ- ences in the potential of the internal signal ground and the power ground. Figures 4 and 5 show an example of a layout with 2 layers.

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Step-Down Converter Design Example Specifications VO = 1.8V @1.2A VIN = 2.7V to 4.2V (3.6V nominal) fS = 1.5MHz Transient droop = 80mV O = 50mV 1.8V Output Inductor VOUT · (VIN(MAX) - VOUT) VIN(MAX) ⋅ ΔIL ⋅ fOSC 1.8 · (4.2 - 1.8) ΔIL = 30% ⋅ IO = 0.3 · 1.2 = 360mA For Sumida 2.2μH inductor (CDRH2D14) with DCR 75mΩ, the ∆IL should be VO L VO VIN 0.312 ΔIL = ⋅ 1 - · T = 312mA IPKL = IO + ΔIL = 1.2 + = 1.356A2 PL = IO 2 ⋅ DCR = 1.22 ⋅ 0.0359 = 51.7mW 1.8V Output Capacitor 2 · ΔILOAD VDROOP · fS 2 · 1.2 0.08 · 1.5 · 106 COUT = = = 20µF; use 22µF ESR ≤ = = 0.16Ω ΔVO ΔIL 0.05 0.312 Select a 22μF, 10mΩ ESR ceramic capacitor to meet the ripple 50mV requirement. VOUT · (VIN - VOUT) VIN · fOSC · L 8 · fOSC · COUT ΔVOUT ≤ · ESR +⎛ 1.8 · (4.2 - 1.8) IRMS = ΔIL ·0.289 = 0.312 · 0.289 = 90.2mArms PCOUT = ESR · IRMS2 = 0.01 · 0.09022 = 81.4 μW

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Input Capacitor Input ripple VPP = 25mV CIN(MIN) = = = 15.4µF; use 22µF 1 VPP IO 0.025 1.2 IO RMSI PCIN = ESR · IRMS 1.2 2= = = 600mArms AAT1145 Losses PTOTAL = IO 2 · RDS(ON)P · D + IO 2 · RDS(ON)N · (1 - D) + (tSW · fS · IO) · VIN 4.2

1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET AAT1145 1.2A Step-Down ConverterSwitchReg TM PRODUCT DATASHEET Advanced Analogic Technologies, Inc.

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Ordering Information

Output Voltage Package Marking 1 Part Number (Tape and Reel)2 Adj. 0.6V to VIN TDFN33-10 QNXYY AAT1145IDE-0.6-T1 Fixed 1.8V TDFN33-10 WUXYY AAT1145IDE-1.8-T1 Package Information3 TDFN33-10 Top View 3.00 ± 0.05 3.00 ± 0.05 Pin 1 dot by marking Pin 1 identification R0.200 Bottom View 1.70 ± 0.05 0.500 BSC0.23 ± 0.05 0.40 ± 0.05 2.40 ± 0.05 Side View 0.05 ± 0.05

0.203 REF

0.75 ± 0.05 All dimensions in millimeters. 1. XYY = assembly and date code. 2. Sample stock is generally held on all part numbers listed in BOLD. 3. The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the lead terminals due to the manufacturing process. A solder fillet at the exposed copper edge cannot be guaranteed and is not required to ensure a proper bottom solder connection.