AAT1106

AAT1106_07 ANALOGICTECH | Alldatasheet

Document overview

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• PDF pages: 18

Technical content

Features

• V IN Range:2.5Vto5.5V
• V OUT : Adjustable0.6VtoV IN
• Upto600mAOutputCurrent
• Upto96%Efficiency
• 1.5MHzSwitchingFrequency
• 100%DutyCycleDropoutOperation
• AdaptiveSlopeCompensatedCurrentMode ControlforExcellentLineandLoadTransient Response
• <1µAShutdownCurrent
• Short-CircuitandThermalFaultProtection
• TSOT23-5Package
• -40°Cto+85°CTemperatureRange

Applications

• CellularPhones,Smartphones
• DigitalStillCameras
• DigitalVideoCameras
• MicroprocessorandDSPCoreSupplies
• MP3andPortableMediaPlayers
• PDAs
• WirelessandDSLModems TypicalApplication AAT1106-1.8 IN EN LX OUT GND VIN 2.5V to 5.5V 4.7µF 10µF 2.2µH VOUT 1.8V

2 1106.2007.07.1.0 PinDescriptions PinConfiguration TSOT23-5 (TopView) Adjustable OutputVersion FixedOutputVersions (AAT1106ICB-0.6) (AAT1106ICB-1.5, AAT1106ICB-1.8) GND FB INLX EN 1 3 4 GND OUT INLX EN 1 3 4 Pin# Symbol Function 1 EN Enablepin.Activehigh.Inshutdown,allfunctionsaredisableddrawing<1µAsupplycurrent. DonotleaveENfloating. 2 GND Groundpin. 3 LX Switchingnode.Connecttheoutputinductortothispin.Connectstothedrainsoftheinternal P-andN-channelMOSFETswitches. 4 IN Supplyinputpin.MustbecloselydecoupledtoGNDwitha2.2µForlargerceramiccapacitor. 5 FB/OUT FB(AAT1106ICB-0.6):Feedbackinputpin.ConnectFBtothecenterpointoftheexternal resistordivider.Thefeedbackthresholdvoltageis0.6V. OUT(AAT1106ICB-1.5,AAT1106ICB-1.8):Outputvoltagepin.

1106.2007.07.1.0 3 Absolute Maximum Ratings Recommended OperatingConditions Symbol Description Value Units θJA ThermalResistance(TSOT23-5) 150 °C/W PD MaximumPowerDissipationatT A =25°C 667 mW Symbol Description Value Units VIN InputSupplyVoltage -0.3to6.0 V VEN ,V FB EN,FBVoltages -0.3toV IN +0.3 V VLX ,V OUT LX,OUTVoltages -0.3toV IN +0.3 V TJ OperatingTemperatureRange -40to+85 °C StorageTemperatureRange -65to+150 °C TLEAD LeadTemperature(soldering,10s) 300 °C 1.AbsoluteMaximumRatingsarethosevaluesbeyondwhichthelifeofadevicemaybeimpaired. 2.T J iscalculatedfromtheambienttemperatureT A andpowerdissipationP D accordingtothefollowingformula:T J =T A +P D xθJA . 3.Thermalresistanceisspecifiedwithapproximately1squareinchof1ozcopper.

4 1106.2007.07.1.0 ElectricalCharacteristics VIN =V EN =3.6V,T A =25°C,unlessotherwisenoted. Symbol Description Conditions Min Typ Max Units Step-DownConverter VIN InputVoltageRange 2.5 5.5 V IQ InputDCSupplyCurrent ActiveMode,V FB =0.5V 270 400 µA ShutdownMode,V FB =0V,V IN =4.2V 0.08 1.0 TA =25°C 0.5880 0.6000 0.6120 VFB RegulatedFeedbackVoltage TA =0°C≤ TA ≤ +85°C 0.5865 0.6000 0.6135 V TA =-40°C≤ TA ≤ +85°C 0.5850 0.6000 0.6150 IFB FBInputBiasCurrent VFB =0.65V -30 30 nA ∆VOUT / OutputVoltageLine VIN =2.5Vto5.5V,IOUT =10mA 0.11 0.40 %/V VOUT /∆VIN Regulation ∆VOUT / OutputVoltageLoad IOUT =10mAto600mA 0.0015 %/mA VOUT /∆IOUT Regulation ILIM MaximumOutputCurrent VIN =3.0V 600 mA FOSC OscillatorFrequency VFB =0.6VorV OUT =100% 1.2 1.5 1.8 MHz TS StartupTime FromEnabletoOutput 100 µs Regulation RDS(ON) P-ChannelMOSFET ILX =300mA 0.30 0.50 ΩN-ChannelMOSFET ILX =300mA 0.20 0.45 PeakInductorCurrent VIN =3V,V FB =0.5VorV OUT =90%; 1.20 ADutyCycle<35% OutputOver-VoltageLockout ∆VOVL =V OVL -V FB 60 mV VEN(L) EnableThresholdLow 0.4 V VEN(H) EnableThresholdHigh 1.4 V IEN InputLowCurrent -1.0 1.0 µA TSD Over-TemperatureShutdown 150 °C Threshold THYS Over-TemperatureShutdown 15 °C Hysteresis 1.100%productiontestat+25°C.Specificationsoverthetemperaturerangeareguaranteedbydesignandcharacterization.

1106.2007.07.1.0 5 TypicalCharacteristics Efficiency vs. Output Current (VOUT = 1.5V; L = 2.2µH; TA = 25°°C) Output Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.6V VIN = 4.2V Efficiency vs. Output Current (VOUT = 1.2V; L = 2.2µH; TA = 25°°C) Output Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 2.7V VIN = 3.6V VIN = 4.2V Efficiency vs. Output Current (VOUT = 1.8V; L = 2.2µH; TA = 25°°C) Output Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 3.6V VIN = 4.2V VIN = 2.7V Efficiency vs. Output Current (VIN = 3.6V; VOUT = 1.8V; TA = 25°°C) Output Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 L = 10µH L = 4.7µH L = 2.2µH L = 1.4µH Efficiency vs. Output Current (VOUT = 2.5V; L = 2.2µH; TA = 25°°C) Output Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 VIN = 3.6V VIN = 4.2V VIN = 2.7V Efficiency vs. Output Current (VIN = 3.6V; VOUT = 2.5V; TA = 25°°C) Output Current (mA) Efficiency (%) 0.1 1 10 100 1000 100 L = 10µH L = 4.7µH L = 2.2µH L = 1.4µH

6 1106.2007.07.1.0 TypicalCharacteristics Feedback Voltage vs. Temperature (VIN = 3.6V) Temperature (°°C) Feedback Voltage (V) 0.597 0.598 0.599 0.600 0.601 0.602 0.603 0.604 -40 -20 0 20 40 60 80 100 RDS(ON) vs. Temperature (VIN = 3.6V) Temperature (°C) RDS(ON) (Ω) -45 -15 15 45 60 90 0.15 0.18 0.21 0.24 0.27 0.30 0.33 0.36 75-30 0 30 P-Channel N-Channel Frequency vs. Input Voltage (VOUT = 1.8V; ILOAD = 150mA; L = 2.2µH) Input Voltage (V) Frequency (MHz) 1.470 1.480 1.490 1.500 1.510 1.520 1.530 1.540 1.550 1.560 RDS(ON) vs. Input Voltage Input Voltage (V) RDS(ON) (ΩΩ) 0.150 0.200 0.250 0.300 0.350 0.400 2 2.5 3 3.5 4 4.5 5 5.5 6 P-Channel MOSFET N-Channel MOSFET Efficiency vs. Input Voltage (VIN = 3.6V; L = 2.2µH; VOUT = 1.8V) Input Voltage (V) Efficiency (%) 2 3 4 56 100 ILOAD = 10mA ILOAD = 100mA ILOAD = 500mA Output Voltage vs. Output Current (VIN = 3.6V; VOUT = 1.8V; L = 2.2µH) Load Current (mA) Output Voltage (V) 0 200 400 600 800 1000 1200 1.64 1.66 1.68 1.7 1.72 1.74 1.76 1.78 1.8 1.82 1.84

1106.2007.07.1.0 7 TypicalCharacteristics Load Transient Response (PWM Mode Only; ILOAD = 100mA to 400mA; L = 2.2µH; CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V) VSW 2V/div VOUT 100mV/div ILOAD 500mA/div 40µs/div Load Transient Response (Light Load Mode to PWM Mode; ILOAD = 28mA to 400mA; L = 2.2µH; CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V) VSW 2V/div VOUT 200mV/div ILOAD 500mA/div PWM Light Load 4µs/div Frequency VS. Temperature Temperature (°C) OSC Frequency (MHz) -50 -25 0 25 50 150 1.10 1.15 1.20 1.25 1.30 1.35 1.40 1.45 1.50 1.55 1.60 100 Input Supply Current vs. Temperature Temperature (°°C) Input Supply Current (µA) -50 -30 -10 10 30 50 70 90 200 220 240 260 280 300 320

(VOUT = 1.8V; CFF = 22pF; RLOAD = 3ΩΩ; CIN = 4.7µF; COUT = 10µF; L = 2.2µH) Time (20µs/div) Output Voltage (mid) (V) InputCurrent (bottom)(A) -0.25 0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 VEN = 3.0V VOUT = 1.8V IIN Startup Waveform (VOUT = 1.8V; CFF = 0pF; RLOAD = 3ΩΩ; CIN = 4.7µF; COUT = 10µF; L = 2.2µH) Time (20µs/div) Output Voltage (top) (V) InputCurrent (bottom)(A) -0.25 0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 VOUT = 1.8V IIN VEN = 3.0V Startup Waveform (VOUT = 1.8V; CFF = 100pF; RLOAD = 3ΩΩ; CIN = 4.7µF; COUT = 10µF; L = 2.2µH) Time (20µs/div) Output Voltage (top) (V) InputCurrent (bottom)(A) -0.25 0.00 0.25 0.50 0.75 1.00 1.25 1.50 1.75 VEN = 3.0V VOUT = 1.8V IIN AAT1106 600mAStep-DownConverter

1106.2007.07.1.0 9 FunctionalBlock Diagram PWM LOGIC NON-OVERLAP CONTROL DRV OSC SLOPE COMP BLANKING EA COMP0.6V SW COUT FB/OUT VOUT 0.6V VIN REFEN VIN VIN 2.7 - 5.5 V IZERO COMP ISENSE COMP QS R Q R RS LATCH SHUTDOWN1 R2* * ForadjustableoutputR1+ R2areexternal 0.65V OVDET GND FunctionalDescription The AAT1106 is a high performance 600mA, 1.5MHz fixed frequency monolithic switch-mode step-down converter which uses a current mode architecture with an adaptive slope compensation scheme.Itminimizesexternalcomponentsizeand optimizes efficiencyoverthecompleteloadrange. Theadaptiveslopecompensationallowsthedevice toremainstableoverawiderrangeofinductorval- ues so that smaller values (1µH to 4.7µH) with associatedlowerDCRcanbeusedtoachievehigh - er efficiency. Apart from the small bypass input capacitor,only a small L-C filter is requiredat the output.Thefixedoutputversionrequiresonlythree externalpowercomponents(C IN ,C OUT ,andL).The adjustableversioncanbe programmedwithexter- nal feedbackto any voltage,rangingfrom 0.6V to the input voltage. It uses internal MOSFETs to achievehighefficiencyandcan generateverylow output voltage by using an internal reference of 0.6V.Atdropout,theconverterdutycycleincreases to100%andtheoutputvoltagetrackstheinputvolt- age minus the low R DS(ON) drop of the P-channel high-sideMOSFET.Theinputvoltagerangeis2.5V to 5.5V. The converter efficiency has been opti- mizedforallloadconditions,rangingfromnoload to 600mAat V IN = 3V.The internal error amplifier and compensation provides excellent transient response,load,andlineregulation. Current ModePWM Control Slope compensated current mode PWM control provides stable switching and cycle-by-cycle cur- rent limit for excellent load and line response and protection of the internal main switch (P-channel MOSFET) and synchronous rectifier (N-channel MOSFET).Duringnormaloperation,theinternalP- channelMOSFETisturnedonforaspecifiedtime toramptheinductorcurrentateachrisingedgeof theinternaloscillator,andisswitchedoffwhenthe feedbackvoltageisabovethe0.6Vreferencevolt- age.Thecurrentcomparator,ICOMP ,limitsthepeak inductor current. When the main switch is off, the synchronous rectifier turns on immediately and stays on until either the inductor current starts to reverse,as indicatedby the currentreversal com - parator, IZERO , or the beginning of the next clock cycle.

10 1106.2007.07.1.0 ControlLoop The AAT1106 is a peak current mode step-down converter. The current through the P-channel MOSFET (high side) is sensed for current loop control, as well as short circuit and overload pro - tection.An adaptive slope compensation signal is addedtothesensedcurrenttomaintainstabilityfor duty cycles greater than 50%. The peak current modeloopappearsasavoltage-programmedcur- rent source in parallel with the output capacitor. Theoutputofthevoltageerroramplifierprograms the current mode loop for the necessary peak switchcurrenttoforceaconstantoutputvoltagefor allloadandlineconditions.Internalloopcompen - sation terminates the transconductance voltage error amplifier output. For fixed voltage versions, theerroramplifierreferencevoltageisinternallyset to program the converter output voltage. For the adjustable output, the error amplifier reference is fixedat0.6V. Enable Theenablepinisactivehigh.Whenpulledlow,the enableinputforcestheAAT1106intoalow-power, non-switching state.The total input current during shutdownislessthan1 µA. CurrentLimitandOver-Temperature Protection. For overload conditions, the peak input current is limited.Tominimizepowerdissipationandstresses under current limit and short-circuit conditions, switching is terminated after entering current limit for a series of pulses. Switching is terminated for sevenconsecutiveclockcyclesafteracurrentlimit has been sensed for a series of four consecutive clock cycles. Thermal protection completely dis - ablesswitchingwheninternaldissipationbecomes excessive.Thejunctionover-temperaturethreshold is 150°C with 15°C of hysteresis. Once an over- temperature or over-current fault conditions is removed,theoutputvoltageautomaticallyrecovers. DropoutOperation Whentheinputvoltagedecreasestowardthevalue oftheoutputvoltage,theAAT1106allowsthemain switch to remain on for more than one switching cycleandincreasesthedutycycleuntilitreaches 100%. The duty cycle D of a step-down converter is definedas: WhereT ON isthemainswitchontimeandF OSC is theoscillatorfrequency(1.5MHz). Theoutputvoltagethenis theinputvoltage minus the voltage drop across the main switch and the inductor.Atlowinputsupplyvoltage,theR DS(ON) of theP-channelMOSFETincreasesandtheefficien - cy of the converter decreases. Caution must be exercised to ensure the heat dissipateddoes not exceedthemaximumjunctiontemperatureoftheIC. Maximum LoadCurrent TheAAT1106willoperatewithaninputsupplyvolt- age as low as 2.5V; however,the maximum load currentdecreasesatlowerinputduetothelargeIR droponthemainswitchandsynchronousrectifier. The slope compensation signal reduces the peak inductor current as a function of the duty cycle to prevent sub-harmonic oscillations at duty cycles greater than 50%. Conversely, the current limit increasesasthedutycycledecreases. D = TON · FOSC · 100% ≈ VOUT VIN

• 100%

1106.2007.07.1.0 11 Applications Information Figure 1 shows the basic application circuit with AAT1106fixedoutputversions. Figure1: BasicApplicationCircuit withFixed OutputVersions. Figure2: BasicApplicationCircuitwith AdjustableOutputVersion. Settingthe Output Voltage Forapplicationsrequiringanadjustableoutputvolt- age, the AAT1106-0.6 adjustable version can be externally programmed. Resistors R1 and R2 of Figure2programtheoutputtoregulateatavoltage higherthan0.6V.Tolimitthebiascurrentrequired fortheexternalfeedbackresistorstringwhilemain - taining good noise immunity, the minimum sug - gested value for R1 is 59k Ω. Although a larger value will further reduce quiescent current, it will alsoincreasetheimpedanceofthefeedbacknode, making it more sensitive to external noise and interference. Table1 summarizes the resistor val- uesforvariousoutputvoltageswithR1settoeither 59k Ω for good noise immunity or 316k Ω for reducednoloadinputcurrent. The adjustableversion of theAAT1106,combined with an external feed forward capacitor (C2 in Figure2),deliversenhancedtransientresponsefor extreme pulsed load applications. The addition of thefeedforwardcapacitortypicallyrequiresalarg - er output capacitor C3 for stability. The external resistorsetstheoutputvoltageaccordingtothefol- lowingequation: Table1: Resistor Selectionfor Output Voltage Setting;Standard 1% Resistor Values SubstitutedClosest tothe Calculated Values. Inductor Selection For most designs, the AAT1106 operates with inductor values of 1µH to 4.7µH. Low inductance values are physically smaller, but require faster switching, which results in some efficiency loss. Theinductorvaluecanbederivedfromthefollow - ingequation: Where ∆IL is inductor ripple current. Large value inductors lower ripple current and small value inductors result in high ripple currents. Choose inductor ripple current approximately 35% of the maximumloadcurrent600mA,or∆IL =210mA. R1 = 59k R1 = 316k ΩΩ VOUT (V) R2 (k ΩΩ) R2 (k ΩΩ) 0.8 19.6 105 0.9 29.4 158 1.0 39.2 210 1.1 49.9 261 1.2 59.0 316 1.3 68.1 365 1.4 78.7 422 1.5 88.7 475 1.8 118 634 1.85 124 655 2.0 137 732 2.5 187 1000 3.3 267 1430 = 1 21 +6.0 R RV ·V or OUT 

• R1- 1VOUT 0.6VR2 = AAT1106-0.6 IN EN LX FB GND VIN 2.5V to 5.5V 4.7µF 10µF 316K 634K 22pF 2.2µH VOUT 1.8V AAT1106-1.8 IN EN LX OUT GND VIN 2.5V to 5.5V 4.7µF 10µF 2.2µH VOUT 1.8V L = VOUT · (VIN - VOUT) VIN · ∆IL · fOSC

12 1106.2007.07.1.0 Table 2: Typical Surface Mount Inductors. Part L (µH) Max DCR (m ΩΩ) Rated DC Current (A) Size WxLxH (mm) 1.4 56.2 2.52 4.7 108.7 1.15 1.0 4.5 1,72 4.7 162 0.84 1.5 120 1.29 4.7 240 0.79 For output voltages above 2.0V, when l ight-load efficiency is important, the minimum recommended inductor size is 2.2µH. For o ptimum voltage-posi- tioning load transients, choose an inductor with DC series resistance in the 50m Ω to 150m Ω range. For hig her effic iency at heav y loads (abov e 200mA), or minimal load r egulation (with some transient overshoot), the resistance should be kept below 100m Ω. The DC current rating of the induc - tor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation (600mA + 105mA). Table 2 li sts some typical surface mount inducto rs that meet target applications for the AAT1106. Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation characteristics. The inductor should not show any appr eciable saturati on under normal load conditions. Some inducto rs may meet the peak and average cur rent ratings yet result in excessive losses due to a high DCR. Always con - sider the losses associated with the DCR and its effect on the total converter efficiency when select- ing an inductor. For example, the 2.2 µH CR43 series indu ctor selec ted from Sumida has a 71.2m Ω DCR and a 1.75ADC current rating. At full load, the inductor DC loss is 25mW which gives a 2.8% loss in efficiency for a 600mA, 1.5V output. Slope Compensation The AAT1106 step-down converter uses peak cur- rent mode control with a unique adapti ve slope compensation scheme to maint ain stability with lower value inductors for duty cycles greater than 50%. Using lower value inductors provides better overall efficiency and also makes it easier to stan - dardize on one inductor for different required out- put voltage levels. In order to do this and keep the step-down converter stable when the duty cycle is greater than 5 0%, the AAT11 06 separates the slope compensation into 2 phase s. The required slope compensation is auto matically detected by an internal circuit using the feedback voltage V FB before the error amp comparison to V REF . When below 50% duty cycle, the slope compensa - tion is 0.284A/µs; but when above 50% duty cycle, the slope compensation is set to 1.136A/µs. The output inductor value must be sel ected so the inductor current down slope me ets the internal slope compensation requirements. Error AmpVREF VFB

1106.2007.07.1.0 13 Below 50% duty cycl e, the slo pe compensation requirement is: Therefore: Above 50% duty cycle, Therefore: With these adaptive settings, a 2.2µH inductor can be used for all output voltages from 0.6V to 5V. Input Capacitor Selection The input capacitor reduces the surge current drawn from the input and switching noise from the device. The input capacitor impedance at the switching fre - quency shall be less than the input source imped - ance to prevent high frequency switching current passing to the input. A low ESR input capacitor sized for maximum RMS current must be used. Ceramic capacitors with X5R or X 7R dielectrics are highly recommended because of their low ESR and small temperature coefficients. A 4.7µF ceramic capacitor is sufficient for most applications. To estimate the required input capacitor size, deter- mine the acceptable inp ut ripple level (V PP ) and solve for C. The calculated value varies with input voltage and is a maximum when V IN is double the output voltage. Always examine the ceramic capacitor DC voltage coefficient characteristics when selecting the proper value. For example, the capacitance of a 10 µF, 6.3V, X5R ceramic capacitor with 5.0V DC applied is actu - ally about 6 µF. The maximum input capacitor RMS current is: The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current for V IN = 2 x V O The term appears in both the input volt- age ripple and input capacitor RMS current equa - tions and is at maximum when V O is twice V IN . This is why the input voltage ripple and the input capaci- tor RMS current ripple are a maximum at 50% duty cycle. The input capacitor provides a low impedance loop for the edges of pulsed current drawn by the AAT1106. Low ESR/ESL X7R and X5R ceram ic capacitors are ideal for this function. To minimize stray inductance, the capacitor should be placed as VO VIN VO VIN IO RMS(MAX)I 2= VO VIN VO VIN VO VIN VO VIN CIN(MIN) = 1 VPP IO VO VIN VO VIN VO VIN CIN = VO VIN VPP IO 2.5 mL = = 2.2µH 2 · Lm = = 1.136A/µs 0.625 mL = = 2.2µH 1.25 2 · Lm = = 0.284A/µs

14 1106.2007.07.1.0 closely as possible to the IC. This keeps the high fre - quency content of the input current localized, mini- mizing EMI and inp ut voltage ripple. The proper placement of the input capacitor (C1) can be seen in the evaluation board layout in Figure 3. A laboratory test set-up typically consists of two long wires run - ning from the bench power supply to the evaluation board input voltage pins. The inductance of these wires, along with the low-ESR ceramic input capac - itor, can create a high Q network that may affect con - verter performance. This problem often becomes apparent in the form of excessive ringing in the out- put voltage during load transients. Errors in the loop phase and gain mea surements can also result. Since the inductance of a short PCB trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applica - tions do not exhibit this problem. In ap plications where the input power source lead inductance can - not be reduced to a level that does not affect the converter performance, a high ESR tantalum or alu - minum electrolytic should be placed in parallel with the low ESR, ESL bypass ceramic. This dampens the high Q network and stabilizes the system. Output Capacitor Selection The output capacitor is required to keep the output voltage ripple small and to ensur e regulation loop stability. The outp ut ca pacitor mus t hav e low impedance at the switching frequency. Ceramic capacitors with X5R or X7R dielectrics are recom - mended due to their low ESR and high ripple cur- rent. The output ripple V OUT is determined by: The output capacitor limits the output ripple and pro - vides holdup during large load transitions. A 4.7 µF to 10 µF X5R or X7R ceramic capacitor typically pro - vides sufficient bulk capacitance to stabilize the out- put during large load transitions and has the ESR and ESL characteristics necessary for low output ripple. The output voltage droop due to a load tran - sient is dominated by the capacitance of the ceram - ic output capacitor. During a step increase in load current, the ceramic output capacitor alone supplies the load current until the loop responds. Within two or three switching cycles, the loop responds and the inductor current increases to match the load current demand. The relationship of the output voltage droop during the three switching cycles to the output capacitance can be estimated by: Once the average inductor current increases to the DC load level, the output voltage recovers. The above equation establishes a limit on the minimum value for the output capacitor with respect to load transients. The internal voltage loop compensation also limits the mi nimum output capacitor value to 4.7 µF. This is due to its effect on the loop crossover frequency (bandwidth), phase margin, a nd gain margin. Increased output capacitance will reduce the crossover frequency with greater phase margin. The maximum output capacitor RMS ripple current is given by: Dissipation due to the RMS current in the ceramic output capacitor ESR is typically minimal, resulting in less than a few degrees rise in hot-spot temperature. Thermal Calculations There are three types of losses associated with the AAT1106 s tep-down converter: switching l osses, conduction losses, and quiescent curr ent losses. Conduction losses are associated with the R DS(ON) characteristics of the power output swit ching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assuming continuous conduction mode(CCM), a simplified form of the losses is given by: PTOTAL IO 2 · (RDSON(HS) · VO + RDSON(LS) · [VIN - VO]) VIN + (tsw · F · IO + IQ) · VIN VOUT · (VIN(MAX) - VOUT) RMS(MAX)I L · F · VIN(MAX) COUT = 3 · ∆ILOAD VDROOP · FS VOUT · (VIN - VOUT) VIN · fOSC · L 8 · fOSC · C3

1106.2007.07.1.0 15 IQ is the step-down converter quiescent current. The term tsw is used to estimate the full load step- down converter switching losses. For the condition where the step-down converter is in dropout at 100% duty cycle, the total device dis - sipation reduces to: Since R DS(ON), quiescent current, and switch ing losses all vary with input voltage, the total losses should be investigated over the complete input volt- age range. Given the total losses, the maximum junction temperature can be derived from the θJA for the TSOT23-5 package which is 150°C/W. Layout Guidance When laying out the PC board, the following steps should be taken to ensure proper operation of the AAT1106. These items are also illustrated graphi- cally in Figure 3. 1. The power traces (GND, LX, IN) should be kept short, direct, and wide to allow large current flow. Place sufficient multiply-layer pads when needed to change the trace layer. 2. The input capacitor (C1) sho uld connect as closely as possible to IN (Pin 4) and GND (Pin 2). 3. The output capacitor C3 and L1 should be con - nected as closely as possible. The connection of L1 to the LX pin should be as short as pos - sible and there should not be any signal lines under the inductor. 4. The feedback FB trace or OUT pin (Pin 5) should be separate from any power trace and connect as closely as possible to the load point. Sensing along a high-current load trace will degrade DC load regulation. If external feedback resistors are used, they should be placed as closely as possi- ble to the FB pin (Pin 5) to minimize the length of the high impedance feedback trace. 5. The resistance of the trace from the load return to the GND (Pin 2) should be kept to a minimum. This will help to minimize any error in DC regu - lation due to differences in the potential of the internal signal ground and the power ground. TJ(MAX) = PTOTAL · ΘJA + TAMB PTOTAL = IO 2 · RDSON(HS) + IQ · VIN

16 1106.2007.07.1.0 a: Top Layer b: Internal GND Plane c: Bottom Layer d: Middle Layer Figure 3: AAT1106 Four-Layer Layout Example with the Internal GND Plane. AAT1106-0.6 IN EN LX FB GND VIN 2.5V to 5.5V 4.7µF 10µF 316K 634K 22pF 2.2µH VOUT 1.8V

1106.2007.07.1.0 17

Ordering Information

All dimensions in millimeters. Output Voltage Package Marking 1 Part Number (Tape & Reel) 2 Adj. 0.6 to V IN TSOT23-5 VVXYY AAT1106ICB-0.6-T1 Fixed 1.5V TSOT23-5 VXXYY AAT1106ICB-1.5-T1 Fixed 1.8V TSOT23-5 VYXYY AAT1106ICB-1.8-T1 0.40 ±± 0.10 0.95 BSC 1.60 ±± 0.10 0.45 ±± 0.15 0.127 ±± 0.55 Detail "A" 2.80 ±± 0.25

1.90 BSC

0° ++10° -0° Top View End View Detail "A"Side View 1.00 ±± 0.10 0.000 ++ 0.130 - 0.000 2.95 ±± 0.15 All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” m eans semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more information, please visit our website at http://www.analogictech.com/pbfree. 1. XYY = assembly and date code. 2. Sample stock is generally held on part numbers listed in BOLD . 3. Package outline exclusive of mold flash and metal burr.

18 1106.2007.07.1.0 Advanced Analogic Technologies, Inc. 830 E. Arques Avenue, Sunnyvale, CA 94085 Phone (408) 737-4600 Fax (408) 737-4611 © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service with - out notice. Except as provided in AnalogicTech’s terms and conditions of sale, AnalogicTech assumes no liability whatsoever, and AnalogicTech disclaims any express or implied war- ranty relating to the sale and/or use of AnalogicTech products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. In order to minimize risks associated with the customer’s applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. Testing and other quality control techniques are utilized to the extent AnalogicTech deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holders.