3506IFE

LT3506 LINER | Alldatasheet

Document overview

• Manufacturer or author: Provided By ALLDATASHEET.COM(FREE DATASHEET DOWNLOAD SITE)
• PDF pages: 24

Technical content

IOUT (A) EFFICIENCY (%) 100 0.5 1.0

3506 TA01b

1.5 2.0 VIN = 5V VOUT = 1.8V VOUT = 3.3V Dual Monolithic 1.6A Step-Down Switching Regulator The LT®3506 is a dual current mode PWM step-down DC/DC converter with internal 2A power switches. Both convert- ers are synchronized to a single oscillator and run with opposite phases, reducing input ripple current. The output voltages are set with external resistor dividers, and each regulator has independent shutdown and soft-start circuits. Each regulator generates a power-good signal when its output is in regulation, easing power supply sequencing and interfacing with microcontrollers and DSPs. The LT3506 switching frequency is 575kHz and the LT3506A is 1.1MHz. These high switching frequencies allow the use of tiny inductors and capacitors, resulting in a very small dual 1.6A output solution. Constant frequency and ceramic capacitors combine to produce low, predictable output ripple voltage. With its wide input range of 3.6V to 25V, the LT3506 regulates a wide variety of power sources, from 4-cell batteries and 5V logic rails to unregulated wall transformers, lead acid batteries and distributed-power supplies. Current mode PWM architecture provides fast transient response with simple compensation components and cycle-by-cycle current limiting. Frequency foldback and thermal shutdown provide additional protection. ■ Disk Drives ■ DSP Power Supplies ■ Wall Transformer Regulation ■ Distributed Power Regulation ■ DSL Modems ■ Cable Modems ■ Wide Input Voltage Range, 3.6V to 25V ■ Two 1.6A Output Switching Regulators with Internal Power Switches ■ Constant Switching Frequency LT3506: 575kHz LT3506A: 1.1MHz ■ Anti-Phase Switching Reduces Ripple ■ Accurate 0.8V Reference, ±1% ■ Independent Shutdown/Soft-Start Pins ■ Independent Power Good Indicators Ease Supply Sequencing ■ Uses Small Inductors and Ceramic Capacitors ■ Small 16-Lead Thermally Enhanced 5mm × 4mm DFN and TSSOP Surface Mount Packages APPLICATIO SU FEATURES DESCRIPTIO U TYPICAL APPLICATIO U Efficiency VIN 4.5V TO 25V PGOOD1 PGOOD2 3506 F011.5nF 22µF47µF 22µF 0.22µF 0.22µF 10.7k 18.7k 100k 100k 33.2k 6.4µH4.7µH VOUT2 3.3V 1.6A VOUT1 1.8V 1.6A 15k D1 D215k 10k 1.5nF 1000pF 2200pF LT3506 VIN2VIN1 GND 1/2 BAT-54A 1/2 BAT-54A D1, D2: ON SEMI MBR5230LT3 BOOST2 SW2 FB2 VC2 RUN/SS2 BOOST1 SW1 FB1 VC1 RUN/SS1 PGOOD1 PGOOD2 , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.

Absolute M A xi M u M R A tings (Note 1) FB1 VC1 PG1 RUN/SS1 RUN/SS2 PG2 VC2 FB2 BOOST1 SW1 VIN1 VIN1 VIN2 VIN2 SW2 BOOST2 TOP VIEW DHD PACKAGE 16-LEAD PLASTIC DFN TJMAX = 125°C, θJA = 43°C/W, θJC = 4.3°C/W EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB FE PACKAGE 16-LEAD PLASTIC TSSOP NARROW TOP VIEW BOOST1 SW1 VIN1 VIN1 VIN2 VIN2 SW2 BOOST2 FB1 VC1 PG1 RUN/SS1 RUN/SS2 PG2 VC2 FB2 TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB Operating Temperature Range (Note 2) PI CO FIGURATIO U UU LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3506EDHD#PBF LT3506EDHD#TRPBF 3506 16-Lead (5mm x 4mm) Plastic DFN –40°C to 85°C LT3506AEDHD#PBF LT3506AEDHD#TRPBF 3506A 16-Lead (5mm x 4mm) Plastic DFN –40°C to 85°C LT3506IDHD#PBF LT3506IDHD#TRPBF 3506 16-Lead (5mm x 4mm) Plastic DFN –40°C to 125°C LT3506AIDHD#PBF LT3506AIDHD#TRPBF 3506A 16-Lead (5mm x 4mm) Plastic DFN –40°C to 125°C LT3506EFE#PBF LT3506EFE#TRPBF 3506EFE 16-Lead Plastic TSSOP Narrow –40°C to 85°C LT3506AEFE#PBF LT3506AEFE#TRPBF 3506AEFE 16-Lead Plastic TSSOP Narrow –40°C to 85°C LT3506IFE#PBF LT3506IFE#TRPBF 3506IFE 16-Lead Plastic TSSOP Narrow –40°C to 125°C LT3506AIFE#PBF LT3506AIFE#TRPBF 3506AIFE 16-Lead Plastic TSSOP Narrow –40°C to 125°C LEAD BASED FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3506EDHD LT3506EDHD#TR 3506 16-Lead (5mm x 4mm) Plastic DFN –40°C to 85°C LT3506AEDHD LT3506AEDHD#TR 3506A 16-Lead (5mm x 4mm) Plastic DFN –40°C to 85°C LT3506IDHD LT3506IDHD#TR 3506 16-Lead (5mm x 4mm) Plastic DFN –40°C to 125°C LT3506AIDHD LT3506AIDHD#TR 3506A 16-Lead (5mm x 4mm) Plastic DFN –40°C to 125°C LT3506EFE LT3506EFE#TR 3506EFE 16-Lead Plastic TSSOP Narrow –40°C to 85°C LT3506AEFE LT3506AEFE#TR 3506AEFE 16-Lead Plastic TSSOP Narrow –40°C to 85°C LT3506IFE LT3506IFE#TR 3506IFE 16-Lead Plastic TSSOP Narrow –40°C to 125°C LT3506AIFE LT3506AIFE#TR 3506AIFE 16-Lead Plastic TSSOP Narrow –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ORDER I FOR ATIOU U W

The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VBOOST = 8V, unless otherwise noted. (Note 2)

ELECTRICAL CHARACTERISTICS

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS VIN(MIN) Undervoltage Lockout ● 3.4 3.6 V IINQ Quiescent Current Not Switching 3.8 4.8 mA IINSD Shutdown Current VRUNSS = 0V 30 45 µA VFB Feedback Voltage –40°C to 85°C, EDHD –40°C to 85°C, EFE –40°C to 125°C, IFE, IDHD 792 784 784 800 800 800 808 816 816 mV mV mV IFB FB Pin Bias Current VFB = 800mV, VC = 0.4V ● 40 100 nA VFB(REG) Reference Line Regulation VIN = 5V to 25V 0.005 %/V gmEA Error Amp GM 350 uMhos AV Error Amp Voltage Gain 400 IVC VC Source Current VC Sink Current VFB = 0.6V, VC = 0V VFB = 1.2V, VC = 1100mV µA µA VVC(THRESH) VC Switching Threshold 0.7 V VVC(CLAMP) VC Clamp Voltage 1.9 V fSW Switching Frequency LT3506 LT3506A 500 575 1.1 650 1.2 kHz MHz Switching Phase (Note 5) 180 Deg DC Maximum Duty Cycle LT3506 LT3506A VFB(SWTHRESH) Frequency Shift Threshold on FB 0.4 V fFOLD Foldback Frequency VFB = 0V 170 kHz ISW Switch Current Limit (Note 3) 2.0 2.6 3.6 A VSW(SAT) Switch VCESAT (Note 4) ISW = 1A 210 mV ILSW Switch Leakage Current 10 µA VBOOST(MIN) Minimum Boost Voltage Above Switch ISW = 1A 1.5 2.5 V IBOOST BOOST Pin Current ISW = 1A 20 30 mA IRUN/SS RUN/SS Current 2.1 µA VRUN/SS(THRESH) RUN/SS Threshold 0.3 0.8 V VFB(PGTHRESH) VFB PG Threshold VFB Rising 720 mV VPG(LOW) PG Voltage Output Low VFB = 640mV, IPG = 250µA 0.22 0.4 V ILPG PG Pin Leakage VPG = 2V 0.1 1 µA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3506E/LT3506AE are guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3506I/LT3506AI are guaranteed and tested over the full –40°C to 125°C operating temperature range. Note 3: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at high duty cycle. Note 4: Switch VCESAT guaranteed by design. Note 5: Switching phase is guaranteed by design.

OUTPUT CURRENT (A) EFFICIENCY (%) 1.0

3506 G01

100 VOUT = 1.8V L = 4.7µH (COILCRAFT MSS6122-472MLB) TA = 25°C VIN = 4.5V VIN = 12V VIN = 25V TEMPERATURE (°C) –50 FREQUENCY (kHz) 700 650 600 550 500 FREQUENCY (MHz) 1.20 1.15 1.10 1.05 1.00 –25 0 2 5 5 0

3506 G10

IOUT (A) EFFICIENCY (%) 100 0.4 0.8

3506 G02

1.2 1.6 VOUT = 3.3V L = 6.4µH (SUMIDA CR54-6R4) TA = 25°C VIN = 5V VIN = 25V VIN = 12V IOUT (A) EFFICIENCY (%) 100 0.4 0.8

3506 G03

1.2 1.6 VOUT = 5V L = 10µH (COOPER UP1B-100) TA = 25°C VIN = 8V VIN = 25V VIN = 15V INPUT VOLTAGE (V)* LOAD CURRENT (A) 1.4 1.6 12 14

3506 G04

1.2 1.0 842 6 10 16 1.8 L = 2.2µH L = 1.5µH L = 1µH TA = 25°C INPUT VOLTAGE (V)* LOAD CURRENT (A) 1.4 1.6

3506 G05

1.2 1.0 5 10 15 25 1.8 L = 4.7µH L = 3.3µH L = 2.2µH SLOPE COMPENSATION REQUIRES L > 2.2µH FOR VIN < 7 WITH VOUT = 3.3V TA = 25°C SW CURRENT (A) SWITCH VOLTAGE (mV) 200 300 2.0

3506 G06

0.5 1.0 1.5 400 TA = 25°C SWITCH CURRENT (A) BOOST CURRENT (mA) 2.0

3506 G07

0.5 1.0 1.5

40 TA = 25°C

DUTY CYCLE (%) CURRENT LIMIT (A) TYPICAL

3506 G08

3.0 2.5 2.0 1.5 1.0 0.5 MINIMUM TA = 25°C Efficiency, VOUT = 1.8V (LT3506A) Efficiency, VOUT = 3.3V (LT3506) Efficiency, VOUT = 5V (LT3506) Maximum Load Current, VOUT = 1.8V (LT3506A) Maximum Load Current, VOUT = 3.3V (LT3506A) Switch VCESAT Boost Pin Current Current Limit vs Duty Cycle TYPICAL PERFOR A CE CHARACTERISTICSU W Frequency vs Temperature

TEMPERATURE (°C) –50 RUN/SS CURRENT (µA) 0.5 1.0 1.5 2.0 3.0 –25 0 2 5 5 0

3506 G12

2.5 1.4 1.2 1.0 0.8 0.6 0.4 0.2 RUNN/SS THRESHOLDS (V) TEMPERATURE (°C) –50 25 75

3506 G13

–25 0 50 100 125 TO SWITCH TO RUN IRUN/SS vs Temperature RUN/SS Thresholds vs Temperature TYPICAL PERFOR A CE CHARACTERISTICSU W BOOST1 (Pin 1), BOOST2 (Pin 8): The BOOST pins are used to provide drive voltages, higher than the input voltage, to the internal bipolar NPN power switches. Tie through a diode from VOUT or from VIN. SW1 (Pin 2), SW2 (Pin 7): The SW pins are the outputs of the internal power switches. Connect these pins to the inductors, catch diodes and boost capacitors. VIN1 (Pins 3, 4): The V IN1 pins supply current to the LT3506’s internal regulator and to the internal power switch connected to SW1. These pins must be locally bypassed. VIN2 (Pins 5, 6): The VIN2 pins supply current to the inter- nal power switch connected to SW2 and must be locally bypassed. Connect these pins directly to VIN1 unless power for Channel 2 is coming from a different source. RUN/SS1 (Pin 13), RUN/SS2 (Pin 12): The RUN/SS pins are used to shut down the individual switching regula - tors and the internal bias circuits. They also provide a soft-start function. To shut down either regulator, pull the RUN/SS pin to ground with an open drain or collector. Tie a capacitor from these pins to ground to limit switch current during start-up. If neither feature is used, leave these pins unconnected. PG1 (Pin 14), PG2 (Pin 11): The Power Good pins are the open collector outputs of an internal comparator. PG remains low until the FB pin is within 10% of the final regulation voltage. As well as indicating output regulation, the PG pins can be used to sequence the two switching regulators. These pins can be left unconnected. The PG outputs are valid when VIN is greater than 3.4V and either of the RUN/SS pins is high. The PG comparators are disabled in shutdown. VC1 (Pin 15), VC2 (Pin 10): The VC pins are the outputs of the internal error amps. The voltages on these pins control the peak switch currents. These pins are normally used to compensate the control loops, but can also be used to override the loops. Pull these pins to ground with an open drain to shut down each switching regulator. FB1 (Pin 16), FB2 (Pin 9): The LT3506 regulates each feedback pin to 800mV. Connect the feedback resistor divider taps to these pins. Exposed Pad (Pin 17): The Exposed Pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be soldered to the circuit board for proper operation. PI FU CTIO SU UU

Figure 2. Block Diagram of the LT3506 with Associated External Components (1 of 2 Regulators Shown)

3506 F02

The LT3506 is a dual, constant frequency, current mode buck regulator with internal 2A power switches. The two regulators share common circuitry including voltage reference and oscillator. In addition, the analog blocks on both regulators share the VIN1 supply voltage, but are otherwise independent. This section describes the opera- tion of the LT3506. If the RUN/SS (run/soft-start) pins are both tied to ground, the LT3506 is shut down and draws 30μA from V IN1. Internal 2μA current sources charge external soft-start capacitors, generating voltage ramps at these pins. If either RUN/SS pin exceeds 0.6V, the internal bias circuits turn on, including the internal regulator, 800mV reference and 575kHz master oscillator. In this state, the LT3506 draws 1.8mA from VIN1, whether one or both RUN/SS pins are high. Neither switching regulator will begin to operate until its RUN/SS pin reaches ~0.8V. The master oscillator generates two clock signals of opposite phase. The two switchers are current mode, step-down regulators. This means that instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. This cur - rent mode control improves loop dynamics and provides cycle-by-cycle current limit. The Block Diagram in Figure 2 shows only one of the two switching regulators. A pulse from the slave oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external induc- tor begins to increase. When this current exceeds a level determined by the voltage at V C, current comparator C1 resets the flip-flop, turning off the switch. The current in the inductor flows through the external Schottky diode, and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output voltage by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.75V, and an active clamp of 1.9V limits the output current. The VC pin is also clamped to the RUN/SS pin voltage. As the internal current source charges the external soft-start capacitor, the current limit increases slowly. Each switcher contains an independent oscillator. This slave oscillator is normally synchronized to the master oscillator. However, during start-up, short-circuit or overload conditions, the FB pin voltage will be near zero and an internal comparator gates the master oscillator clock signal. This allows the slave oscillator to run the regulator at a lower frequency. This frequency foldback behavior helps to limit switch current and power dissipation under fault conditions. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient opera- tion. A power good comparator trips when the FB pin is at 90% of its regulated value. The PG output is an open collector transistor that is off when the output is in regulation, al - lowing an external resistor to pull the PG pin high. Power good is valid when the LT3506 is enabled (either RUN/SS pin is high) and VIN is greater than ~3.4V. (Refer to the Block Diagram)

APPLICATIO S I FOR ATIOW U UU FB Resistor Network The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resis- tors according to: R1 = R2(VOUT/0.8 – 1) The parallel combination of R1 and R2 should be 10k or less to avoid bias current errors. Reference designators refer to the Block Diagram in Figure 2. Input Voltage Range The minimum input voltage is determined by either the LT3506’s minimum operating voltage of ~3.6V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: DC = (VOUT + VD)/(VIN – VSW + VD) where VD is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.3V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = (VOUT + VD)/DCMAX - VD + VSW with DCMAX = 0.89 (0.78 for the LT3506A). A more detailed analysis includes inductor loss and the dependence of the diode and switch drop on operating current. A common application where the maximum duty cycle limits the input voltage range is the conversion of 5V to 3.3V. The maximum load current that the LT3506 can deliver at 3.3V depends on the accuracy of the 5V input supply. With a low loss inductor (DCR less than 80mW), the LT3506 can deliver 1.2A for VIN > 4.7V and 1.6A for VIN > 4.85V. The maximum input voltage is determined by the absolute maximum ratings of the VIN and BOOST pins and by the minimum duty cycle DCMIN = 0.08 (0.15 for the LT3506A): VIN(MAX) = (VOUT + VD)/DCMIN – VD + VSW. This limits the maximum input voltage to ~21V with VOUT = 1.2V and ~15V with V OUT = 0.8V. For the LT3506A the maximum input voltage is ~8V with VOUT=0.8V. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the absolute maximum rating. Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L = 2 • (VOUT + VD) for the LT3506 L = (VOUT + VD) for the LT3506A where VD is the voltage drop of the catch diode (~0.4V) and L is in μH. With this value the maximum load current will be ~1.6A, independent of input voltage. The inductor’s RMS current rating must be greater than your maximum load current and its saturation current should be about 30% higher. To keep efficiency high, the series resistance (DCR) should be less than 0.1W. Table 1 lists several vendors and types that are suitable. Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value provides a slightly higher maximum load current, and will reduce the output volt - age ripple. If your load is lower than 1.6A, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher ef- ficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. There are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Also, low inductance may result in dis - continuous mode operation, which may be acceptable, but further reduces maximum load current. For details of maximum output current and discontinuous mode opera- tion, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50%(V OUT/VIN < 0.5), there is a minimum inductance required to avoid subharmonic oscillations. See Application Note 19 for detailed informa- tion on subharmonic oscillations. The following discussion assumes continuous inductor current.

ISWPK = ILPK = IOUT + ΔIL/2. LT3506 will be able to deliver the required output current. when IOUT is less than ΔIL/2 as calculated above. Table 1. Inductors

tor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. With two switchers operating at the same frequency but with different phases and duty cycles, cal- culating the input capacitor RMS current is not simple. However, a conservative value is the RMS input current for the channel that is delivering most power (V OUT • IOUT). This is given by: I I V V V V I INRMS OUT OUT IN OUT IN OUT= −( ) < and is largest when V IN = 2V OUT (50% duty cycle). As the second, lower power channel draws input current, the input capacitor’s RMS current actually decreases as the out-of-phase current cancels the current drawn by the higher power channel. Considering that the maximum load current from a single channel is ~1.6A, RMS ripple current will always be less than 0.8A. The high frequency of the LT3506 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 22μF (less than 10μF for the LT3506A). The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple and the capacitors can handle plenty of ripple current. They are also comparatively robust and can be used in this application at their rated voltage. X5R and X7R types are stable over temperature and applied voltage, and give dependable service. Other types (Y5V and Z5U) have very large tem- perature and voltage coefficients of capacitance, so they may have only a small fraction of their nominal capacitance in your application. While they will still handle the RMS ripple current, the input voltage ripple may become fairly large, and the ripple current may end up flowing from your input supply or from other bypass capacitors in your system, as opposed to being fully sourced from the local input capacitor. An alternative to a high value ceramic capacitor is a lower value along with a larger electrolytic capacitor, for example a 1μF ceramic capacitor in parallel with a low ESR tantalum capacitor. For the electrolytic capacitor, a value larger than 22mF (10mF for the LT3506A) will be required to meet the ESR and ripple current requirements. Because the input capacitor is likely to see high surge currents when the input source is applied, tantalum capacitors should be surge rated. The manufacturer may also recommend operation below the rated voltage of the capacitor. Be sure to place the 1μF ceramic as close as possible to the VIN and GND pins on the IC for optimal noise immunity. A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plug - ging the circuit into a live power source) this tank can ring, doubling the input voltage and damaging the LT3506. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Output Capacitor Selection The output capacitor filters the inductor current to gen - erate an output with low voltage ripple. It also stores energy in order satisfy transient loads and to stabilize the LT3506’s control loop. Because the LT3506 operates at a high frequency, you don’t need much output capacitance. Also, the current mode control loop doesn’t require the presence of output capacitor series resistance (ESR). For these reasons, you are free to use ceramic capacitors to achieve very low output ripple and small circuit size. Estimate output ripple with the following equations: VRIPPLE = ΔIL/(8 • f • COUT) for ceramic capacitors, and VRIPPLE = ΔIL • ESR for electrolytic capacitors (tantalum and aluminum); where ΔIL is the peak-to-peak ripple current in the induc- tor. The RMS content of this ripple is very low, and the RMS current rating of the output capacitor is usually not of concern. Another constraint on the output capacitor is that it must have greater energy storage than the induc - tor; if the stored energy in the inductor is transferred to the output, you would like the resulting voltage step to be small compared to the regulation volt - age. For a 5% overshoot, this requirement becomes COUT > 10L(ILIM/VOUT)2. APPLICATIO S I FOR ATIOW U UU

performance. The last equation gives a good starting point. data sheet and experiment to get the desired performance. value of COUT, you may not be able to tolerate this loss. better transient response for large changes in load current. Table 2 lists several capacitor vendors. Table 2. Low-ESR Surface Mount Capacitors such as the Microsemi UPS120. currents may have more than 1A of average diode current. tifier 20BQ030 (both 2A, 30V) would be good choices.

3.3V, use a small Schottky diode (such as the BAT-54). Absolute Maximum Ratings section. the LT3506 in case VINB is held low while VIN is present.

3506 F03

Figure 3. Generating the Boost Voltage

ILOAD (A) 0.001 INPUT VOLTAGE (V) 4.0 4.5

3506 G14

3.5 3.0 0.01 0.1 1 5.5 5.0 VIN TO START BOOST DIODE TIED TO OUTPUT BOOST DIODE TIED TO INPUT VIN TO RUN TA = 25°C DBOOST = 1N5817 ILOAD (A) 0.001 INPUT VOLTAGE (V) 5.5 6.0

3506 G15

5.0 4.5 0.01 0.1 1 7.0 6.5 VIN TO START BOOST DIODE TIED TO OUTPUT BOOST DIODE TIED TO INPUT VIN TO RUN TA = 25°C DBOOST = 1N5817 The minimum input voltage of an LT3506 application is limited by the minimum operating voltage (<3.6V) and by the maximum duty cycle as outlined above. For proper start-up, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT3506 is turned on with its RUN/SS pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. The plots below show the minimum load current to start and to run as a function of input voltage for 3.3V and 5V outputs. In many cases the discharged output capacitor will present a load to the switcher which will allow it to start. The plots show the worst-case situation where V IN is ramping very slowly. Use a Schottky diode (such as the BAT-54) for the lowest start-up voltage. Minimum Input Voltage, VOUT = 3.3V (LT3506A) Minimum Input Voltage, VOUT = 5V (LT3506A) APPLICATIO S I FOR ATIOW U UU

achieve low output ripple and small circuit size. output current proportional to the voltage at the V C pin. amplifier output current, resulting in two poles in the loop. output capacitor ESR or from a resistor in series with CC. in shutdown mode. They also provide a soft-start function. ~30μA. Internal 2μA current sources pull up on each pin. and the quiescent current increases to ~3.5mA. switch current and therefore input current during start-up. where COUT is the value of the output capacitor. sources will charge these pins to ~2.5V. Figure 4. Circuit Model for Frequency Compensation

3506 F04

and VIN is greater than ~2.4V. be at least twice the value of the capacitor on RUN/SS1. Figure 5. Sequencing the Outputs

3506 F05

to a short circuit or a collapsing input voltage). will oscillate and pull extra current from the input. desired outputs with the lowest step-down ratio possible. than stepping down from 5V to 2.5V. an example of the latter approach. Figure 6. Shorted Input Protection

3506 F06

3506 F07

Figure 7. Subtracting the Current when the Switch is ON (a) From the Current when the Switch in OFF (b) Reveals the Path Switched; Keep these Nodes as Small as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane.

will be held high when the input to the LT3506 is absent. input, as shown in Figure 6. shows the high-di/dt paths in the buck regulator circuit. ideally at the ground terminal of the output capacitor C2. nent placement with trace and via locations. these layers will spread the heat dissipated by the LT3506. Thermal Considerations section.

3506 F08

Figure 8. A Good PCB Layout Ensures Proper Low EMI Operation

APPLICATIO S I FOR ATIOW U UU Single, Low-Ripple 3.2A Output The LT3506 can generate a single, low-ripple 3.2A output if the outputs of the two switching regulators are tied together and share a single output capacitor. By tying the two FB pins together and the two V C pins together, the two channels will share the load current. There are several advantages to this two-phase buck regulator. Ripple cur- rents at the input and output are reduced, reducing volt- age ripple and allowing the use of smaller, less expensive capacitors. Although two inductors are required, each will be smaller than the inductor required for a single-phase regulator. This may be important when there are tight height restrictions on the circuit. The Typical Applications section shows circuits with maximum heights of 1.4mm, 1.8mm and 2.1mm. There is one special consideration regarding the two phase circuit. When the difference between the input voltage and output voltage is less than 2.5V, then the boost circuits may prevent the two channels from properly sharing current. If, for example, channel 1 gets started first, it can supply the load current, while channel 2 never switches enough current to get its boost capacitor charged. In this case, channel 1 will supply the load until it reaches current limit, the output voltage drops, and channel 2 gets started. The solution is to generate a boost supply generated from either SW pin that will service both BOOST pins. The low profile, single output 5V to 3.3V converter shown in the Typical Applications section shows how to do this. Other Linear Technology Publications Application notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design note 100 shows how to generate a dual (+ and –) output supply using a buck regulator

1.8V and 1.2V Outputs with Sequencing 1.8V and ±5V Outputs VIN 4.5V TO 21V

3506 TA01

68µF47µF 22µF 0.22µF 0.22µF 32.4k 18.7k 16.2k 3.3µH 4.7µH VOUT2 1.2V 1.5A VOUT1 1.8V 1.5A 15k 4.7nF 1500pF 1000pF BOOST1 SW1 FB1 VC1 RUN/SS1 RUN/SS2 PGOOD2 BOOST2 SW2 FB2 VC2 PGOOD1 LT3506 VIN2VIN1 GND OUTPUT CURRENTS CAN INCREASE TO 1.6A WHEN VIN>12V.D1, D2: ON SEMICONDUCTOR MBRS230LT3 D3: BAT-54A L1: COILCRAFT MSS6122-472 L2: TDK SLF7028-3R3M PGOOD D1 D2 D3a D3b 100k 20k15k VIN 7V TO 25V

3506 TA02

22µF 2.2nF 47µF 22µF 22µF 0.22µF 0.22µF 2.2µF 13.3k 18.7k 69.8k 4.7µH VOUT2 0.6A VOUT1 1.8V 1.5A 15k 47k 15k 15k 100k 4.7nF BOOST1 SW1 FB1 VC1 RUN/SS1 PGOOD2 PGOOD1 BOOST2 SW2 FB2 VC2 RUN/SS2 LT3506 VIN2VIN1 GND L1: COILCRAFT MSS6122-472 L2: COILTRONICS CTX5-1A IOUT3 SHOULD NEVER EXCEED 1/2 OF IOUT2. SEE DESIGN NOTE 100 FOR DETAILS ON GENERATING DUAL OUTPUTS USING A BUCK REGULATOR. D1: ON SEMICONDUCTOR MBRS230LT3 D2: ON SEMICONDUCTOR MBRM130LT3 D3: BAT-54A D4: ON SEMICONDUCTOR MBR0530 PGOOD D1 D2 D3a D3b VOUT2 4.7µH VOUT3 –5V 0.3AD4 1500pF 1500pF

Low Ripple, Low Profile 1.2V, 3A Converter, Maximum Height = 2mm Two Stage Step Down, Up to 25V Input to 1.2V Output VIN 4.5V TO 21V

3506 TA03

68µF 22µF 0.22µF 0.22µF 32.4k 16.2k 20k 1000pF 4.1µH 4.1µH VOUT2 1.2V 100k 4.7nF VC1 VC2 RUN/SS1 RUN/SS2 PGOOD1 PGOOD2 BOOST1 SW1 BOOST2 SW2 FB1 FB2 LT3506 VIN2VIN1 GND D1, D2: DIODES, INC. B230A D3: BAT-54A L1, L2: SUMIDA CDRH5D18-4R1 PGOOD D3a D3b VOUT VIN 8V TO 25V

3506 TA04

68µF47µF 22µF 0.22µF 0.22µF 32.4k 100k 69.8k 16.2k 2.2µH 10µH VOUT2 1.2V 1.5A VOUT1 13.3k 15k 20k 4.7nF BOOST1 SW1 FB1 VC1 RUN/SS1 RUN/SS2 BOOST2 SW2 FB2 VC2 PGOOD1 PGOOD2 LT3506 VIN2VIN1 GND L1: COOPER UP1B-100 L2: COOPER UP0.4C-2R2 D1, D2: ON SEMICONDUCTOR MBRS230LT3 D3: BAT-54A PGOOD D1 D2 D3a D3b 1500pF 1000pF

Low Ripple, Low Profile 0.8V, 3A Converter, Maximum Height = 1mm VIN 3.6V TO 8V

3506 TA05

68µF 22µF 0.1µF 0.1µF 10k 1.5µH 1.5µH VOUT 0.8V 100k 4.7nF VC1 VC2 RUN/SS1 RUN/SS2 PGOOD1 PGOOD2 BOOST1 SW1 BOOST2 SW2 FB1 FB2 LT3506AEDHD VIN2VIN1 GND D1, D2: DIODES, INC. DFLS230L D3: BAT-54AT L1, L2: COILCRAFT LPO6610-152ML PGOOD D3a D3b VOUT 20k 1000pF Low Profile 1.8V and 1.3V Outputs with Sequencing, Maximum Height = 1.2mm VIN 4.5V TO 10V

3506 TA06

47µF22µF 10µF 0.1µF 0.1µF 28k 18.7k 17.4k 2.2µH 2.2µH VOUT2 1.3V 1.6A VOUT1 1.8V 1.5A 15k 4.7nF 1.5nF 1.5nF BOOST1 SW1 FB1 VC1 RUN/SS1 RUN/SS2 PGOOD2 BOOST2 SW2 FB2 VC2 PGOOD1 LT3506AEDHD VIN2VIN1 GND D1, D2: DIODES, INC. DFLS230L D3: BAT-54AW L1, L2: COILCRAFT LPS4012-222 PGOOD D1 D2 D3a D3b 100k 15k10k

4.00 ±0.10 (2 SIDES) 5.00 ±0.10 (2 SIDES) NOTE: 1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WJGD-2) IN JEDEC PACKAGE OUTLINE MO-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.40 ± 0.10 BOTTOM VIEW—EXPOSED PAD 2.44 ± 0.10 (2 SIDES) 0.75 ±0.05 R = 0.115 TYP R = 0.20 TYP 4.34 ±0.10 (2 SIDES) 169 PIN 1 TOP MARK (SEE NOTE 6)

0.200 REF

0.00 – 0.05 (DHD16) DFN 0504 0.25 ± 0.05 PIN 1 NOTCH

0.50 BSC

4.34 ±0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 2.44 ±0.05 (2 SIDES) 3.10 ±0.05 0.70 ±0.05 4.50 ±0.05 PACKAGE OUTLINE 0.25 ± 0.05 PAckAge DescRiPtion 16-Lead Plastic DFN (5mm × 4mm) (Reference LTC DWG # 05-08-1707)

Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa- tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. PAckAge DescRiPtion 16-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation BA FE16 (BA) TSSOP 0204 0.09 – 0.20 (.0035 – .0079) 0° – 8° 0.25 REF 0.50 – 0.75 (.020 – .030) 4.30 – 4.50* (.169 – .177) 1 3 4 5 6 7 8 10 9 4.90 – 5.10* (.193 – .201) 16 1514 13 12 11 1.10 (.0433) MAX 0.05 – 0.15 (.002 – .006) 0.65 (.0256) BSC 2.74 (.108) 2.74 (.108) 0.195 – 0.30 (.0077 – .0118) TYP MILLIMETERS (INCHES) *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS 2. DIMENSIONS ARE IN RECOMMENDED SOLDER PAD LAYOUT 3. DRAWING NOT TO SCALE 0.45 ±0.05

0.65 BSC

4.50 ±0.10 6.60 ±0.10 1.05 ±0.10 2.74 (.108) 2.74 (.108) SEE NOTE 4 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT 6.40 (.252) BSC

Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com  LINEAR TECHNOLOGY CORPORATION 2006 LT 0807 REV B • PRINTED IN USA PART NUMBER DESCRIPTION COMMENTS LT1765 25V, 2.75A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, S8, TSSOP16E Packages LT1766 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, TSSOP16/TSSOP16E Packages LT1767 25V, 1.2A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, MS8, MS8E Packages LT1940/LT1940L Dual Monolithic 1.4A, 1.1MHz Step- Down Switching Regulator VIN: 3.6V to 25V, VOUT(MIN) = 1.25V, IQ = 3.8mA, TSSOP16E Packages LTC3407/LTC3407-2 Dual 600mA/800mA, 1.5MHz, Synchronous Step-Down Regulator VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40mA, MSE Package LT3493 1.2A, 750kHz Step-Down Switching Regulator in 2mm × 3mm DFN VIN: 3.6V to 36V, VOUT(MIN) = 0.78V, IQ = 1.9mA, 2mm × 3mm DFN Package LT3505 1.2A, 3MHz Step-Down Switching Regulator in 3mm × 3mm DFN VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 2mA, DFN or MSE10 Package LTC3548 Dual 800mA and 400mA, 2.25MHz, Synchronous Step-Down Regulator VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, 3mm × 3mm DFN or MSE10 Package LTC3549 Dual 300mA, 2.25MHz, Synchronous Step-Down Regulator VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, 3mm × 3mm DFN Package LTC3701 Two Phase, Dual, 500kHz, Constant Frequency, Current Mode, High Efficiency Step-Down DC/DC Controller VIN: 2.5V to 10V, VOUT(MIN) = 0.8V, IQ = 460µA, SSOP-16 Package LTC3736 Dual Two Phase, No RSENSE™, Synchronous Controller with Output Tracking VIN: 2.75V to 9.8V, VOUT(MIN) = 0.6V, IQ = 300µA, 4mm × 4mm QFN or SSOP-24 Packages LTC3737 Dual Two Phase, No RSENSE DC/DC Controller with Output Tracking VIN: 2.75V to 9.8V, VOUT(MIN) = 0.6V, IQ = 220µA, 4mm × 4mm QFN or SSOP-24 Packages No RSENSE is a trademark of Linear Technology Corporation. RelAteD PARts