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Technical content

n Isolated PFC Flyback with Minimum Number of External Components n VIN and VOUT Limited Only by External Components n Active Power Factor Correction n Low Harmonic Distortion n No Opto-Coupler Required n Constant-Current and Constant-Voltage Regulation n Accurate Regulated Voltage and Current (±5% Typical) n Energy Star Compliant (<0.5W No Load Operation) n Thermally Enhanced 16-lead MSOP Package TYPICAL APPLICATION FEATURES DESCRIPTION Isolated No Opto-Coupler Flyback Controller with Active PFC The L T®3798 is a constant-voltage/constant-current iso- lated flyback controller that combines active power factor correction (PFC) with no opto-coupler required for output voltage feedback into a single-stage converter. A L T3798 based design can achieve a power factor of greater than 0.97 by actively modulating the input current, allowing compli- ance with most Harmonic Current Emission requirements. The L T3798 is well suited for a wide variety of off-line applications. The input range can be scaled up or down, depending mainly on the choice of external components. Efficiencies higher than 86% can be achieved with output power levels up to 100W. In addition, the L T3798 can easily be designed into high DC input applications. Universal Input 24W PFC Bus Converter VOUT vs IOUT

APPLICATIONS

n Offline 5W to 100W+ Applications n High DC V IN Isolated Applications n Offline Bus Converter (12V , 24V or 48V Outputs) L, L T , L TC, L TM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499 and 7471522.

3798 TA01a

VIN_SENSE EN/UVLO VIN DCM FB VREF CTRL2 CTRL1 OVP GATE SENSE INTVCC GND L T3798 COMP+VC COMP– 560μF × 2 4.7pF 10μF 2.2nF 4:1:1 2.2μF 0.1μF 20Ω 20Ω 0.05Ω 499k 499k 95.3k 22pF 100k 100k 4.99k 90.9k CTRL3 90V TO 265V AC 0.1μF 40.2k100k 16.5k221k 4.7μF 24V IOUT (A)

23.50 VOUT (V)

24.25 24.00 23.75 24.50 0.6 0.8 10.2 0.4

3798 TA01b

VAC = 265V VAC = 220V VAC = 120V VAC = 90V

PIN CONFIGURATION ABSOLUTE MAXIMUM RATINGS FB, V V Operating Temperature Range (Note 2) (Note 1) CTRL1 CTRL2 CTRL3 V REF OVP VC COMP+ COMP– V IN_SENSE SENSE GATE INTV CC EN/UVLO V IN DCM FB TOP VIEW GND MSE PACKAGE 16-LEAD PLASTIC MSOP θJA = 50°C/W , θJC = 10°C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE L T3798EMSE#PBF L T3798EMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 125°C L T3798IMSE#PBF L T3798IMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 125°C Consult L TC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container . For more information on lead free part marking, go to: http://www.linear .com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear .com/tapeandreel/ PARAMETER CONDITIONS MIN TYP MAX UNITS Input Voltage Range 10 38 V VIN Quiescent Current VEN/UVLO = 0.2V VEN/UVLO = 1.5V , Not Switching 45 60 70 μA μA VIN Quiescent Current, INTVCC Overdriven V INTVCC = 11V 60 μA VIN Shunt Regulator Voltage I = 1mA 40 V VIN Shunt Regulator Current Limit 8m A INTVCC Quiescent Current VEN/UVLO = 0.2V VEN/UVLO = 1.5V , Not Switching 12.5 1.8 15.5 2.2 17.5 2.7 μA mA EN/UVLO Pin Threshold EN/UVLO Pin Voltage Rising l 1.21 1.25 1.29 V EN/UVLO Pin Hysteresis Current EN/UVLO=1V 8 10 12 μA VIN_SENSE Threshold Turn Off 27 μA VREF Voltage 0 μA Load 200μA Load l l 1.97 1.95 2.0 1.98 2.03 2.03 V V CTRL1/CTRL2/CTRL3 Pin Bias Current CTLR1/CTRL2/CTRL3 = 1V ±30 nA ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 18V , INTVCC = 11V , unless otherwise noted.

ELECTRICAL CHARACTERISTICS

Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The L T3798E is guaranteed to meet specified performance from 0°C to 125°C junction temperature. Specification over the –40°C and The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 18V , INTVCC = 11V , unless otherwise noted. 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The 3798I is guaranteed to meet specified performance from –40°C to 125°C operating junction temperature range. Note 3: Current flows out of the FB pin. PARAMETER CONDITIONS MIN TYP MAX UNITS SENSE Current Limit Threshold VIN_SENSE = 150μA 96 102 107 mV Minimum SENSE Current Limit Threshold V IN_SENSE = 34μA 14 mV Minimum SENSE Current Limit Threshold V IN_SENSE = 21μA 4 mV SENSE Input Bias Current Current Out of Pin, SENSE = 0V 15 μA FB Voltage l 1.22 1.25 1.28 V FB voltage Line Regulation 10V < V IN < 35V 0.01 0.03 %/V FB Pin Bias Current (Note 3), FB = 1V 4.05 4.25 4.4 μA FB Error Amplifier Voltage Gain ΔVVC/ΔVFB, CTRL1=1V , CTRL2=2V , CTRL3=2V 180 V/V FB Error Amplifier T ransconductance ΔI = 5μA 170 UMHOS Current Error Amplifier Voltage Gain ΔV COMP+/ΔVCOMP–, CTRL1 = 1V , CTRL2 = 2V , CTRL3 = 2V 100 V/V Current Error Amplifier T ransconductance ΔI = 5μA 50 UMHOS Current Loop Voltage Gain ΔVCTRL/ΔVSENSE,1000pF Cap from COMP+ to COMP– 21 V/V DCM Current Turn-On Threshold Current Out of Pin 80 μA Maximum Oscillator Frequency COMP + = 0.95V , VIN_SENSE = 150μA 150 kHz Minimum Oscillator Frequency COMP+ = 0V , VFB <VOVP 4k Hz Minimum Oscillator Frequency COMP+ = 0V , VFB >VOVP 0.5 kHz Backup Oscillator Frequency 20 kHz Linear Regulator INTV CC Regulation Voltage No Load 9.8 10 10.4 V Dropout (VIN-INTVCC)I INTVCC = –10mA, VIN = 10V 500 900 mV Current Limit Below Undervoltage Threshold 12 25 mA Current Limit Above Undervoltage Threshold 80 120 mA Gate Driver t r GATE Driver Output Rise Time CL = 3300pF , 10% to 90% 18 ns tf GATE Driver Output Fall Time CL = 3300pF , 90% to 10% 18 ns GATE Output Low (VOL) 0.01 V GATE Output High (VOH) INTVCC- 50mV V

TYPICAL PERFORMANCE CHARACTERISTICS VREF vs Temperature VREF vs VIN SENSE Current Limit Threshold vs Temperature Minimum Oscillator Frequency vs Temperature Backup Oscillator Frequency vs Temperature Maximum Oscillator Frequency vs Temperature EN/UVLO Threshold vs Temperature EN/UVLO Hysteresis Current vs Temperature V IN IQ vs Temperature TA = 25°C, unless otherwise noted. TEMPERATURE (°C) –50 0 50 75–25 25 100 125 150

3798 G01

1.2 EN/UVLO (V) 1.22 1.26 1.24 1.28 1.3 RISING FALLING TEMPERATURE (°C) –50 0 50 75–25 25 100 125 150

3798 G02

10.5EN/UVLO HYSTERESIS CURRENT (μA) 11.5 TEMPERATURE (°C) –50 0 50 75–25 25 100 125 150

3798 G03

IQ (μA) 100 VIN = 12V VIN = 24V TEMPERATURE (°C) –50

1.900 VREF (V)

1.925 1.975 2.000 2.025 2.100 2.075 0 50 75 1.950 2.050 –25 25 100 125 150

3798 G05

VIN = 24V WITH NO LOAD VIN = 24V WITH 200μA LOAD VIN (V) 1.95 VREF (V) 1.96 1.98 1.99 2.01 2.05 2.03 2.04 1.97 2.02 15 2520 30 35 40 200μA LOAD TEMPERATURE (°C) –50 CURRENT LIMIT (mA) 100 120 0 50 75 –25 25 100 125 150

3798 G06

MIN ILIM VIN_SENSE = 34μA MIN ILIM VIN_SENSE = 21μA TEMPERATURE (°C) –50

120 FREQUENCY (kHz)

–25 25 100 125 150

3798 G06a

TEMPERATURE (°C) –50 FREQUENCY (kHz) 0 50 75 –25 25 125 100150

3799 G07

VFB < VOVP VOVP > VFB TEMPERATURE (°C) –50 FREQUENCY (kHz) 0 50 75 –25 25 125 100150

3798 G07a

TYPICAL PERFORMANCE CHARACTERISTICS INTVCC vs Temperature INTVCC vs VIN VIN Shunt Voltage vs Temperature Maximum Shunt Current vs Temperature TA = 25°C, unless otherwise noted. Leakage Inductance Blanking Time vs SENSE Current Limit Threshold TEMPERATURE (°C) –50

9.5 INTVCC (V)

9.75 10.25 10.5 0 50 75 –25 25 100 125 150

3798 G08

VIN (V) 51 5 25 3010 20 35 40

3798 G09

INTVCC (V) 9.2 9.4 9.6 9.8 10.2 TEMPERATURE (°C) –50 0 50 75–25 25 100 125 150

3798 G10

VIN SHUNT VOLTAGE (V) 39.5 40.5 41.5 ISHUNT = 1mA TEMPERATURE (°C) –50 0 50 75–25 25 100 125 150

3798 G11

SHUNT CURRENT (mA) SENSE CURRENT LIMIT THRESHOLD (mV) 0 60 8020 40 100 120

3798 G12

0.2LEAKAGE INDUCTANCE BLANKING TIME (μs) 0.4 0.6 0.8 1.2 1.4 1.6 1.8 Output Voltage vs Input Voltage VOUT vs Temperature Output Current vs Input Voltage VIN (VAC) 23.6 VOUT (V) 23.8 24.6 24.2 170 210 230 24.4 150 130 110190 250 270

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PAGE 17 SCHEMATIC: UNIVERSAL VIN (VAC) 0.90 OUTPUT CURRENT (A)0.95 1.00 1.05 1.10 210 230130110 170 150 190 250 270

3798 G14

PAGE 17 SCHEMATIC: UNIVERSAL VOUT = 22V TEMPERATURE (°C) –50 VOUT (V)24 24.5 0 50 75 23.5 –25 25 125 100150

3798 G012a

PAGE 17 SCHEMATIC: UNIVERSAL VAC = 120V VAC = 220V

CTRL1, CTRL2, CTRL3 (Pin 1, Pin 2, Pin 3): Current Output Adjustment Pins. These pins control the output current. The lowest value out of the three CTRL inputs is compared to negative input of the operational amplifier. V REF (Pin 4): Voltage Reference Output Pin. Typically 2V . This pin drives a resistor divider for the CTRL pin, either for analog dimming or for temperature limit/compensation of output load. Can supply up to 200μA. OVP (Pin 5): Overvoltage Protection. This pin accepts a DC voltage to compare to the sample and hold’s voltage output information. When output voltage information is above the OVP, the part divides the minimum switching frequency by 8, around 500Hz. This protects devices con- nected to the output. This also allows the part to operate with very little power consumption with no load to meet energy star requirements. VC (Pin 6): Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate the switching regulator . A 100pF capacitor in parallel helps eliminate noise. COMP +, COMP– (Pin 7, Pin 8): Compensation Pins for Internal Error Amplifier . Connect a capacitor between these two pins to compensate the internal feedback loop. FB (Pin 9): Voltage Loop Feedback Pin. FB is used to regulate the output voltage by sampling the third wind- ing. If the converter is used in current mode, the FB pin will normally be at a voltage level lower than 1.25V , and will reach the steady state of 1.25V if it detects an open output condition. DCM (Pin 10): Discontinuous Conduction Mode Detection Pin. Connect a capacitor and resistor in series with this pin to the third winding. V IN (Pin 11): Input Voltage. This pin supplies current to the internal start-up circuitry and to the INTVCC LDO. This pin must be locally bypassed with a capacitor . A 42V shunt regulator is internally connected to this pin. EN/UVLO (Pin 12): Enable/Undervoltage Lockout. A resis- tor divider connected to VIN is tied to this pin to program the minimum input voltage at which the L T3798 will turn on. When below 1.25V , the part will draw 60μA with most of the internal circuitry disabled and a 10μA hysteresis current will be pulled out of the EN/UVLO pin. When above 1.25V , the part will be enabled and begin to switch and the 10μA hysteresis current is turned off. INTV CC (Pin 13): Regulated Supply for Internal Loads and GATE Driver. Supplied from VIN and regulates to 10V (typical). INTVCC must be bypassed with a 4.7μF capacitor placed close to the pin. PIN FUNCTIONS Power Factor vs Input Voltage Efficiency vs Input Voltage VIN (VAC) 0.90 0.92 0.91 0.95 0.96 0.97 0.98 0.99 1.00 150 210 190230 0.93 0.94 130 110 170 250 270

3798 G15

PAGE 17 SCHEMATIC: UNIVERSAL VIN (VAC) EFFICIENCY (%) 100 150 210 190 230130 110 170 250 270

3798 G16

PAGE 17 SCHEMATIC: UNIVERSAL TYPICAL PERFORMANCE CHARACTERISTICSTA = 25°C, unless otherwise noted.

VOUT– VRECTIFIED C7L1A N:1 L1B SW1 1.22V VIN R14 R13 L1C t R15 VIN_SENSEEN/UVLODCM STARTUP INTERNAL REG VREF OVP FB S&H COMP+ COMP– CTRL1 CTRL3 CTRL2 FB S&H VC 1.22V R10 A5 A6MINIMUM MUL TIPLIER R12 600mV ONE SHOT CURRENT COMPARATOR R Q S S MASTER LATCH LOW OUTPUT CURRENT OSCILLATOR R11 DRIVER GND GATE INTVCC SENSE 3798 BD GATE (Pin 14): N-Channel FET Gate Driver Output. Switches between INTVCC and GND. Driven to GND during shutdown state and stays high during low voltage states. SENSE (Pin 15): The Current Sense Input for the Control Loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor , R SENSE, in the source of the NFET . The negative terminal of the current sense resistor should be connected to the GND plane close to the IC. V IN_SENSE (Pin 16): Line Voltage Sense Pin. The pin is used for sensing the AC line voltage to perform power factor correction. Connect a resistor in series with the line voltage to this pin. If no PFC is needed, connect this pin to INTV CC with a 25k resistor. GND (Exposed Pad Pin 17): Ground. The exposed pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The exposed pad must be soldered to the circuit board for proper operation.

The L T3798 is a current mode switching controller IC designed specifically for generating a constant current/ constant voltage supply in an isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage and cur- rent on the isolated secondary side of the transformer must be communicated to the primary side in order to maintain regulation. Historically, this has been done with an opto-isolator . The L T3798 uses a novel method of using the external MOSFETs peak current information from the sense resistor to calculate the output current of a flyback converter without the need of an opto-coupler . Active power factor correction is becoming a requirement for offline power supplies and the power levels are de- creasing. A power factor of one is achieved if the current drawn is proportional to the input voltage. The L T3798 modulates the peak current limit with a scaled version of the input voltage. This technique can provide power factors of 0.97 or greater . The Block Diagram shows an overall view of the system. The external components are in a flyback topology configura- tion. The third winding senses the output voltage and also supplies power to the part in steady-state operation. The V IN pin supplies power to an internal LDO that generates 10V at the INTVCC pin. The novel control circuitry consists of two error amplifiers, a minimum circuit, a multiplier , a transmission gate, a current comparator , a low output current oscillator and a master latch, which will be ex- plained in the following sections. The part also features a sample-and-hold to sample the output voltage from the third winding. A comparator is used to detect discontinu- ous conduction mode (DCM) with a cap connected to the third winding. The part features a 1.9A gate driver. The L T3798 is designed for both off-line and DC applica- tions. The EN/UVLO and a resistor divider can be configured for a micropower hysteretic start-up. In the Block Diagram, R3 is used to stand off the high voltage supply voltage. The internal LDO starts to supply current to the INTV CC when VIN is above 2.5V . The VIN and INTVCC capacitors are charged by the current from R3. When V IN exceeds the turn-on threshold and INTVCC is in regulation at 10V , the OPERATION part begins to switch. The VIN hysteresis is set by the EN/ UVLO resistor divider . The third winding provides power to VIN when its voltage is higher than the V IN voltage. A voltage shunt is provided for fault protection and can sink 8mA of current when V IN is over 40V . During a typical cycle, the gate driver turns the external MOSFET on and a current flows through the primary wind- ing. This current increases at a rate proportional to the input voltage and inversely proportional to the magnetizing inductance of the transformer . The control loop determines the maximum current and the current comparator turns the switch off when the current level is reached. When the switch turns off, the energy in the core of the transformer flows out the secondary winding through the output diode, D1. This current decreases at a rate proportional to the output voltage. When the current decreases to zero, the output diode turns off and voltage across the secondary winding starts to oscillate from the parasitic capacitance and the magnetizing inductance of the transformer . Since all windings have the same voltage across them, the third winding rings too. The capacitor connected to the DCM pin, C1, trips the comparator A2, which serves as a dv/dt detector , when the ringing occurs. This timing information is used to calculate the output current and will be described below. The dv/dt detector waits for the ringing waveform to reach its minimum value and then the switch turns back on. This switching behavior is similar to zero volt switching and minimizes the amount of energy lost when the switch is turned back on and improves efficiency as much as 5%. Since this part operates on the edge of continuous conduction mode and discontinuous conduction mode, the operating mode is called critical conduction mode (or boundary conduction mode). Primary Side Control Loops The L T3798 achieves constant current/constant voltage operation by using two separate error amplifiers. These two amplifiers are then fed to a circuit that outputs the lower voltage of the two, shown as the "minimum" block in the Block Diagram. This voltage is converted to a current before being fed into the multiplier .

formation available on the primary side of the transformer . time and a height of the peak secondary winding current. node thus increasing the current comparator input. the output voltage control loop. Figure 1. Secondary Diode Current and Switch Waveforms

3798 F01

IN_SENSE to INTVCC instead of the AC line voltage. capable of 8mA(typical) of current at room temperature. pins voltage is below 1.25V and 0μA when above 1.25V . Figure 2. The UVLO threshold for VIN rising is: diode, and ITC is the internal current source for the FB pin. drop of the output rectifying diode.

3798 F02

Figure 2. Undervoltage Lockout (UVLO)

Programming Output Current The maximum output current depends on the supply volt- age and the output voltage in a flyback topology. With the V IN_SENSE pin connected to 100μA current source and a DC supply voltage, the maximum output current is determined at the minimum supply voltage, and the maximum output voltage using the following equation: IOUT(MAX) = 2•(1– D)• NPS 42•R SENSE where D = VOUT •NPS VOUT •NPS + VIN The maximum control voltage to achieve this maximum output current is 2V • (1-D). It is suggested to operate at 95% of these values to give margin for the part’s tolerances. When designing for power factor correction, the output current waveform is going to have a half sine wave squared shape and will no longer be able to provide the above currents. By taking the integral of a sine wave squared over half a cycle, the average output current is found to be half the value of the peak output current. In this case, the recommended maximum average output current is as follows: IOUT(MAX) = 2•(1−D) • NPS 42•R SENSE

• 47.5% where D = VOUT •NPS VOUT •NPS + VIN The maximum control voltage to achieve this maximum output current is (1-D) • 47.5%. For control voltages below the maximum, the output cur- rent is equal to the following equation: IOUT = CTRL • NPS 42•R SENSE The VREF pin supplies a 2V reference voltage to be used with the control pins. To set an output current, a resistor divider is used from the 2V reference to one of the control pins. The following equation sets the output current with a resistor divider: R1= R2 2NPS 42•I OUT •R SENSE where R1 is the resistor connected to the VREF pin and the CTRL pin and R2 is the resistor connected to the CTRL pin and ground. Setting VIN_SENSE Resistor The VIN_SENSE resistor sets the current feeding the internal multiplier that modulates the current limit for power factor correction. At the maximum line voltage, VMAX, the current is set to 360μA. Under this condition, the resistor value is equal to (V MAX/360μA). For DC input or non-PFC AC input applications, connect a 25k resistor from V IN_SENSE to INTV CC instead of the AC line voltage. Critical Conduction Mode Operation Critical conduction mode is a variable frequency switching scheme that always returns the secondary current to zero with every cycle. The L T3798 relies on boundary mode and discontinuous mode to calculate the critical current because the sensing scheme assumes the secondary current returns to zero with every cycle. The DCM pin uses a fast current input comparator in combination with a small capacitor to detect dv/dt on the third winding. To eliminate false trip- ping due to leakage inductance ringing, a blanking time of between 600ns and 2μs is applied after the switch turns off, depending on the current limit shown in the Leakage In- ductance Blanking Time vs SENSE Current Limit Threshold curve in the Typical Performance Characteristics section. The detector looks for 80μA of current through the DCM pin due to falling voltage on the third winding when the secondary diode turns off. This detection is important since the output current is calculated using this comparator’s output. This is not the optimal time to turn the switch on because the switch voltage is still close to V IN + VOUT • NPS and would waste all the energy stored in the parasitic ca- pacitance on the switch node. Discontinuous ringing begins when the secondary current reaches zero and the energy in the parasitic capacitance on the switch node transfers OPERATION

to the input capacitor . This is a second-order network composed of the parasitic capacitance on the switch node and the magnetizing inductance of the primary winding of the transformer . The minimum voltage of the switch node during this discontinuous ring is V IN – VOUT • NPS. The L T3798 turns the switch back on at this time, during the discontinuous switch waveform, by sensing when the slope of the switch waveform goes from negative to positive using the dv/dt detector . This switching technique may increase efficiency by 5%. Sense Resistor Selection The resistor , R SENSE, between the source of the external N-channel MOSFET and GND should be selected to provide an adequate switch current to drive the application without exceeding the current limit threshold. For applications without power factor correction, select a resistor according to: RSENSE = 2(1– D)NPS IOUT •4 2 •95% where D = VOUT •NPS VOUT •NPS + VIN For applications with power factor correction, select a resistor according to: RSENSE = 2(1– D)NPS IOUT •4 2 • 47.5% where D = VOUT •NPS VOUT •NPS + VIN Minimum Current Limit The L T3798 features a minimum current limit of approxi- mately 18% of the peak current limit. This is necessary when operating in critical conduction mode since low current limits would increase the operating frequency to a very high frequency. The output voltage sensing circuitry needs a minimum amount of flyback waveform time to sense the output voltage on the third winding. The time needed is 350ns. The minimum current limit allows the use of smaller transformers since the magnetizing primary inductance does not need to be as high to allow proper time to sample the output voltage information. To help improve crossover distortion of the line input current, a second minimum current limit of 6% becomes active when the V IN_SENSE current is lower than 27μA. Since the off-time becomes very short with this lower minimum current limit, the sample-and-hold is deactivated. Universal Input The L T3798 operates over the universal input voltage range of 90VAC to 265VAC. In the Typical Performance Characteristics section, the Output Voltage vs V IN and the Output Current vs V IN graphs, show the output voltage and output current line regulation for the first application picture in the Typical Applications section. Selecting Winding Turns Ratio Boundary mode operation gives a lot of freedom in selecting the turns ratio of the transformer . We suggest to keep the duty cycle low, lower N PS, at the maximum input voltage since the duty cycle will increase when the AC waveform decreases to zero volts. A higher N PS increases the output current while keeping the primary current limit constant. Although this seems to be a good idea, it comes at the expense of a higher RMS current for the secondary-side diode which might not be desirable because of the primary side MOSFET’s superior performance as a switch. A higher N PS does reduce the voltage stress on the secondary-side diode while increasing the voltage stress on the primary- side MOSFET. If switching frequency at full output load is kept constant, the amount of energy delivered per cycle by the transformer also stays constant regardless of the N PS. Therefore, the size of the transformer remains the same at practical N PS’s. Adjusting the turns ratio is a good way to find an optimal MOSFET and diode for a given application.

ratio. An RCD clamp can be used in place of the TVS clamp. Figure 3. TVS & RCD Switch Voltage Clamps

3798 F03

decrease the amount of ringing by using an RC snubber . Figure 4. Observed Waveforms at MOSFET Drain when

3798 F04

value of the capacitance will decrease the overshoot. be tightly controlled to ensure a consistent output current. Table 1. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted twice the line frequency for PFC to work properly. negative input and output of the operational amplifier . application, the compensation capacitor is 0.1μF. below the switching frequency for optimal performance.

MOSFET and Diode Selection With a strong 1.9A gate driver , the L T3798 can effectively drive most high voltage MOSFETs. A low Qg MOSFET is recommended to maximize efficiency. In most applications, the R DS(ON) should be chosen to limit the temperature rise of the MOSFET . The drain of the MOSFET is stressed to V OUT • NPS + VIN during the time the MOSFET is off and the secondary diode is conducting current. But in most applications, the leakage inductance voltage spike exceeds this voltage. The voltage of this stress is determined by the switch voltage clamp. Always check the switch waveform with an oscilloscope to make sure the leakage inductance voltage spike is below the breakdown voltage of the MOS- FET. A transient voltage suppressor and diode are slower than the leakage inductance voltage spike, therefore causing a higher voltage than calculated. The secondary diode stress may be as much as V OUT + 2

• VIN/NPS due to the anode of the diode ringing with the secondary leakage inductance. An RC snubber in parallel with the diode eliminates this ringing, so that the reverse voltage stress is limited to V OUT + VIN/NPS. With a high NPS and output current greater than 3A, the IRMS through the diode can become very high and a low forward drop Schottky is recommended. Discontinuous Mode Detection The discontinuous mode detector uses AC-coupling to detect the ringing on the third winding. A 22pF capacitor with a 30k resistor in series is recommended in most designs. Depending on the amount of leakage inductance ringing, an additional current may be needed to prevent false tripping from the leakage inductance ringing. A resis- tor from INTV CC to the DCM pin adds this current. Up to an additional 100μA of current may be needed in some cases. The DCM pin is roughly 0.7V , therefore the resistor value is selected using the following equation: R = 10V – 0.7V I where I is equal to the additional current into the DCM pin. Power Factor Correction/Harmonic Content The L T3798 attains high power factor and low harmonic content by making the peak current of the main power switch proportional to the line voltage by using an internal multiplier . A power factor of >0.97 is easily attainable for most applications by following the design equations in this data sheet. With proper design, L T3798 applications can easily meet most harmonic standards. Operation Under Light Output Loads The L T3798 detects output overvoltage conditions by looking at the voltage on the third winding. The third winding voltage is proportional to the output voltage when the main power switch is off and the secondary diode is conducting current. Sensing the output voltage requires delivering power to the output. When the output current is very low, this periodic delivery of output current can exceed the load current. The OVP pin sets the output overvolt- age threshold. When the output of the sample-and-hold is above this voltage, the minimum switching frequency is divided by 8 as shown in Figure 5. This OVP threshold needs to be set above 1.35V and should be set out of the way of output voltage transients. The output clamp point is set with the following formula: V OUT = VOVP(R4 + R5)/(NST • R5)–(VF + (R4•ITC)/NST) The VOVP pin voltage may be provided by a resistor divider from the VREF pin. This frequency division greatly reduces the output current delivered to the output but a Zener or resistor is required to dissipate the remaining output cur- rent. The Zener diode’s voltage needs to be 5% higher than the output voltage set by the resistor divider connected to the FB pin. Multiple Zener diodes in series may be needed for higher output power applications to keep the Zener’s temperature within the specification.

reached its turn-on voltage. voltage to very high voltage DC input voltage applications. for high voltage offline supply voltages.

3798 F05

Figure 5. Switching Waveforms When Output Figure 6. Switching Waveforms When Output Short-Circuits

3798 F06

3798 TA02

VIN_SENSE VIN DCM FB VREF CTRL2 CTRL1 GATE SENSE INTVCC GND L T3798 COMP+VC COMP– C10 560μF × 2 BR1: DIODES, INC. HD06 C8: VISHAY 440LD22-R D1: CENTRAL SEMICONDUCTOR CMR1U-06M D2, D3: DIODES INC. BAV20W D4: CENTRAL SEMICONDUCTOR CMR1U-02M M1: FAIRCHILD FDPF15N65 T1: COILCRAFT JA4429-AL Z1: FAIRCHILD SMBJ170A Z2: CENTRAL SEMICONDUCTOR CMZ5937B 4.7pFC5 10μ F 2.2nF 4:1:1 C7, 0.1μF R16 20Ω R17 20ΩD2 RS 0.05Ω 499k 499k 100k 100k R15 4.99k 22pF R14 90.9k CTRL3 EN/UVLO 40.2k R11 100k R10 16.5k C3 2.2μF OVP R12 221k 4.7μF R13 “Y1 CAP” 24V 95.3k 90V TO 265V AC 0.1μF 0.068μF BR1L1 33mH 800μH Universal Input 24W PFC Bus Converter

3798 TA03

VIN_SENSE VIN DCM FB VREF CTRL2 CTRL1 GATE SENSE INTVCC GND L T3798 COMP+VC COMP– C10 1000μF ×2 C11 10μ F BR1: DIODES, INC. HD06 C8: VISHAY 440LD22-R C11: MURATA GRM32ER7YA106KA12L D1: CENTRAL SEMICONDUCTOR CMR1U-06M D2, D3: DIODES INC. BAV20W D4: DIODES INC. SBR20A200CTB M1: INFINEON IPB60R165CP T1: WÜRTH ELEKTRONIK 750811248 Z1: FAIRCHILD SMBJ170A Z2: CENTRAL SEMICONDUCTOR CMZ5937B 22pF 10μ F 2.2nF 4:1:1 1μF C7, 0.1μF R16 20Ω R17 47ΩD2 RS 0.03Ω 499k 499k 100k 100k R15 5.49k 22pF R14 100k CTRL3 EN/UVLO OVP 40.2k R10 31.6k R12 221k R11 100k 2.4M 301k 4.7μF R13 33k 24V 90V TO 265V AC 0.22μF 0.1μF BR1L2 27mH 1mH “Y1 CAP” Universal Input 48W PFC Application

MSOP (MSE16) 0911 REV E 0.53 ± 0.152 (.021 ± .006) SEATING PLANE 0.18 (.007) 1.10 (.043) MAX 0.17 – 0.27 (.007 – .011) TYP 0.86 (.034) REF 0.50 (.0197) BSC 16151413121110 12345678 1 8 NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.254 (.010) 0° – 6° TYP DETAIL “A” DETAIL “A” GAUGE PLANE 5.23 (.206) MIN 3.20 – 3.45 (.126 – .136) 0.889 ± 0.127 (.035 ± .005) RECOMMENDED SOLDER PAD LAYOUT 0.305 ± 0.038 (.0120 ± .0015) TYP 0.50 (.0197) BSC BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ± 0.102 (.112 ± .004) 2.845 ± 0.102 (.112 ± .004) 4.039 ± 0.102 (.159 ± .004) (NOTE 3) 1.651 ± 0.102 (.065 ± .004) 1.651 ± 0.102 (.065 ± .004) 0.1016 ± 0.0508 (.004 ± .002) 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 0.280 ± 0.076 (.011 ± .003) REF 4.90 ± 0.152 (.193 ± .006) DETAIL “B” DETAIL “B” CORNER TAIL IS PART OF THE LEADFRAME FEATURE. FOR REFERENCE ONL Y NO MEASUREMENT PURPOSE

0.12 REF

0.35 REF 16-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1667 Rev E) Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However , no responsibility is assumed for its use. Linear Technology Corporation makes no representa- tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. Please refer to http://www.linear .com/designtools/packaging/ for the most recent package drawings.

Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear .com © LINEAR TECHNOLOGY CORPORATION 2012 LT 0212 • PRINTED IN USA RELATED PARTS TYPICAL APPLICATION PART NUMBER DESCRIPTION COMMENTS L T3799/L T3799-1 Offline Isolated Flyback LED Controller with Active PFC No Opto-Coupler Required, TRIAC Dimmable, VIN and VOUT Limited Only by External Components, MSOP-16E L T3748 100V Isolated Flyback Controller 5V ≤ V IN ≤ 100V , No Opto Flyback , MSOP-16 with High Voltage Spacing L T3573/L T3574/L T3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch L T3511/L T3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch L T3757/L T3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive L T3957/L T3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch L TC3803/L TC3803-3/L TC3803-5 200kHz/300kHz Flyback Controllers V IN and VOUT Limited Only by External Components L TC3805/L TC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited Only by External Components

3798 TA04

VIN_SENSE VIN DCM FB VREF CTRL2 CTRL1 GATE SENSE INTVCC GND L T3798 COMP+VC COMP– C10 10μF C2: TDK C5750X7S2A106M C8: VISHAY 440LD22-R C10: MURATA GRM32ER7YA106KA12L D1: DIODES INC. DFLS1150 D2, D3: DIODES INC. BAV20W D4: ON SEMICONDUCTOR MBR20200CT D5: DIODES INC. DFLS2100 15pF 10μ F 2.2nF 22pF 1:1:0.8 0.1μF C7, 22nF R17 20ΩD2 “Y1 CAP” RS 0.004Ω 24k TO INTVCC 6.8k 6.8k R15 5.9k R14 100k CTRL3 EN/UVLO OVP 40.2k R10 34.8k 16.2k R12 221k R11 100k 402k 51.1k 4.7μF R13 15k 28V 10μF VIN 20V TO 60V M1: FAIRCHILD FDP2532 T1: WÜRTH ELEKTRONIK 750312872 Z1: DIODES INC. SMCJ60A Z2: CENTRAL SEMICONDUCTOR CMZ59398 Z3: FAIRCHILD SMBJ170A 112W Wide DC Input Industrial Power Supply