MPZ2012S101A

TPA2013D1 TI

Document overview

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Technical content

FEATURES

APPLICATIONS

DESCRIPTION

IN– IN+ AGND PGND VOUT+ VOUT– V OUTCC V INCC

10 F/c109

22 F/c109

1 F/c109

To□Battery Differential Input CIN CIN V FBCC SDb SDd ShutDown□Boost ShutDown□ClassD GAINGain□(V /Float/GND)CC 2.2□to□6.2 H/c109 GPIO 50□kΩ 500□kΩ TPA2013D1 TPA2013D1 SLOS520 AUGUST 2007 www.ti.com 2.7-W CONSTANT OUTPUT POWER CLASS-D AUDIO AMPLIFIER WITH INTEGRATED BOOST CONVERTER Cell Phones High Efficiency Integrated Boost Converter (Over 90% Efficiency) PDA GPS 2.2-W into an Ω Load from a 3.6-V Supply Portable Electronics 2.7-W into an Ω Load from a 3.6-V Supply Operates from 1.8 V to 5.5 V Efficient Class-D Prolongs Battery Life The TPA2013D1 is a high efficiency Class-D audio Independent Shutdown for Boost Converter power amplifier with an integrated boost converter. It and Class-D Amplifier drives up to 2.7 W (10% THD+N) into a Ω speaker. Differential Inputs Reduce RF Common Noise With 85% typical efficiency, the TPA2013D1 helps extend battery life when playing audio. Built-in INPUT Low Pass Filter Decreases RF and Out of Band Noise Sensitivity The built-in boost converter generates the voltage rail Synchronized Boost and Class-D Eliminates for the Class-D amplifier. This provides a louder Beat Frequencies audio output than a stand-alone amplifier connected directly to the battery. It also maintains a consistent Thermal and Short-Circuit Protection loudness, regardless of battery voltage. Additionally, Available in 2.275 mm x 2.275 mm 16-ball the boost converter can be used to supply external WCSP and mm x mm 20-Lead QFN devices. Packages The TPA2013D1 has an integrated low pass filter to Selectable Gain Settings of V/V, V/V, and improve RF rejection and reduce out-of-band noise, V/V increasing the signal to noise ratio (SNR). A built-in PLL synchronizes the boost converter and Class-D switching frequencies, thus eliminating beat frequencies and improving audio quality. All outputs are fully protected against shorts to ground, power supply, and output-to-output shorts. Please be aware that an important notice concerning availability, standard warranty, and use in critical sheet. PRODUCTION DATA information is current as of publication date. Copyright 2007, Texas Instruments Incorporated Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

www.ti.com DEVICE INFORMATION VCCFB SW SW V CCOUT V CCIN SDb IN+IN– PGNDPGND SDd VDD PGND RGP (QFN)□Package (Top□View) TPA2013D1RGP 6 10 1620 GAIN AGND VOUT– VOUT– VOUT+ VOUT+ VOUT+ VCCFB SWVCCOUTVCCIN SDbIN+ IN–PGND AGNDSDdPGNDVOUT– VDDGAINVOUT+ PGND D1 D2 D3 D4 C1 C2 C3 C4 B1 B3 B4B2 A1 A2 A3 A4 YZH (WCSP)□Package (Top□View) TPA2013D1YZH TPA2013D1 SLOS520 AUGUST 2007 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. BOOST CONVERTER TERMINAL FUNCTIONS TERMINAL I/O I Positive audio input IN I Negative audio input VOUT+ 13, 14, O Positive audio output VOUT 11, O Negative audio output SDb I Shutdown terminal for the Boost Converter SDd I Shutdown terminal for the Class D Amplifier SW 18, Boost and rectifying switch input V CC OUT Boost converter output connect to V CC IN GAIN I Gain selection pin V C C IN Class-D audio power amplifier voltage supply connect to V CC OUT V CC FB I Voltage feedback V DD Supply voltage AGND Analog ground connect all GND pins together PGND 10, D1, C2, Power ground connect all GND pins together Thermal Solder the thermal pad on the bottom of the QFN package to the GND plane of the PCB. Die Pad N/A P Pad It is required for mechanical stability and will enhance thermal performance. Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com PWM and Level Shifter H-Bridge Internal Oscillator Biases, Control, and References Regulator Vmax Control Gate Control Anti- Ringing PGND AGND BG□Control PGNDAGND V OUTCC Vref AGND IN+ IN– SDb VDD SW V OUTCC V FBCC V INCC VOUT+ VOUT– PGND SDd Res. Array GAIN TPA2013D1 SLOS520 AUGUST 2007 Functional Block Diagram Table BOOST CONVERTER MODE CONDITION CASE OUTPUT CURRENT MODE OF OPERATION V DD V CC Low Continuous (fixed frequency) V DD V CC High Continuous (fixed frequency) V DD V CC Low Discontinuous (variable frequency) V DD V CC High Discontinuous (variable frequency) Table DEVICE CONFIGURATION Boost Class D SDb SDd Comments Converter Amplifier low low OFF OFF Device is in shutdown mode Iq μ A Boost converter is off. Class-D Audio Power Amplifier (APA) can be driven by an low high OFF ON external pass transistor connected to the battery. high low ON OFF Class-D APA is off. Boost Converter is on and can be used to drive an external device. Boost converter and Class-D APA are on. Normal operation. Boost converter can be high high ON ON used to drive an external device in parallel to the Class-D APA within the limits of the boost converter output current. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com ABSOLUTE MAXIMUM RATINGS DISSIPATION RATINGS AVAILABLE OPTIONS RECOMMENDED OPERATING CONDITIONS TPA2013D1 SLOS520 AUGUST 2007 over operating free-air temperature range (unless otherwise noted) (1) VALUE UNIT V DD Supply voltage 0.3 to V V I Input voltage, Vi: SDb SDd IN+, IN V CC FB 0.3 to V DD 0.3 V Continuous total power dissipation See Dissipation Rating Table T A Operating free-air temperature range to C T J Operating junction temperature range to 150 C T stg Storage temperature range to 150 C (1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operations of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. PACKAGE T A C DERATING FACTOR (1) T A C T A C ball WCSP 1.5 W 12.4 mW/ C W 0.8 W pin QFN 2.5 W 20.1 mW/ C 1.6 W 1.3 W (1) Derating factor measured with JEDEC High K board. T A PACKAGED DEVICES (1) PART NUMBER SYMBOL 16-ball, 2.275 mm 2.275 mm WCSP TPA2013D1YZH BTH 0.05mm tolerance) C TO C 20-pin, mm mm QFN TPA2013D1RGP (2) BTI (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com (2) The RGP package is only available taped and reeled. To order, add suffix R to the end of the part number for a reel of 3000 (e.g., TPA2013D1RGPR). MIN MAX UNIT V DD Supply voltage 1.8 5.5 V V IH High-level input voltage SDb SDd 1.3 V V IL Low-level input voltage SDb SDd 0.35 V I IH High-level input current SDb SDd 5.8 V DD 5.5 V CC 5.5 V μ A I IL Low-level input current SDb SDd -0.3 V DD 5.5 V CC 5.5 V μ A T A Operating free-air temperature C Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com DC CHARACTERISTICS TPA2013D1 SLOS520 AUGUST 2007 T A C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT Class-D audio power amplifier V CC 5.5 V voltage supply range, V CC IN SDd SDb V DD 1.8 R L Ω 0.04 1.5 SDd SDb V DD 3.6 R L Ω 0.04 1.5 SDd SDb V DD 4.5 R L Ω 0.02 1.5 I SD Shutdown quiescent current μ A SDd SDb 0.35 V DD 1.8 R L Ω 0.03 1.5 SDd SDb 0.35 V DD 3.6 R L Ω 0.03 1.5 SDd SDb 0.35 V DD 4.5 R L Ω 0.02 1.5 Boost converter quiescent SDd SDb 1.3 V DD 3.6 V CC 5.5 No 1.3 I DD mA current Load, No Filter V DD 3.6, V cc 5.5 No Load, No Filter 4.3 Class D amplifier quiescent I CC mA current V DD 4.5, V cc 5.5 No Load, No Filter 3.6 SDd SDb 1.3V, V DD 3.6, V cc 5.5 No Load, No 16.5 Boost converter and audio Filter I DD power amplifier quiescent mA SDd SDb 1.3V, V DD 4.5, V cc 5.5 No Load, No 18.5 current, Class D (1) Filter Boost converter switching 500 600 700 kHz frequency f Class D switching frequency 250 300 350 kHz UVLO Under voltage lockout 1.7 V Gain input low level Gain V/V (6dB) 0.35 V GAIN Gain input mid level Gain V/V (15.5 dB) (floating input) 0.7 0.8 V Gain input high level Gain V/V (20 dB) 1.35 V Class D Power on reset ON 2.8 V POR D threshold (1) I DD is calculated using I DD CC V CC )/(V DD η where I CC is the class D amplifier quiescent current; η 40%, which is the boost converter efficiency when class D amplifier has no load. To achieve the minimal 40% η it is recommended to use the suggested inductors in table and to follow the layout guidelines. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com BOOST CONVERTER DC CHARACTERISTICS CLASS D AMPLIFIER DC CHARACTERISTICS AC CHARACTERISTICS TPA2013D1 SLOS520 AUGUST 2007 T A C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V CC Output voltage range 3.0 5.5 V V FB Feedback voltage 490 500 510 mV I OL Output current limit, Boost_max 1300 1500 1700 mA R ON_PB PMOS switch resistance 220 m Ω R ON_NB NMOS resistance 170 m Ω No Load, 1.8 V V DD 5.2 Line regulation mV/V V CC 5.5 V V DD 3.6 I L 500 mA, Load regulation mV/A V CC 5.5 V I L Start up current limit, Boost 0.4 I Boost mA T A C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT V in 100 mV, V DD 1.8 V CC R L Ω 0.5 2.2 CMR Input common mode range V in 100 mV, V DD 2.5 V CC 3.6 R L Ω 0.5 2.8 V V in 100 mV, V DD 3.6 V CC 5.5 R L Ω 0.5 4.7 R L Ω V icm 0.5 and V icm V CC 0.8, differential CMRR Input common mode rejection dB inputs shorted V CC 3.6 Av V/V, IN+ IN V ref R L Ω V CC 3.6 Av V/V, IN+ IN V ref R L Ω Output offset voltage V OO mV Class-D V CC 3.6 Av V/V, IN+ IN V ref R L Ω V CC 5.5 Av V/V, IN+ IN V ref R L Ω Gain V/V dB) R in Input Impedance Gain V/V (15.5 dB) k Ω Gain V/V (20 dB) 9.5 OUTP High-side FET On-state 0.36 series resistance R DS(on) OUTP Low-side FET On-state 0.36 series resistance I OUTx 300 mA; V CC 3.6 V Ω OUTN High-side FET On-state 0.36 series resistance R DS(on) OUTN Low-side FET On-state 0.36 series resistance Low Gain GAIN 0.35 V 1.8 2.2 V/V A V Mid Gain GAIN 0.8 V 5.7 6.3 V/V High Gain GAIN 1.35 V 9.5 10.5 V/V T A V DD 3.6V, R L Ω L 4.7 μ H (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT t START Start up time 1.8 V V DD 5.5 C IN μ F 7.5 ms THD+N 1%, V CC 5.5 V DD 3.6 85% R L Ω Pout 1.7 C boost μ F η Efficiency THD+N 1%, V CC 5.5 V DD 4.2 87.5% R L Ω Pout 1.7 W Thermal Shutdown Threshold 150 C Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com CLASS D AMPLIFIER AC CHARACTERISTICS TPA2013D1 IN+ IN OUT+ OUT– VDD VDD GND CI CI Measurement Output Load 30□kHz Low-Pass Filter Measurement Input T A V DD 3.6V, R L Ω L 4.7 μ H (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT KSVR Output referred power supply V DD 3.6 V CC 5.5V, 200 mV PP ripple, f 217 dB Class-D rejection ratio Hz f kHz, P o 1.7 V CC 5.5 V f kHz, P o 1.2 V CC 4.5 V THD+N Total harmonic distortion noise Class-D f kHz, P o 2.2 V CC 5.5 V 10% f kHz, P o V CC 5.5 V 0.1% Vn Output integrated noise floor Av dB (2V/V) Class-D μ Vrms Output integrated noise floor Av dB (2V/V) A-weighted THD+N 10%, V CC 5.5V, V DD 3.6V R L Ω 2.2 THD+N 1%, V CC 5.5V, V DD 3.6V R L Ω 1.7 THD+N 1%, V CC 4.5V, V DD 3.6V R L Ω 1.2 P O Maximum output power W THD+N 10%, V CC 5.5V, V DD 3.6V R L Ω 2.7 THD+N 1%, V CC 5.5V, V DD 3.6V R L Ω 2.2 THD+N 1%, V CC 4.5V, V DD 3.6V R L Ω 1.9 TEST SET-UP FOR GRAPHS (1) C I was shorted for any common-mode input voltage measurement. All other measurements were taken with a μ F C I (unless otherwise noted). (2) A 33- μ H inductor was placed in series with the load resistor to emulate a small speaker for efficiency measurements. (3) The 30-kHz low-pass filter is required, even if the analyzer has an internal low-pass filter. An RC low-pass filter (100 Ω 47-nF) is used on each output for the data sheet graphs. (4) L 4.7 μ H is used for the boost converter unless otherwise noted. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com TYPICAL CHARACTERISTICS PO − Output Power − W 100 0.0 0.5 1.0 1.5 Efficiency − % G001 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 4.5 V VDD = 4.2 V VDD = 3.6 V VDD = 2.5 V VDD = 1.8 V PO − Output Power − W 100 Efficiency − % G002 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 5.5 V VDD = 4.2 V VDD = 3.6 V VDD = 2.5 V VDD = 1.8 V PO − Output Power − W 0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.0 0.5 1.0 1.5 PD − Power Dissipation − W G003 VDD = 3.6 V VDD = 1.8 V Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 4.5 V VDD = 2.5 V VDD = 4.2 V PO − Output Power − W 0.0 0.1 0.2 0.3 0.4 0.5 0.6 PD − Power Dissipation − W G004 VDD = 1.8 V Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 5.5 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V PO − Output Power − W 0.0 0.2 0.4 0.6 0.8 1.0 0.0 0.5 1.0 1.5 IDD − Supply Current − A G005 VDD = 1.8 V Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 4.5 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V PO − Output Power − W 0.0 0.2 0.4 0.6 0.8 1.0 1.2 IDD − Supply Current − A G006 VDD = 1.8 V Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 5.5 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V TPA2013D1 SLOS520 AUGUST 2007 EFFICIENCY EFFICIENCY vs vs OUTPUT POWER OUTPUT POWER Figure Figure POWER DISSIPATION POWER DISSIPATION vs vs OUTPUT POWER OUTPUT POWER Figure Figure SUPPLY CURRENT SUPPLY CURRENT vs vs OUTPUT POWER OUTPUT POWER Figure Figure Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com VDD − Supply Voltage − V 0.0 0.5 1.0 1.5 2.0 2.5 PO − Output Power − W G007 Gain = 2 V/V R L = 8 Ω + 33 µH THD = 1% V CC = 4.5 V L = 2.2 µH L = 3.3 µHL = 6.2 µH L = 4.7 µH VDD − Supply Voltage − V 0.0 0.5 1.0 1.5 2.0 2.5 PO − Output Power − W G008 L = 3.3 µH L = 6.2 µH L = 4.7 µH Gain = 2 V/V R L = 8 Ω + 33 µH THD = 1% V CC = 5.5 V VDD − Supply Voltage − V 0.0 0.5 1.0 1.5 2.0 2.5 PO − Output Power − W G009 Gain = 2 V/V R L = 8 Ω + 33 µH THD = 10% V CC = 4.5 V L = 2.2 µH L = 3.3 µHL = 6.2 µH L = 4.7 µH VDD − Supply Voltage − V 0.0 0.5 1.0 1.5 2.0 2.5 PO − Output Power − W G010 L = 6.2 µH L = 4.7 µH L = 3.3 µH Gain = 2 V/V R L = 8 Ω + 33 µH THD = 10% V CC = 5.5 V R L − Load Resistance − Ω 0.0 0.5 1.0 1.5 2.0 2.5 3.0 4 8 12 16 20 24 28 32 PO − Output Power − W G011 f = 1 kHz Gain = 2 V/V V CC = 4.5 V THD = 1% THD = 10% R L − Load Resistance − Ω 0.0 0.5 1.0 1.5 2.0 2.5 3.0 4 8 12 16 20 24 28 32 PO − Output Power − W G012 f = 1 kHz Gain = 2 V/V V CC = 5.5 V THD = 1% THD = 10% TPA2013D1 SLOS520 AUGUST 2007 TYPICAL CHARACTERISTICS (continued) OUTPUT POWER OUTPUT POWER vs vs SUPPLY VOLTAGE SUPPLY VOLTAGE Figure Figure OUTPUT POWER OUTPUT POWER vs vs SUPPLY VOLTAGE SUPPLY VOLTAGE Figure Figure 10. OUTPUT POWER OUTPUT POWER vs vs LOAD LOAD Figure 11. Figure 12. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com PO − Output Power − W 0.01 0.1 1 3 100 0.1 0.01 G013 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 4.5 V10 VDD = 1.8 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V THD+N − % 0.01 0.1 1 3 PO − Output Power − W 0.1 0.01 G014 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 5.5 V10 VDD = 1.8 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V THD+N − % 100 PO − Output Power − W 0.01 0.1 1 5 0.1 0.01 G015 Gain = 2 V/V R L = 4 Ω + 33 µH VCC = 5.5 V10 VDD = 1.8 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V THD+N − % 100 f − Frequency − Hz 20 100 1k 20k 0.1 0.01 0.001 G016 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 4.5 V VDD = 1.8 V1 PO = 0.075 W PO = 0.2 W 10k PO = 0.025 W THD+N − % f − Frequency − Hz 0.1 0.01 0.001 G017 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 4.5 V VDD = 3.6 V1 PO = 0.05 W PO = 1 W 20 100 1k 20k 10k PO = 0.25 W THD+N − % 20 100 1k 20k 10k f − Frequency − Hz 0.1 0.01 0.001 G018 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 5.5 V VDD = 1.8 V1 PO = 0.075 W PO = 0.025 W PO = 0.2 W THD+N − % TPA2013D1 SLOS520 AUGUST 2007 TYPICAL CHARACTERISTICS (continued) TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs OUTPUT POWER OUTPUT POWER Figure 13. Figure 14. TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs OUTPUT POWER FREQUENCY Figure 15. Figure 16. TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY FREQUENCY Figure 17. Figure 18. Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com 20 100 1k 20k 10k f − Frequency − Hz 0.1 0.01 0.001 G019 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 5.5 V VDD = 3.6 V1 PO = 0.05 W PO = 1 W PO = 0.25 W THD+N − % 20 100 1k 20k 10k f − Frequency − Hz 0.1 0.01 0.001 G020 Gain = 2 V/V P O = 250 mW R L = 4 Ω + 33 µH VCC = 4.5 V1 VDD = 1.8 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V THD+N − % 20 100 1k 20k 10k f − Frequency − Hz 0.1 0.01 0.001 G021 Gain = 2 V/V P O = 250 mW R L = 4 Ω + 33 µH VCC = 5.5 V1 VDD = 1.8 V VDD = 2.5 V VDD = 4.2 V VDD = 3.6 V THD+N − % −120 −100 −80 −60 −40 −20 f − Frequency − Hz PSRR − dB G022 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 4.5 V 20 100 1k 20k 10k VDD = 3.6 V VDD = 4.2 V VDD = 2.5 V VDD = 1.8 V −120 −100 −80 −60 −40 −20 f − Frequency − Hz PSRR − dB G023 Gain = 2 V/V R L = 8 Ω + 33 µH VCC = 5.5 V 20 100 1k 20k 10k VDD = 3.6 V VDD = 2.5 V VDD = 1.8 V VDD = 4.2 V −120 −100 −80 −60 −40 −20 f − Frequency − Hz CMRR − dB G024 Gain = 2 V/V R L = 8 Ω VCC = 4.5 V 20 100 1k 20k 10k VDD = 3.6 V VDD = 2.5 V VDD = 1.8 V VDD = 4.2 V TPA2013D1 SLOS520 AUGUST 2007 TYPICAL CHARACTERISTICS (continued) TOTAL HARMONIC DISTORTION NOISE TOTAL HARMONIC DISTORTION NOISE vs vs FREQUENCY FREQUENCY Figure 19. Figure 20. TOTAL HARMONIC DISTORTION NOISE POWER SUPPLY REJECTION RATIO vs vs FREQUENCY FREQUENCY Figure 21. Figure 22. POWER SUPPLY REJECTION RATIO COMMON-MODE REJECTION RATIO vs vs FREQUENCY FREQUENCY Figure 23. Figure 24. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com −90 −80 −70 −60 −50 −40 −30 −20 −10 f − Frequency − Hz CMRR − dB G025 Gain = 2 V/V R L = 8 Ω VCC = 5.5 V 20 100 1k 20k 10k VDD = 3.6 V VDD = 1.8 V VDD = 2.5 V VDD = 4.2 V 100 IO − Output Current − A 0.01 0.1 1 VCC = 4.5 V VDD = 1.8 V VDD = 2.5 V VDD = 3.6 V VDD = 4.2 V Boost Efficiency − % G026 100 IO − Output Current − A 0.01 0.1 1 VCC = 5.5 V VDD = 1.8 V VDD = 2.5 V VDD = 3.6 V VDD = 4.2 V Boost Efficiency − % G027 VDD − Supply Voltage − V 100 ICC = 250 mA L = 4.7 µH Boost Efficiency − % VCC = 4.5 V VCC = 5.5 V G028 VDD − Supply Voltage (Boost) − V 0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 L = 4.7 µH IOM − Max. Continuous Output Current − A VCC = 4.5 V VCC = 5.5 V G029 t − Time − ms 0 2 4 6 8 10 12 14 16 18 20 V − Voltage − V VCC OUT SDb , SDd Start Time 7.5 ms G030 TPA2013D1 SLOS520 AUGUST 2007 TYPICAL CHARACTERISTICS (continued) COMMON-MODE REJECTION RATIO BOOST EFFICIENCY vs vs FREQUENCY OUTPUT CURRENT Figure 25. Figure 26. BOOST EFFICIENCY BOOST EFFICIENCY vs vs OUTPUT CURRENT SUPPLY VOLTAGE Figure 27. Figure 28. MAXIMUM CONTINUOUS OUTPUT CURRENT vs SUPPLY VOLTAGE (BOOST) Start-Up Time Figure 29. Figure 30. Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com APPLICATION INFORMATION FULLY DIFFERENTIAL AMPLIFIER Advantages of Fully Differential Amplifiers BOOST CONVERTER Boost Terms TPA2013D1 SLOS520 AUGUST 2007 The TPA2013D1 is a fully differential amplifier with differential inputs and outputs. The fully differential amplifier consists of a differential amplifier with common-mode feedback. The differential amplifier ensures that the amplifier outputs a differential voltage on the output that is equal to the differential input times the gain. The common-mode feedback ensures that the common-mode voltage at the output is biased around V CC regardless of the common-mode voltage at the input. The fully differential TPA2013D1 can still be used with a single-ended input; however, the TPA2013D1 should be used with differential inputs when in a noisy environment, like a wireless handset, to ensure maximum noise rejection. Input-coupling capacitors not required: The fully differential amplifier allows the inputs to be biased at voltage other than mid-supply. The inputs of the TPA2013D1 can be biased anywhere within the common mode input voltage range listed in the Recommended Operating Conditions table. If the inputs are biased outside of that range, input-coupling capacitors are required. Midsupply bypass capacitor, C (BYPASS) not required: The fully differential amplifier does not require a bypass capacitor. Any shift in the midsupply affects both positive and negative channels equally and cancels at the differential output. Better RF-immunity: GSM handsets save power by turning on and shutting off the RF transmitter at a rate of 217 Hz. The transmitted signal is picked-up on input and output traces. The fully differential amplifier cancels the signal better than the typical audio amplifier. The TPA2013D1 consists of a boost converter and a Class-D amplifier. The boost converter takes a low supply voltage, V DD and increases it to a higher output voltage, V CC V CC is the power supply for the Class-D amplifier. The two main passive components necessary for the boost converter are the boost inductor and the boost capacitor. The boost inductor stores current, and the boost capacitor stores charge. As the Class-D amplifier depletes the charge in the boost capacitor, the boost inductor charges it back up with the stored current. The cycle of charge/discharge occurs at a frequency of f boost The TPA2013D1 allows a range of V CC voltages, including setting V CC lower than V DD The following is a list of terms and definitions used in the boost equations found later in this document. C Minimum boost capacitance required for a given ripple voltage on V CC L Boost inductor f boost Switching frequency of the boost converter. I CC Current pulled by the Class-D amplifier from the boost converter. I L Average current through the boost inductor. and Resistors used to set the boost voltage. V CC Boost voltage. Generated by the boost converter. Voltage supply for the Class-D amplifier. V DD Supply voltage to the IC. Δ I L Ripple current through the inductor. Δ V Ripple voltage on V CC due to capacitance. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com SETTING THE BOOST VOLTAGE CC /c180/c230 /c246 /c231 /c247 /c232 /c248 (1) INDUCTOR SELECTION CCL CC DD VI = I V 0.8 /c230 /c246/c180/c231 /c247 /c180/c232 /c248 (2) DD CC DD L CC V (V V )L = I f V boost /c180 /c45 /c68 /c180 /c180 (3) TPA2013D1 SLOS520 AUGUST 2007 Use Equation to determine the value of for a given V CC The maximum recommended value for V CC is 5.5 The typical value of the V CC FB pin is 500 mV. The current through the resistor divider should be about 100 times greater than the current into the V CC FB pin, typically 0.01 μ Based on those two values, the recommended value of is 500 k Ω V CC must be greater than V and less than or equal to 5.5 SURFACE MOUNT INDUCTORS Working inductance decreases as inductor current increases. If the drop in working inductance is severe enough, it may cause the boost converter to become unstable, or cause the TPA2013D1 to reach its current limit at a lower output power than expected. Inductor vendors specify currents at which inductor values decrease by a specific percentage. This can vary by 10% to 35%. Inductance is also affected by dc current and temperature. TPA2013D1 INDUCTOR EQUATIONS Inductor current rating is determined by the requirements of the load. The inductance is determined by two factors: the minimum value required for stability and the maximum ripple current permitted in the application. Use Equation to determine the required current rating. Equation shows the approximate relationship between the average inductor current, I L to the load current, load voltage, and input voltage CC V CC and V DD respectively). Insert I CC V CC and V DD into Equation to solve for I L The inductor must maintain at least 90% of its initial inductance value at this current. The minimum working inductance is 2.2 μ A lower value may cause instability. Ripple current, Δ I L is peak-to-peak variation in inductor current. Smaller ripple current reduces core losses in the inductor as well as the potential for EMI. Use Equation to determine the value of the inductor, Equation shows the relationship between inductance V DD V CC the switching frequency, f boost and Δ I L Insert the maximum acceptable ripple current into Equation to solve for Δ I L is inversely proportional to Minimize Δ I L as much as is necessary for a specific application. Increase the inductance to reduce the ripple current. Note that making the inductance too large will prevent the boost converter from responding to fast load changes properly. Typical inductor values for the TPA2013D1 are 4.7 μ H to 6.8 μ Select an inductor with a small dc resistance, DCR. DCR reduces the output power due to the voltage drop across the inductor. Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com CAPACITOR SELECTION TPA2013D1 SLOS520 AUGUST 2007 SURFACE MOUNT CAPACITORS Temperature and applied dc voltage influence the actual capacitance of high-K materials. Table shows the relationship between the different types of high-K materials and their associated tolerances, temperature coefficients, and temperature ranges. Notice that a capacitor made with X5R material can lose up to 15% of its capacitance within its working temperature range. High-K material is very sensitive to applied dc voltage. X5R capacitors can have losses ranging from to 45% of their initial capacitance with only half of their dc rated voltage applied. For example, if Vdc is applied to a μ F X5R capacitor, the measured capacitance at that point may show 0.85 μ 0.55 μ or somewhere in between. Y5V capacitors have losses that can reach or exceed 50% to 75% of their rated value. In an application, the working capacitance of components made with high-K materials is generally much lower than nominal capacitance. A worst case result with a typical X5R material might be 10% tolerance, 15% temperature effect, and 45% dc voltage effect at 50% of the rated voltage. This particular case would result in a working capacitance of 42% (0.9 0.85 0.55) of the nominal value. Select high-K ceramic capacitors according to the following rules: Use capacitors made of materials with temperature coefficients of X5R, X7R, or better. Use capacitors with dc voltage ratings of at least twice the application voltage. Use minimum V capacitors for the TPA2013D1. Choose a capacitance value at least twice the nominal value calculated for the application. Multiply the nominal value by a factor of for safety. If a μ F capacitor is required, use μ The preceding rules and recommendations apply to capacitors used in connection with the TPA2013D1. The TPA2013D1 cannot meet its performance specifications if the rules and recommendations are not followed. Table Typical tolerance and temperature coefficient of capacitance by material Material COG/NPO X7R X5R Typical Tolerance 10% 80/ 20% Temperature Coefficient 30ppm 15% 22/ 82% Temperature Range, C C C -30/85 C Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com /c40 /c41CC CC DD boost CC I V VC = 2 V f V /c180 /c45/c180 /c68 /c180 /c180 (4) /c40 /c41CC CC DD boost CC I V VC = V f V /c180 /c45 /c68 /c180 /c180 (5) TPA2013D1 SLOS520 AUGUST 2007 TPA2013D1 CAPACITOR EQUATIONS The value of the boost capacitor is determined by the minimum value of working capacitance required for stability and the maximum voltage ripple allowed on V CC in the application. The minimum value of working capacitance is μ Do not use any component with a working capacitance less than μ For X5R or X7R ceramic capacitors, Equation shows the relationship between the boost capacitance, to load current, load voltage, ripple voltage, input voltage, and switching frequency CC V CC Δ V DD f boost respectively). Insert the maximum allowed ripple voltage into Equation to solve for A factor of is included to implement the rules and specifications listed earlier. For aluminum or tantalum capacitors, Equation shows the relationship between he boost capacitance, to load current, load voltage, ripple voltage, input voltage, and switching frequency CC V CC Δ V DD f boost respectively). Insert the maximum allowed ripple voltage into Equation to solve for Solve this equation assuming ESR is zero. Capacitance of aluminum and tantalum capacitors is normally not sensitive to applied voltage so there is no factor of included in Equation However, the ESR in aluminum and tantalum capacitors can be significant. Choose an aluminum or tantalum capacitor with ESR around m Ω For best perfornamce using of tantalum capacitor, use at least a V rating. Note that tantalum capacitors must generally be used at voltages of half their ratings or less. Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com RECOMMENDED INDUCTOR AND CAPACITOR VALUES BY APPLICATION CLASS-D REQUIREMENTS DECOUPLING CAPACITORS INPUT CAPACITORS )I I 1f =c (2 R C/c180 /c112 /c180 (6) TPA2013D1 SLOS520 AUGUST 2007 Use Table as a guide for determining the proper inductor and capacitor values. Table Recommended Values Class-D Class-D Minimum Required Max Output Max I L L Inductor Vendor C (2) Capacitor Vendor Load V DD V CC Δ V Power (A) μ Part Numbers μ Part Numbers Ω (V) (V) (mVpp) (W) (1) 3.3 Toko DE2812C Kemet C1206C106K8PACTU 4.3 0.70 Coilcraft DO3314 Murata GRM32ER61A106KA01B Murata LQH3NPN3R3NG0 Taiyo Yuden LMK316BJ106ML-T 4.7 Murata LQH32PN4R7NN0 1.6 5.5 1.13 Murata GRM32ER71A226KE20L Toko DE4514C Taiyo Yuden LMK316BJ226ML-T Coilcraft LPS4018-472 3.3 4.6 1.53 Murata LQH55PN3R3NR0 TDK C4532X5R1A336M Toko DE4514C 6.2 2.3 1.8 5.5 Sumida Murata GRM32ER61A476KE20L CDRH5D28NP-6R2NC Taiyo Yuden LMK325BJ476MM-T (1) All power levels are calculated at THD unless otherwise noted (2) All values listed are for ceramic capacitors. The correction factor of is included in the values. The TPA2013D1 is a high-performance Class-D audio amplifier that requires adequate power supply decoupling to ensure the efficiency is high and total harmonic distortion (THD) is low. Place a low equivalent-series-resistance (ESR) ceramic capacitor, typically μ F as close as possible to the device VDD lead. This choice of capacitor and placement helps with higher frequency transients, spikes, or digital hash on the line. Additionally, placing this decoupling capacitor close to the TPA2013D1 is important for the efficiency of the Class-D amplifier, because any resistance or inductance in the trace between the device and the capacitor can cause a loss in efficiency. Place a capacitor of μ F or greater between the power supply and the boost inductor. The capacitor filters out high frequency noise. More importantly, it acts as a charge reservoir, providing energy more quickly than the board supply, thus helping to prevent any droop. The TPA2013D1 does not require input coupling capacitors if the design uses a differential source that is biased within the common mode input range. Use input coupling capacitors if the input signal is not biased within the recommended common-mode input range, if high pass filtering is needed, or if using a single-ended source. The input capacitors and input resistors form a high-pass filter with the corner frequency, fc, determined in Equation The value of the input capacitor is important to consider as it directly affects the bass (low frequency) performance of the circuit. Speakers in wireless phones cannot usually respond well to low frequencies, so the corner frequency can be set to block low frequencies in this application. Not using input capacitors can increase output offset. Use Equation to find the required the input coupling capacitance. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com I )I 1C = (2 f R c/c180 /c112 /c180 /c180 (7) FILTER FREE OPERATION AND FERRITE BEAD FILTERS 1□nF Ferrite Chip□Bead OUTP OUTN 1□nF Ferrite Chip□Bead OPERATION WITH DACs AND CODECs STEREO OPERATION APPLICATION TPA2013D1 SLOS520 AUGUST 2007 Any mismatch in capacitance between the two inputs inputs will cause a mismatch in the corner frequencies. Choose capacitors with a tolerance of 10% or better. A ferrite bead filter can often be used if the design is failing radiated emissions without an LC filter and the frequency sensitive circuit is greater than MHz. This filter functions well for circuits that just have to pass FCC and CE because FCC and CE only test radiated emissions greater than MHz. When choosing a ferrite bead, choose one with high impedance at high frequencies, and very low impedance at low frequencies. In addition, select a ferrite bead with adequate current rating to prevent distortion of the output signal. Use an LC output filter if there are low frequency MHz) EMI sensitive circuits and/or there are long leads from amplifier to speaker. Figure shows a typical ferrite bead output filters. Figure 31. Typical Ferrite Chip Bead Filter Suggested Chip Ferrite Bead Load Vendor Part Number Size Ω Murata BLM18EG121SN1 0603 Ω TDK MPZ2012S101A 0805 When using switching amplifiers with CODECs and DACs, sometimes there is an increase in the output noise floor from the audio amplifier. This occurs when mixing of the output frequencies of the CODEC/DAC with the switching frequencies of the audio amplifier input stage. The noise increase can be solved by placing a low-pass filter between the CODEC/DAC and audio amplifier. This filters off the high frequencies that cause the problem and allow proper performance. The TPA2013D1 has a two pole low pass filter at the inputs. The cutoff frequency of the filter is set to approximately 100kHz. The integrated low pass filter of the TPA2013D1 eliminates the need for additional external filtering components. A properly designed additional low pass filter may be added without altering the performance of the device. Use the boost converter of the TPA2013D1 to supply the power for another audio amplifier when stereo operation is required. Ensure the gains of the amplifiers match each other. This prevents one channel from sounding louder than the other. Use Equation through Equation to determine R1, R2, boost inductor, and the boost capacitor values. Figure is an example schematic. The TPA2032D1 is a good choice for this application; the gain is internally set to V/V, the power supply is compatible with V CC OUT of the TPA2013D1, and the output power of the TPA2032D1 is on par with the TPA2013D1. Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com TPA2032D1 VDD IN– IN+ GND VO+ SHUTDOWN CI CI VO– VDD SW IN– IN+ AGND PGND VOUT+ VOUT– V OUTCC V INCC

47 F/c109

To□Battery Left Channel Input Right Channel Input CIN CIN V FBCC SDb SDd ShutDown□Boost ShutDown□ClassD GAIN

4.7 H/c109

62.5□kΩ 500□kΩ TPA2013D1 LED DRIVER FOR DIGITAL STILL CAMERAS GPIO LXCL-PWF3 NDS355N 1 /c87 CKG57NX5R1C107M VDD SW IN– IN+ AGND PGND VOUT+ VOUT– V OUTCC V INCC

100 F/c109

To□Battery Differential Input CIN CIN V FBCC SDb SDd ShutDown□Boost ShutDown□ClassD GAINGain

6.2 H/c109

50□kΩ 500□kΩ TPA2013D1 TPA2013D1 SLOS520 AUGUST 2007 Figure 32. TPA2013D1 in Stereo With the TPA2032D1 Use the boost converter of the TPA2013D1 as a power supply for the flash LED of a digital still camera. Use a microprocessor or other device or synchronize the flash to shutter sound that typically comes from the speaker of a digital still camera. Figure shows a typical circuit for this application. LEDs, switches, and other components will vary by application. Figure 33. LED Driver Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com BYPASSING THE BOOST CONVERTER Toko 1098AS-4R7M Toshiba□CRS□06 Schottky□Diode VDD SW IN– IN+ AGND PGND VOUT+ VOUT– V OUTCC V INCC To□Battery Left Channel Input CIN CIN V FBCC SDb SDd GAIN GND□=□Bypass V =□Boost□ModeDD 50□kΩ 500□kΩ TPA2013D1 GPIO EFFICIENCY AND THERMAL INFORMATION JA 1 1= = = 80.64 C/WDerating Factor 0.0124/c113 /c176 (8) TPA2013D1 SLOS520 AUGUST 2007 Bypass the boost converter to drive the Class-D amplifier directly from the battery. Place a Shottky diode between the SW pin and the V CC IN pin. Select a diode that has an average forward current rating of at least 1A, reverse breakdown voltage of V or greater, and a forward voltage as small as possible. See Figure for an example of a circuit designed to bypass the boost converter. Do not configure the circuit to bypass the boost converter if V DD is higher than V CC when the boost converter is enabled SDb 1.3 V); V DD must be lower than V CC for proper operation. V DD may be set to any voltage within the recommended operating range when the boost converter is disabled SDb 0.3V). Place a logic high on SDb to place the TPA2013D1 in boost mode. Place a logic low on SDb to place the TPA2013D1 in bypass mode. Figure 34. Bypass Circuit The maximum ambient temperature depends on the heat-sinking ability of the PCB system. The derating factors for the YZH and RGP packages are shown in the dissipation rating table. Apply the same principles to both packages. Using the YZH package, and converting this to θ JA Given θ JA of 80.64 C/W, the maximum allowable junction temperature of 150 and the maximum internal dissipation of 0.317 W DD 3.6 P O 1.7 W), the maximum ambient temperature is calculated with the following equation: Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

www.ti.com A J JA DmaxT Max = T Max P = 150 80.64 (0.317) = 124 C/c45 /c113 /c45 /c176 (9) BOARD LAYOUT Copper Trace□Width Solder Pad□Width Solder□Mask Opening Copper Trace ThicknessSolder□Mask Thickness TPA2013D1 SLOS520 AUGUST 2007 Equation shows that the calculated maximum ambient temperature is 124 C at maximum power dissipation under the above conditions. The TPA2013D1 is designed with thermal protection that turns the device off when the junction temperature surpasses 150 C to prevent damage to the IC. Also, using speakers more resistive than Ω dramatically increases the thermal performance by reducing the output current and increasing the efficiency of the amplifier. In making the pad size for the WCSP balls, use nonsolder mask defined (NSMD) land. With this method, the solder mask opening is made larger than the desired land area, and the opening size is defined by the copper pad width. Figure and Table show the appropriate diameters for a WCSP layout. Figure 35. Land Pattern Dimensions Table Land Pattern Dimensions SOLDER PAD SOLDER MASK COPPER STENCIL STENCIL COPPER PAD DEFINITIONS OPENING THICKNESS OPENING THICKNESS Nonsolder mask 275 μ m 375 μ m oz max (32 μ 275 μ m x 275 μ m Sq. 125 μ m thick defined (NSMD) (+0.0, μ (+0.0, μ (rounded corners) NOTES: Circuit traces from NSMD defined PWB lands should be μ m to 100 μ m wide in the exposed area inside the solder mask opening. Wider trace widths reduce device stand off and impact reliability. Recommend solder paste is Type or Type Best reliability results are achieved when the PWB laminate glass transition temperature is above the operating the range of the intended application. For a PWB using a Ni/Au surface finish, the gold thickness should be less 0.5 mm to avoid a reduction in thermal fatigue performance. Solder mask thickness should be less than μ m on top of the copper circuit pattern. Best solder stencil performance is achieved using laser cut stencils with electro polishing. Use of chemically etched stencils results in inferior solder paste volume control. Trace routing away from WCSP device should be balanced in X and Y directions to avoid unintentional component movement due to solder wetting forces. Copyright 2007, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Link(s): TPA2013D1

www.ti.com Trace Width TPA2013D1 SLOS520 AUGUST 2007 Recommended trace width at the solder balls is μ m to 100 μ m to prevent solder wicking onto wider PCB traces. For high current pins (SW, PGND, VOUT+, VOUT V CC IN, and V CC OUT) of the TPA2013D1, use 100 μ m trace widths at the solder balls and at least 500 μ m PCB traces to ensure proper performance and output power for the device. For low current pins (IN IN+, SDd SDb GAIN, V CC FB, V DD of the TPA2013D1, use μ m to 100 μ m trace widths at the solder balls. Run IN- and IN+ traces side-by-side to maximize common-mode noise cancellation. Submit Documentation Feedback Copyright 2007, Texas Instruments Incorporated Product Folder Link(s): TPA2013D1

Orderable Device Status(1) Package Type Package Drawing Pins Package Qty Eco Plan(2) Lead/Ball FinishMSL Peak Temp (3) TPA2013D1RGPT PREVIEW QFN RGP 20 250 TBD Call TI Call TI TPA2013D1YZHR ACTIVE DSBGA YZH 16 3000 Green (RoHS & no Sb/Br) SNAGCU Level-1-260C-UNLIM TPA2013D1YZHT PREVIEW DSBGA YZH 16 250 TBD Call TI Call TI (1)The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2)Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontentfor the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS):TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt):This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br):TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. PACKAGE OPTION ADDENDUM www.ti.com 10-Sep-2007 Addendum-Page 1

X: Max = Y: Max = 2330 µm, Min = 2330 µm, Min = 2230 µm 2230 µm

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