LT8302

LT8302_15 LINER

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Technical content

8302faFor more information www.linear .com/L T8302 Typical applicaTion FeaTures DescripTion 42VIN Micropower No-Opto Isolated Flyback Converter with 65V/3.6A Switch The LT®8302 is a monolithic micropower isolated flyback converter. By sampling the isolated output voltage directly from the primary-side flyback waveform, the part requires no third winding or opto-isolator for regulation. The output voltage is programmed with two external resistors and a third optional temperature compensation resistor. Bound- ary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 3.6A, 65V DMOS power switch is integrated along with all the high voltage circuitry and control logic into a thermally enhanced 8-lead SO package. The LT8302 operates from an input voltage range of 2.8V to 42V and delivers up to 18W of isolated output power. The high level of integration and the use of boundary and low ripple burst modes result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. 2.8V to 32VIN/5VOUT Isolated Flyback Converter

applicaTions

n 2.8V to 42V Input Voltage Range n 3.6A, 65V Internal DMOS Power Switch n Low Quiescent Current: 106µA in Sleep Mode 380µ A in Active Mode n Quasi-Resonant Boundary Mode Operation at Heav y Load n Low Ripple Burst Mode® Operation at Light Load n Minimum Load < 0.5% (Typ) of Full Output n No T ransformer Third Winding or Opto-Isolator Required for Output V oltage Regulation n Accurate EN/UVLO Threshold and Hysteresis n Internal Compensation and Soft-Start n Temperature Compensation for Output Diode n Output Short-Circuit Protection n Thermally Enhanced 8-Lead SO Package n Isolated Automotive, Industrial, Medical Power Supplies n Isolated Auxiliary/Housekeeping Power Supplies L, LT, LT C, LT M, Linear Technology, the Linear logo and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, 7471522. Efficiency vs Load Current VIN L T8302 SW 9µH VIN 2.8V TO 32V 3:1 1µH RFB RREF EN/UVLO 470pF 10µF 1µF 220µF 10mA TO 1.1A (VIN = 5V) 10mA TO 2.0A (VIN = 12V) 10mA TO 2.9A (VIN = 24V) VOUT– 39/uni03A9 154k 115k 10k

8302 TA01a

LOAD CURRENT (A) EFFICIENCY (%) 1.5 2.5

8302 TA01b

0.5 1.0 2.0 3.0 VIN = 5V VIN = 12V VIN = 24V FRONT PAGE APPLICATION

8302fa For more information www.linear .com/L T8302 pin conFiguraTionabsoluTe MaxiMuM raTings EN/UV Op erating Junction Temperature Range (Notes 3, 4) LT830 LT830 0°C to 150°C LT830 55°C to 150°C 65°C to 150°C 0°C (Note 1) TOP VIEW TC RREF RFB SW EN/UVLO INTVCC VIN GND S8E PACKAGE 8-LEAD PLASTIC SO GND θJA = 33°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB orDer inForMaTion LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LT8302ES8E#PBF LT8302ES8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C LT8302IS8E#PBF LT8302IS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C LT8302HS8E#PBF LT8302HS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 150°C LT8302MPS8E#PBF LT8302MPS8E#TRPBF 8302 8-Lead Plastic SO –55°C to 150°C Consult LT C Marketing for parts specified with wider operating temperature ranges. Consult LT C Marketing for information on nonstandard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

8302faFor more information www.linear .com/L T8302 elecTrical characTerisTics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VEN/UVLO = VIN, CINTVCC = 1µF to GND, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT VIN VIN Voltage Range l 2.8 42 V IQ VIN Quiescent Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 0.5 106 380 2 µA µA µA µA EN/UVLO Shutdown Threshold For Lowest Off I Q l 0.3 0.75 V EN/UVLO Enable Threshold Falling l 1.178 1.214 1.250 V EN/UVLO Enable Hysteresis 14 mV IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.3 –0.1 2.5 0.1 2.7 0.1 µA µA µA V INTVCC INTVCC Regulation Voltage IINTVCC = 0mA to 10mA 2.85 3 3.1 V IINTVCC INTVCC Current Limit VINTVCC = 2.8V 10 13 16 mA INTVCC UVLO Threshold Falling 2.39 2.47 2.55 V INTVCC UVLO Hysteresis 105 mV (RFB – VIN) Voltage IRFB = 75µA to 125µA –50 50 mV RREF Regulation Voltage l 0.98 1.00 1.02 V RREF Regulation Voltage Line Regulation 2.8V ≤ V IN ≤ 42V –0.01 0 0.01 %/V VTC TC Pin Voltage 1.00 V ITC TC Pin Current VTC = 1.2V VTC = 0.8V 12 15 –200 18 µA µA fMIN Minimum Switching Frequency 11.3 12 12.7 kHz tON(MIN) Minimum Switch-On Time 160 ns tOFF(MAX) Maximum Switch-Off Time Backup Timer 170 µs ISW(MAX) Maximum Switch Current Limit 3.6 4.5 5.4 A ISW(MIN) Minimum Switch Current Limit 0.78 0.87 0.96 A RDS(ON) Switch On-Resistance ISW = 1.5A 80 mΩ ILKG Switch Leakage Current VSW = 65V 0.1 0.5 µA tSS Soft-Start Timer 11 ms Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 65V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 65V as shown in Figure 5. Note 3: The LT8302E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT8302I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8302H is guaranteed over the full –40°C to 150°C operating junction temperature range. The LT8302MP is guaranteed over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125°C. Note 4: The LT8302 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.

8302fa For more information www.linear .com/L T8302 Typical perForMance characTerisTics Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms VIN Shutdown Current VIN Quiescent Current, Sleep Mode VIN Quiescent Current, Active Mode Output Load and Line Regulation Output Temperature Variation Switching Frequency vs Load Current TA = 25°C, unless otherwise noted. TEMPERATURE (°C) –50 OUTPUT VOLTAGE (V) 5.1 5.2 5.3 25 75

8302 G02

5.0 4.9 –25 0 50 100 150 125 4.8 4.7 FRONT PAGE APPLICATION VIN = 12V IOUT = 1A RTC = 115k RTC = OPEN VSW 20V/DIV VOUT 50mV/DIV 2µs/DIV FRONT PAGE APPLICATION V IN = 12V IOUT = 2A

8302 G04

2µs/DIV FRONT PAGE APPLICATION V IN = 12V IOUT = 0.5A

8302 G05

20µs/DIV FRONT PAGE APPLICATION V IN = 12V IOUT = 10mA

8302 G06

VIN (V) IQ (µA)

8302 G07

TJ = 150°C TJ = 25°C TJ = –55°C VIN (V) IQ (µA) 100 110 120 130 140 10 20 30 40

8302 G08

TJ = 150°C TJ = –55°C TJ = 25°C VIN (V) IQ (µA) 380 400 420

8302 G09

TJ = 150°C TJ = –55°C TJ = 25°C LOAD CURRENT (A) 0 0.5 FREQUENCY (kHz) 200 500 1.0 2.0 2.5

8302 G03

1.5 3.0 VIN = 5V VIN = 12V VIN = 24V FRONT PAGE APPLICATION LOAD CURRENT (A) OUTPUT VOLTAGE (V) 5.15 1.5

8302 G01

5.00 4.90 0.5 1.0 2.0 4.85 4.80 5.20 5.10 5.05 4.95 2.5 3.0 VIN = 5V VIN = 12V VIN = 24V

8302faFor more information www.linear .com/L T8302 Typical perForMance characTerisTics INTVCC Voltage vs VIN INTVCC UVLO Threshold (RFB-VIN) Voltage RREF Regulation Voltage RREF Line Regulation TC Pin Voltage EN/UVLO Enable Threshold EN/UVLO Hysteresis Current INTVCC Voltage vs Temperature TA = 25°C, unless otherwise noted. TEMPERATURE (°C) –50 VEN/UVLO (V) 1.220 1.230 150

8302 G10

1.210 1.200 0 50 100–25 25 75 125 1.240 1.215 1.225 1.205 1.235 RISING FALLING TEMPERATURE (°C) –50 IHYST (µA) 25 75 150

8302 G11

–25 0 50 100 125 TEMPERATURE (°C) –50

2.80 VINTVCC (V)

2.85 2.90 2.95 3.00 0 50 100 150

8302 G12

3.05 3.10 –25 25 75 125 IINTVCC = 0mA IINTVCC = 10mA VIN (V) VINTVCC (V) 2.95 3.00 3.05 35 4020 25 30

8302 G13

2.90 2.85 10 15 45 2.80 3.10 IINTVCC = 0mA IINTVCC = 10mA TEMPERATURE (°C) –50 2.2 VINTVCC (V) 2.3 2.4 2.5 2.6 0 50 100 150

8302 G14

2.7 2.8 –25 25 75 125 FALLING RISING TEMPERATURE (°C) –50 VOLTAGE (mV) 150

8302 G15

–20 –40 0 50 100–25 25 75 125 –10 –30

30 IRFB = 125µA

IRFB = 100µA IRFB = 75µA TEMPERATURE (°C) –50

0.990 VRREF (V)

0.992 0.996 0.998 1.000 1.010 1.004 0 50 75

8302 G16

0.994 1.006 1.008 1.002 –25 25 100 125 150 VIN (V) VRREF (V) 1.002 1.006 1.010

8302 G17

0.998 0.994 1.000 1.004 1.008 0.996 0.992 0.990 10 20 30 50 TEMPERATURE (°C) –50 VTC (V)1.1 1.3 150

8302 G18

0.9 0.7 0 50 100–25 25 75 125 1.5 1.0 1.2 0.8 1.4

8302fa For more information www.linear .com/L T8302 Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time RDS(ON) Switch Current Limit Maximum Switching Frequency Typical perForMance characTerisTics TA = 25°C, unless otherwise noted. TEMPERATURE (°C) –50 RESISTANCE (m/uni03A9) 120 160 200 25 75 150

8302 G19

–25 0 50 100 125 TEMPERATURE (°C) –50 ISW (A) 25 75 150

8302 G20

–25 0 50 100 125 MAXIMUM CURRENT LIMIT MINIMUM CURRENT LIMIT TEMPERATURE (°C) –50 FREQUENCY (kHz) 300 400 500 25 75 150

8302 G21

–25 0 50 100 125 TEMPERATURE (°C) –50 FREQUENCY (kHz) 25 75 150

8302 G22

–25 0 50 100 125 TEMPERATURE (°C) –50 TIME (ns) 100 200 300 400 –25 0 25 50

8302 G23

TEMPERATURE (°C) –50 TIME (ns) 100 200 300 400 –25 0 25 50

8302 G24

8302faFor more information www.linear .com/L T8302 pin FuncTions EN/UVLO ( Pin 1): Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8302. Pull the pin below 0.3V to shut down the LT8302. This pin has an ac- curate 1.214V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resistor divider from V IN to ground. A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. INTVCC (Pin 2): Internal 3V Linear Regulator Output. The INTVCC pin is supplied from VIN and powers the internal control circuitry and gate driver. Do not overdrive the INTVCC pin with any external supply, such as a third winding supply. Locally bypass this pin to ground with a minimum 1µF ceramic capacitor. VIN (Pin 3): Input Supply. The VIN pin supplies current to the internal circuitry and serves as a reference voltage for the feedback circuitry connected to the R FB pin. Locally bypass this pin to ground with a capacitor. GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed pad provides both electrical contact to ground and good thermal contact to the printed circuit board . Solder the exposed pad directly to the ground plane. SW (Pin 5): Drain of the Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. RFB (Pin 6): Input Pin for External Feedback Resistor. Connect a resistor from this pin to the transformer primary SW pin. The ratio of the RFB resistor to the RREF resistor, times the internal voltage reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. RREF (Pin 7): Input Pin for External Ground Referred Ref- erence Resistor. The resistor at this pin should be in the range of 10k, but for convenience in selecting a resistor divider ratio, the value may range from 9.09k to 11.0k. TC (Pin 8): Output Voltage Temperature Compensation. The voltage at this pin is proportional to absolute temperature (PTAT) with temperature coefficient equal to 3.35mV/°K, i.e., equal to 1V at room temperature 25°C. The TC pin voltage can be used to estimate the LT8302 junction tem- perature. Connect a resistor from this pin to the RREF pin to compensate the output diode temperature coefficient.

8302fa For more information www.linear .com/L T8302 block DiagraM DRIVER INTVCC VIN N:1 A2 RSENSE TC 8302 BD RREF RREF REN2 REN1 RTC RFB gm 1.214V OSCILLATOR LDO BOUNDARY DETECTOR START-UP , REFERENCE, CONTROL PTAT VOL TAGE R M1 GND 4, EXPOSED PAD PIN 9 Q S VIN VIN CIN RFB SW L1A L1B C OUT DOUT VOUT+ VOUT– 25µA INTVCC

1 EN/UVLO

2.5µA 1:4 – +

8302faFor more information www.linear .com/L T8302 operaTion The LT8302 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation bound- ary. Opto-isolator circuits waste output power, and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unit- to-unit variation and aging over lifetime. Circuits employing extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT8302 samples the isolated output voltage through the primary-side flyback pulse waveform. In this manner, neither opto-isolator nor extra transformer winding is re- quired for regulation. Since the LT8302 operates in either boundar y conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method im- proves load regulation without the need of external load compensation components. The LT8302 is a simple to use micropower isolated fly - back converter housed in a thermally enhanced 8-lead SO package. The output voltage is programmed with two external resistors. An optional TC resistor provides easy output diode temperature compensation. By integrating the loop compensation and soft-start inside, the part reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switching regulators including reference, regulators, oscillator, logic, current amplifier, current comparator, driver, and power switch. The novel sections include a flyback pulse sense circuit, a sample- and-hold error amplifier, and a boundary mode detector, as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Quasi-Resonant Boundary Mode Operation The LT8302 features quasi-resonant boundary conduction mode operation at heavy load, where the chip turns on the primary power switch when the secondary current is zero and the SW rings to its valley. Boundary conduction mode is a variable frequency, variable peak-current switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around VIN. A boundary mode detector senses this event and turns the power switch back on at its valley.

8302fa For more information www.linear .com/L T8302 operaTion Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduc - tion mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit subharmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter, boundary conduction mode in - creases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switching and gate charge losses. To avoid this scenario, the LT8302 has an additional internal oscillator, which clamps the maximum switching frequency to be less than 380kHz. Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8302 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch cur- rent limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8302 starts to fold back the switching frequency while keeping the minimum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sample-and- hold error amplifier. Meanwhile, the part switches between sleep mode and active mode, thereby reducing the effec- tive quiescent current to improve light load efficiency. In this condition, the LT8302 runs in low ripple Burst Mode operation. The typical 12kHz minimum switching frequency determines how often the output voltage is sampled and also the minimum load requirement.

8302faFor more information www.linear .com/L T8302 applicaTions inForMaTion Output Voltage The RFB and RREF resistors as depicted in the Block Diagram are external resistors used to program the output voltage. The LT8302 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier, which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the V IN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: VFLBK = (VOUT + VF + ISEC • ESR) • NPS VF = Output diode forward voltage ISEC = T ransformer secondary current ESR = Total impedance of secondary circuit NPS = T ransformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current, I RFB, by the R FB resistor and the flyback pulse sense circuit (M2 and M3). This current, I RFB, also flows through the RREF resistor to generate a ground-referred voltage. The resulting voltage feeds to the inverting input of the sample- and-hold error amplifier. Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the VFLBK equation can be assumed to be zero. The internal reference voltage, V REF, 1.00V, feeds to the noninverting input of the sample-and-hold error ampli- fier. The relatively high gain in the overall loop causes the voltage at the RREF pin to be nearly equal to the internal reference voltage VREF. The resulting relationship between VFLBK and VREF can be expressed as: VFLBK RFB ⎠⎟•RREF = VREF or VFLBK = VREF • RFB RREF VREF = Internal reference voltage 1.00V Combination with the previous V FLBK equation yields an equation for VOUT, in terms of the RFB and RREF resistors, transformer turns ratio, and diode forward voltage: VOUT = VREF • RFB RREF ⎠⎟• 1 NPS ⎠⎟– VF Output Temperature Compensation The first term in the VOUT equation does not have tempera- ture dependence, but the output diode forward voltage, VF, has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coefficient pro- duces approximately 200 mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible ef - fect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however, the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. The LT8302 junction temperature usually tracks the output diode junction temperature to the first order. To compensate the negative temperature coefficient of the output diode, a resistor, R TC, connected between the TC and RREF pins generates a proportional-to-absolute-temperature (PTAT) current. The PTAT current is zero at 25°C, flows into the RREF pin at hot temperature, and flows out of the RREF pin at cold temperature. With the R TC resistor in place, the output voltage equation is revised as follows: VOUT = VREF • RFB RREF

• 1 NPS – VF TO( ) – VTC / T( ) • T –TO( ) • RFB RTC
• 1 NPS – VF / T( ) • T–TO( ) TO=Room temperature 25° C VF / T( ) = Output diode forward voltage temperature coefficient VTC / T( ) = 3.35mV/ C

8302fa For more information www.linear .com/L T8302 applicaTions inForMaTion To cancel the output diode temperature coefficient, the following two equations should be satisfied: VOUT = VREF • RFB RREF

• 1 NPS – VF TO( ) VTC/ T( ) • RFB RTC
• 1 NPS = – VF/ T( ) Selecting Actual RREF, RFB, RTC Resistor Values The LT8302 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the R FB and R TC resistor values. Therefore, a simple 2-step sequential process is recommended for selecting resistor values. Rearrangement of the expression for VOUT in the previous sections yields the starting value for RFB: RFB = RREF •NPS • VOUT + VF TO( )( ) VREF VOUT = Output voltage VF (TO) = Output diode forward voltage at 25°C = ~0.3V NPS = T ransformer effective primary-to-secondary turns ratio The equation shows that the RFB resistor value is indepen- dent of the RTC resistor value. Any RTC resistor connected between the TC and RREF pins has no effect on the output voltage setting at 25°C because the TC pin voltage is equal to the RREF regulation voltage at 25°C. The R REF resistor value should be approximately 10k because the LT8302 is trimmed and specified using this value. If the RREF resistor value varies considerably from 10k, additional errors will result. However, a variation in RREF up to 10% is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB/RREF ratios. First, build and power up the application with the starting RREF, RFB values (no RTC resistor yet) and other compo- nents connected, and measure the regulated output volt- age, VOUT(MEAS). The new RFB value can be adjusted to: RFB(NEW) = VOUT VOUT(MEAS)
• RFB Second, with a new RFB resistor value selected, the output diode temperature coefficient in the application can be tested to determine the R TC value. Still without the R TC resistor, the VOUT should be measured over temperature at a desired target output load. It is very important for this evaluation that uniform temperature be applied to both the output diode and the LT8302. If freeze spray or a heat gun is used, there can be a significant mismatch in temperature between the two devices that causes sig- nificant error. Attempting to extrapolate the data from a diode data sheet is another option if there is no method to apply uniform heating or cooling such as an oven. With at least two data points spreading across the operating temperature range, the output diode temperature coef - ficient can be determined by: – δVF/δT( ) = VOUT T1( ) – VOUT T2( ) T1– T2 Using the measured output diode temperature coefficient, an exact R TC value can be selected with the following equation: RTC = δVTC/δT( ) – δVF/δT( )
• RFB NPS Once the RREF, RFB, and RTC values are selected, the regula- tion accuracy from board to board for a given application will be very consistent, typically under ±5% when includ- ing device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ±1%). However, if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT.

current at the expense of a higher switch voltage. the amount of output power at given input voltages. Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output Figure 3. Output Power for 12V Output Figure 4. Output Power for 24V Output

8302 F02

8302 F03

8302 F04

8302 F01

and may cause instability at light load. information should be carefully considered. shows the details of these transformers. Table 1. Predesigned T ransformers–Typical Specifications

8302faFor more information www.linear .com/L T8302 applicaTions inForMaTion Turns Ratio Note that when choosing an RFB/RREF resistor ratio to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast, the use of simple ratios of small integers, e.g., 3:1, 2:1, 1:1, etc., provides more freedom in settling total turns and mutual inductance. Typically, choose the transformer turns ratio to maximize available output power. For low output voltages (3.3V or 5V), a N:1 turns ratio can be used with multiple pri - mary windings relative to the secondary to maximize the transformer’s current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (V LEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 65V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure NPS < 65V – VIN(MAX) – VLEAKAGE VOUT + VF For larger N:1 values, choose a transformer with a larger physical size to deliver additional current. In addition, choose a large enough inductance value to ensure that the switch-off time is long enough to accurately sample the output voltage. For lower output power levels, choose a 1:1 or 1:N trans- former for the absolute smallest transformer size. A 1:N transformer will minimize the magnetizing inductance (and minimize size), but will also limit the available output power. A higher 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer speci- fies turns ratio accuracy within ±1%. Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8302, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output voltage regulation will be maintained independent of winding re- sistance due to the boundary/discontinuous conduction mode operation of the LT8302. Leakage Inductance and Snubbers T ransformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored en- ergy must be dissipated. It is very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Fig - ure 5, the reflected output voltage on the primary plus VIN should be kept below 50V. This leaves at least 15V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound transformers or for excessive leakage inductance.

Figure 5. Maximum Voltages for SW Pin Flyback Waveform boundary mode detector for approximately 250ns. should be limited to less than 250ns. (RC + DZ) snubber circuit in Figure 6 is recommended. SW pin from exceeding its 65V absolute maximum rating. Figure 6. (RC + DZ) Snubber Circuit

8302 F06

converted to heat and will not be delivered to the load. tor in series with a 39Ω resistor is a good starting point. voltage rating higher than the maximum SW pin voltage. diode. A 1.5W Zener diode is typically recommended.

LT8302 in shutdown with quiescent current less than 2µA.

8302 F07

Figure 7. Undervoltage Lockout (UVLO) minimum amount of time and with a minimum frequency. serve as a minimum load if pre-loading is not acceptable.

Step 1: Select the transformer turns ratio. different transformer turns ratio. Table 2. Switch Voltage Stress and Output Current Capability vs Step 2: Determine the primary inductance. 60% larger than the minimum values calculated above. LPRI = 9µH is then chosen in this example.

8302faFor more information www.linear .com/L T8302 applicaTions inForMaTion Example: ISW = 5V •1.5A •2 0.8 •12V •0.57 fSW = 277kHz The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 7A is necessary to work with the LT8302. The 750311564 from Würth is chosen as the flyback transformer. Step 3: Choose the output diode. Tw o main criteria for choosing the output diode include forward current rating and reverse-voltage rating. The maximum load requirement is a good first-order guess at the average current requirement for the output diode. Under output short-circuit condition, the output diode needs to conduct much higher current. Therefore, a con- servative metric is 60% of the maximum switch current limit multiplied by the turns ratio: IDIODE(MAX) = 0.6 • ISW(MAX) • NPS Example: IDIODE(MAX) = 8.1A Next calculate reverse voltage requirement using maxi- mum V IN: VREVERSE = VOUT + VIN(MAX) NPS Example: VREVERSE = 5V + 32V 3 = 15.7V The PDS835L (8A, 35V diode) from Diodes Inc. is chosen. Step 4: Choose the output capacitor. The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. Use the following equation to calculate the output capacitance: COUT = LPRI •ISW2 2 •VOUT •ΔVOUT Example: Design for output voltage ripple less than ±1% of V OUT, i.e., 100mV. COUT = 9µH• 4.5A( ) 2 2 •5V •0.1V = 182µF Remember ceramic capacitors lose capacitance with ap- plied voltage. The capacitance can drop to 40% of quoted capacitance at the maximum voltage rating. So a 220µF, 6.3V rating ceramic capacitor is chosen. Step 5: Design snubber circuit. The snubber circuit protects the power switch from leak- age inductance voltage spike. A (RC + DZ) snubber is recommended for this application. A 470pF capacitor in series with a 39Ω resistor is chosen as the RC snubber. The maximum Zener breakdown voltage is set according to the maximum V IN: VZENNER(MAX) ≤ 60V – VIN(MAX) Example: VZENNER(MAX) ≤ 60V – 32V = 28V A 24V Zener with a maximum of 26V will provide optimal protection and minimize power loss. So a 24V, 1.5W Zener from Central Semiconductor (CMZ5934B) is chosen. Choose a diode that is fast and has sufficient reverse voltage breakdown: VREVERSE > VSW(MAX) VSW(MAX) = VIN(MAX) + VZENNER(MAX) Example: VREVERSE > 60V A 100V, 1A diode from Diodes Inc. (DFLS1100) is chosen.

8302fa For more information www.linear .com/L T8302 applicaTions inForMaTion Step 6: Select the RREF and RFB resistors. Use the following equation to calculate the starting values for RREF and RFB: RFB = RREF •NPS • VOUT + VF TO( )( ) VREF RREF = 10k Example: 1.00V = 159k For 1% standard values, a 158k resistor is chosen. Step 7: Adjust RFB resistor based on output voltage. Build and power up the application with application com- ponents and measure the regulated output voltage. Adjust RFB resistor based on the measured output voltage: RFB(NEW) = VOUT VOUT(MEASURED)

• RFB Example: RFB = 5V 5.14V •158k = 154k Step 8: Select R TC resistor based on output voltage temperature variation. Measure output voltage in a controlled temperature envi- ronment like an oven to determine the output temperature coefficient. Measure output voltage at a consistent load current and input voltage, across the operating tempera- ture range. Calculate the temperature coefficient of V – δVF/δT( ) = VOUT T1( ) – VOUT T2( ) T1– T2 RTC = 3.35mV/°C – δVF/δT( )
• RFB NPS Example: 100°C – 0°C( ) = 1.48mV /°C RTC = 3.35mV/°C 1.48mV/°C • 154 ⎝⎜ ⎞ ⎠⎟ = 115k Step 9: Select the EN/UVLO resistors. Determine the amount of hysteresis required and calculate R1 resistor value: VIN(HYS) = 2.5µA • R1 Example: Choose 2V of hysteresis, R1 = 806k Determine the UVLO thresholds and calculate R2 resistor value: VIN(UVLO+)=1.228V • R1+ R2( ) R2 + 2.5µA •R1 Example: Set V IN UVLO rising threshold to 7.5V: R2 = 232k VIN(UVLO+) = 7.5V VIN(UNLO–) = 5.5V Step 10: Ensure minimum load. The theoretical minimum load can be approximately estimated as: ILOAD(MIN)= 9µH• 0.96A( ) 2 •12.7kHz 2 •5V =10.5mA Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the con - verter delivers more energy than what is consumed at the output. The real minimum load for this application is about 10mA. In this example, a 500Ω resistor is selected as the minimum load.

8302faFor more information www.linear .com/L T8302 Typical applicaTions 8V to 32VIN/12VOUT Isolated Flyback Converter 8V to 32VIN/3.3VOUT Isolated Flyback Converter Efficiency vs Load Current Load and Line Regulation AMBIENT TEMPERATURE (°C) –50 OUTPUT VOLTAGE (V) 3.30 3.40 150

8302 TA03b

3.20 3.10 0 50 100–25 25 75 125 3.50 3.25 3.35 3.15 3.45 RTC = 105k RTC = OPEN VIN = 12V IOUT = 1A Output Temperature Variation LOAD CURRENT (mA) EFFICIENCY (%) 200 400 600 800 1000

8302 TA02b

VIN = 12V VIN = 24V LOAD CURRENT (mA)

11.2 OUTPUT VOLTAGE (V)

11.4 11.6 11.8 12.0 12.2 12.4 200 400 600 800 1000 1200

8302 TA02c

VIN = 12V VIN = 24V VIN L T8302 SW 9µH VIN 8V TO 32V 1:1 9µH RFB RREF 470pFZ1 10µF 1µF 47µF V OUT+ 12V 5mA TO 0.8A (V IN = 12V) 5mA TO 1.1A (VIN = 24V) VOUT– 39/uni03A9 121k OPEN 232k 806k 10k D1: DIODES DFLS1100 D2: DIODES PDS540 T1: WURTH 750313443 Z1: CENTRAL CMZ5934B

8302 TA02aTC

9µH VIN 8V TO 32V 4:1 0.56µH RFB RREF 470pFZ1 10µF 1µF 470µF V OUT+ 3.3V 20mA TO 2.7A (V IN = 12V) 20mA TO 3.8A (VIN = 24V) VOUT– 39/uni03A9 140k 105k 232k 806k 10k D1: DIODES DFLS1100 D2: DIODES PDS1040L T1: WURTH 750311625 Z1: CENTRAL CMZ5934B

8302 TA03TC

8302fa For more information www.linear .com/L T8302 Typical applicaTions 8V to 36VIN/±12VOUT Isolated Flyback Converter 8V to 36VIN/24VOUT Isolated Flyback Converter 8V to 36VIN/48VOUT Isolated Flyback Converter VIN L T8302 SW 9µH VIN 8V TO 36V 1:1:1 9µH RFB RREF 470pFZ1 10µF 1µF 22µF V OUT1+ 12V 5mA TO 0.4A (V IN = 12V) 5mA TO 0.55A (VIN = 24V) VOUT2– 39/uni03A9 121k OPEN 232k 806k 10k D1: DIODES DFLS1100 D2, D3: DIODES PDS360 T1: WURTH 750313443 Z1: CENTRAL CMZ5934B

8302 TA04TC

9µH 22µF V OUT2+ 12V 5mA TO 0.4A (V IN = 12V) 5mA TO 0.55A (VIN = 24V) VOUT2– EN/UVLO GND INTVCC VIN L T8302 SW 9µH VIN 8V TO 36V 1:2 36µH RFB RREF 470pFZ1 10µF 1µF 10µF V OUT+ 24V 2.5mA TO 0.4A (V IN = 12V) 2.5mA TO 0.55A (VIN = 24V) VOUT– 39/uni03A9 121k OPEN 232k 806k 10k D1: DIODES DFLS1100 D2: DIODES SBR2U150SA T1: WURTH 750313445 Z1: CENTRAL CMZ5934B

8302 TA05TC

9µH VIN 8V TO 36V 1:4 144µH RFB RREF 470pFZ1 10µF 1µF 2.2µF V OUT+ 48V 1.2mA TO 0.2A (V IN = 12V) 1.2mA TO 0.27A (VIN = 24V) VOUT– 39/uni03A9 121k OPEN 232k 806k 10k D1: DIODES DFLS1100 D2: DIODES SBR1U200P1 T1: WURTH 750313457 Z1: CENTRAL CMZ5934B

8302 TA06TC

8302faFor more information www.linear .com/L T8302 Typical applicaTions VIN L T8302 SW 9µH VIN 8V TO 32V 3:1 1µH RFB RREF 470pFZ1 D1 D2C1 10µF 1µF 220µF 10µF 4.7µF V OUT+ 5V/1.1A (VIN = 5V) 5V/2.0A (VIN = 12V) 5V/2.9A (VIN = 24V) VOUT– 39/uni03A9 5/uni03A9 V CC DRAIN L T8309 GATE INTV CCM1 GND 154k OPEN 232k 806k 2.1k 10k D1: DIODES DFLS1100 D2: CENTRAL CMMSH1-60 M1: INFINEON BSC059N04LS T1: WURTH 750311564 Z1: CENTRAL CMZ5934B

8302 TA07

LOAD CURRENT (A) EFFICIENCY (%) 1.5 2.5

8302 TA07b

0.5 1.0 2.0 3.0 8V to 32VIN/5VOUT Isolated Flyback Converter with LT8309 –4V to –42VIN/12VOUT Buck-Boost Converter –18V to –42VIN/–12VOUT Negative Buck Converter Efficiency vs Load Current Efficiency vs Load Current Efficiency vs Load Current VIN SW L T8302 12µH D1 GND RFB RREF EN/UVLO 47µF D1: DIODES PMEG6030EP L1: WÜRTH 744770112 Z1: CENTRAL CMHZ5243B 1µF V IN –4V TO –42V 10µF 10k

8302 TA08a

12V/0.45A (VIN = –5V) 12V/0.8A (VIN = –12V) 12V/1.1A (VIN = –24V) 12V/1.3A (VIN = –42V)R4 118k INTV CC LOAD CURRENT (mA) EFFICIENCY (%) 200 400 800 600 1000 1200 1400

8302 TA08b

VIN = –5V VIN = –12V VIN = –24V VIN = –42V VIN L T8302 12µH VOUT –12V 1.8A D1 Z1 SW R REF EN/UVLO RFBEN/UVLO D1: DIODES PMEG6030EP L1: WÜRTH 744770112 Z1: CENTRAL CMHZ5243B 1µFV IN –18V TO –42V 10µF 47µF 10k

8302 TA09a

LOAD CURRENT (mA) EFFICIENCY (%) 100 500 1000 1500 2000

8302 TA09b

VIN = –18V VIN = –24V VIN = –42V

8302fa For more information www.linear .com/L T8302 package DescripTion Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. .016 – .050 (0.406 – 1.270) .010 – .020 0°– 8° TYP .008 – .010 (0.203 – 0.254) S8E 1013 REV A .053 – .069 (1.346 – 1.752) .014 – .019 (0.355 – 0.483) TYP .004 – .010 (0.101 – 0.254) .080 – .098 (2.032 – 2.489) .118 – .138 (2.997 – 3.505) .050 (1.270) BSC 1 2 3 4 .150 – .157 (3.810 – 3.988) NOTE 3 8 7 .005 (0.13) MAX 6 5 .189 – .197 (4.801 – 5.004) NOTE 3 .228 – .244 (5.791 – 6.197) .160 ±.005 (4.06 ±0.127) .118 (2.99) REF RECOMMENDED SOLDER PAD LAYOUT .045 ±.005 (1.143 ±0.127) .050 (1.27) BSC INCHES (MILLIMETERS) NOTE: 1. DIMENSIONS IN 2. DRAWING NOT TO SCALE 3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm) 8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad (Reference LTC DWG # 05-08-1857 Rev A) .089 (2.26) REF .030 ±.005 (0.76 ±0.127) TYP .245 (6.22) MIN

8302faFor more information www.linear .com/L T8302 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa- tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER A 11/14 Modified I Q and IHYS Conditions Modified LPRI Equation Modified Schematic Updated Related Parts

8302fa For more information www.linear .com/L T8302  LINEAR TECHNOLOGY CORPORATION 2013 LT

1114 REV A • PRINTED IN USA

Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear .com/L T8302 relaTeD parTs Typical applicaTion 4V to 42VIN/48VOUT Boost Converter Efficiency vs Load Current VIN SW L T8302 22µH D1 GND RFB RREF EN/UVLO C3 10µF D1: DIODES PDS560 L1: WÜRTH 7443551221 Z1: CENTRAL CMHZ5262B 1µF V IN 4V TO 42V 10µF 10k

8302 TA10a

48V/1.4A (VIN = 42V) 48V/0.8A (VIN = 24V) 48V/0.4A (VIN = 12V) 48V/0.15A (VIN = 5V) 464k INTV CC LOAD CURRENT (mA) EFFICIENCY (%) 100 250 500

8302 TA10b

VIN = 5V VIN = 12V VIN = 24V VIN = 42V PART NUMBER DESCRIPTION COMMENTS LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flyback 5-Lead TSOT-23 LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 LT3573/LT3574 LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V ≤ V IN ≤ 100V, No-Opto Flyback, MSOP-16(12) LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC V IN and VOUT Limited Only by External Components LT3757A/LT3759 LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/80V Boost/Flyback Converters Monolithic with Integrated 5A/3.3A Switch