A6265KLPTR-T

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Description

The A6265 is a DC-to-DC converter controller, providing a programmable constant current output for driving high power LEDs in series. Driving the LEDs in series ensures identical currents and uniform brightness. For automotive applications, optimum performance is achieved when driving between 2 and 10 LEDs at currents up to 1 A. The A6265 provides a cost-effective solution using an external logic-level MOSFET and minimum additional external components. The maximum LED current is set with a single external sense resistor and can be modified using a current reference input. Direct PWM control is possible via the Enable input, which also provides a shutdown mode. This DC-DC converter can be configured as a ground-referenced boost converter or as a supply-referenced boost converter providing buck-boost capability. The buck-boost topology used ensures that there is no leakage path through the LEDs when in shutdown and no inrush current at power-up. Integrated diagnostics and two fault outputs give indication of undervoltage, chip overtemperature, output open circuit, LED short circuit and LED undercurrent, and can be configured to provide short to supply and short to ground protection for the LED connections. A unique feature is the ability to detect one or more shorted LEDs. The device is provided in a 16-pin TSSOP package with exposed thermal pad (suffix LP). It is lead (Pb) free, with 100% matte tin leadframe plating. A6265-DS, Rev. 5 Features and Benefits ▪ AEC Q100 Grade 0 Automotive Qualified ▪ Constant current LED drive ▪ 6.5 to 50 V supply ▪ Boost or buck-boost modes ▪ Drives between 2 and 10 LEDs in series ▪ Programmable switching frequency 100 to 700 kHz ▪ Open LED overvoltage indication and protection ▪ Single and multiple LED short indication ▪ LED short to ground and supply protection ▪ PWM dimming control ▪ 10 μA shutdown current including LED leakage Automotive High Current LED Controller Package: 16-pin TSSOP with exposed thermal pad (suffix LP) Applications: Typical Application Diagrams Not to scale A6265 Buck-Boost Mode (Supply-referenced boost) Boost Mode ▪ Automotive high power LED lighting systems ▪ Fog lights, reversing lights, daytime running lights ▪ Headlights LN LP LF SG SP SN GND LA Fault Flags VBAT 12 V or 24 V Power net (50 V max) VREG FF1 FF2 EN IREF CKOUT OSC Enable A6265 VIN GND LA VBAT 12 V or 24 V (50 V max) LN LP LF SG SP SN Fault Flags Power net FF1 FF2 EN IREF CKOUT OSC Enable A6265 VREG VIN Boost Mode VBAT(min) (V) Maximum Quantity of LEDs 71 2 81 3 91 5 Vf of each LED = 3.5 V, D(max) = 85% Buck Boost Mode VBAT(min) (V) Maximum Quantity of LEDs 71 0 81 1 91 3 Vf of each LED = 3.5 V, D(max) = 85%

Automotive High Current LED ControllerA6265 2Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A. Absolute Maximum Ratings With respect to GND at TA = 25°C, unless otherwise specified Characteristic Symbol Notes Rating Unit Load Supply Voltage V IN –0.3 to 50 V Pins FF1, EN –0.3 to 50 V Pins FF2, CKOUT –0.3 to 6.5 V Pin OSC –0.3 to 6.5 V Pin SG –0.3 to 6.5 V Pins LA, LN –0.3 to 50 V Pin LF With respect to LA –6 to 6 V Pin LP With respect to LN –6 to 6 V Pin SP, SN –0.3 to 5 V Pin VREG –0.3 to 7 V Pin IREF –0.3 to 7 V Junction Temperature T J(max) 150 °C Storage Temperature Range T stg –55 to 150 °C Operating Temperature Range T A Range K –40 to 150 °C Selection Guide Part Number Packing Package A6265KLPTR-T 4000 pieces per 13-in. reel 16-pin TSSOP with exposed thermal pad Thermal Characteristics may require derating at maximum conditions, see application information Characteristic Symbol Test Conditions* Value Unit Package Thermal Resistance (Junction to Ambient) RθJA On 4-layer PCB based on JEDEC standard 34 ºC/W On 2-layer PCB with 3.8 in.2 of copper area each side 43 ºC/W Package Thermal Resistance (Junction to Exposed Pad) RθJP 2 ºC/W *Additional thermal information available on the Allegro website

Automotive High Current LED ControllerA6265 3Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. Pin-out Diagram EN FF1 FF2 CKOUT IREF OSC SN SP LF LA LN LP VIN VREG GND SG PAD Terminal List Table Number Name Function

1 EN Enable chip

2 FF1 Fault flag

3 FF2 Fault flag

4 CKOUT Oscillator output, with phase shift

5 IREF Current reference

6 OSC Oscillator input/frequency set

7 SN Switch current sense –ve input

8 SP Switch current sense +ve input

9 SG Switch gate drive

10 GND Ground

11 VREG Internal regulator capacitor

12 VIN Main supply

13 LP Load current sense +ve input

14 LN Load current sense –ve input

15 LA LED string voltage sense

16 LF Reference LED voltage sense

– PAD Exposed thermal pad

Automotive High Current LED ControllerA6265 4Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. Functional Block Diagrams SG VIN GND VREG Slope Gen OSC Osc EN FF1 VREG Fault Detect SP SN LN LP LA Shut- down Control Logic RSL RSS ROSC VBAT LF LED Open and Short Detect IREF QR S Temp Monitor IREF CKOUT FF2 Buck-Boost Boost SG VIN GND Slope Gen OSC Osc EN FF1 VREG Fault Detect SP SN LN LP LA Shut- down Control Logic RSS ROSC VBAT LF LED Open and Short Detect Temp Monitor IREF See Functional Description section. CKOUT FF2 RSL IREF VREG AL AE AC AS AL AE AC AS AL AE AC AS QR S

Automotive High Current LED ControllerA6265 5Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. Supply and Reference VIN Functional Operating Range2 6.5 – 50 V VIN Quiescent Current IINQ SG open circuit – – 8 mA IINS EN = GND – 4 10 μA VREG Output Voltage V REG I REG = 0 to 2 mA 4.75 5 5.25 V Gate Output Drive Turn-On Time t r CLOAD = 1 nF, 20% to 80% – 35 – ns Turn-Off Time t f CLOAD = 1 nF, 80% to 20% – 35 – ns Maximum Duty Cycle D t ON × fOSC 80 85 – % Pull-Up On Resistance R DS(on)UP TJ = 25°C, IGHx = –100 mA – 1.7 – Ω TJ = 150°C, IGHx = –100 mA – – 3.5 Ω Pull-Down On Resistance R DS(on)DN TJ = 25°C, IGLx = 100 mA – 1 – Ω TJ = 150°C, IGLx = 100 mA – – 2 Ω Output High Voltage V SGH I SG = –100 μA VREG – 0.1 – V REG V Output Low Voltage V SGL I SG = 100 μA – – 0.1 V Logic Inputs and Outputs Fault Output (Open Drain) V OL I OL = 1 mA, fault not asserted – – 0.4 V Fault Output FF1 Sink Current1 I OH(snk) 0.4 V < VO < 50 V, fault not asserted – 1.3 – mA Fault Output FF1 Leakage Current1 IOH1(lkg) VO = 12 V, fault asserted –1 – 1 μA Fault Output FF2 Leakage Current1 I OH2(lkg) VO = 5 V, fault asserted –5 – 5 μA Input Low Voltage V IL – – 0.8 V Input High Voltage V IH 2 – – V Input Hysteresis V Ihys 225 330 – mV Enable Input Internal Clamp Voltage V ENC – 8.4 – V Enable Input Current Limit Resistor R EN Between EN and internal clamp – 200 – k Ω Boost Mode Select Voltage V LNB Defined by V LN – – 1.2 V Buck-Boost Mode Select Voltage V LNBB Defined by V LN 4.0 – – V Disable Time t DIS fOSC = 350 kHz – 94 – ms Oscillator Oscillator Frequency f OSC ROSC = 43 kΩ – 500 – kHz ROSC = 62 kΩ 315 350 385 kHz ROSC = VREG – 350 – kHz OSC Pin Voltage V OSC R OSC = 62 kΩ 1.15 1.2 1.25 V CKOUT Output Delay t DC OSC input rise to CKOUT rise – 150 – ns OSC Input Low Voltage V OIL – – 0.8 V OSC Input High Voltage V OIH 3.5 – – V OSC Input Hysteresis V Oihys 300 600 – mV ELECTRICAL CHARACTERISTICS1 Valid at TJ = –40°C to 150°C, VIN = 8 to 40 V; unless otherwise noted Characteristics Symbol Test Conditions Min. Typ. Max. Unit Continued on the next page…

Automotive High Current LED ControllerA6265 6Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. ELECTRICAL CHARACTERISTICS1 (continued) Valid at TJ = –40°C to 150°C, VIN = 8 to 40 V; unless otherwise noted Characteristics Symbol Test Conditions Min. Typ. Max. Unit Oscillator (continued) OSC Watchdog Period t OSWD Between successive rising edges 7 – – μs CKOUT Output High Voltage V COH I OH = –1 mA V REG – 1 – V REG V CKOUT Output Low Voltage V COL I OL = 1 mA – – 0.4 V LED Current Sense Input Bias Current LN (BB mode)

3 I LN LP = LN = VIN – 6 – μA

Input Bias Current LP (BB mode)3 I LP LP = LN = VIN – 220 – μA Input Bias Current LN (B mode)3 I LN LP = LN = 0 V – –2 – μA Input Bias Current LP (B mode)3 I LP LP = LN = 0 V – –25 – μA Differential Input Voltage (Active) V IDL EN = High, VIDL = VLP – VLN – 100 – mV Differential Input Voltage (PWM off) V IDLO EN = Low, VIDL = VLP – VLN – 8 – mV Input Common-Mode Range (BB mode)3 VCMLH VLP = VLN VIN –V IN + 1 V Input Common-Mode Range (B mode)3 VCMLL VLP = VLN 0 – 1 V Current Error E ISL [(10 × I LED × RSL) – 1] × 100 –5 – 5 % Switch Current Sense Input Bias Current I BIASS SP = SN = 0 to 2 V –30 – – μA Maximum Differential Input Voltage4 V IDS VIDS = VSP – VSN with D = 50% 330 410 490 mV Input Source Current I INS V IDS = 400 mV – 400 – μA Input Common-Mode Range V CMS VSP = VSN 0 – 2 V Diagnostics and Protection Fault Blank Timer5 tFB Start-up – 3 – ms VREG Undervoltage Turn-Off V REGUV Decreasing VREG 4.1 4.3 4.5 V VREG Undervoltage Hysteresis V REGUVhys 130 150 160 mV LED String Short Voltage V SCL 430 505 580 mV Non-Reference LED Short Offset Voltage VSCO 160 200 240 mV Reference LED Short Offset Voltage V SCOR 430 505 580 mV LED Open Voltage V OCL 5 5.5 6 V LF Bias Current (BB mode)3 I LF LF = LA = VIN + 1.7 V – 46 – μA LA Bias Current (BB mode)3 I LA LF = LA = VIN + 1.7 V – 170 – μA LF Bias Current (B mode)3 ILF LF = 1.7 V – 8 – μA LA Bias Current (B mode)3 I LA LA = 1.7 V – 34 – μA LED Undercurrent Voltage Difference6 VUCL – 1 – mV Open Fault Time-Out t OTO fOSC = 350 kHz – 94 – ms Overtemperature Warning Threshold T JF Temperature increasing – 170 – ºC Overtemperature Hysteresis T Jhys Recovery = T JF – TJhys –1 5– º C 1For input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin. 2Function is correct but parameters are not guaranteed below the general limit (8 V). 3BB mode = buck-boost (supply-referenced) mode, B mode = boost (ground-referenced) mode. 4Parameters guaranteed by design. 5Fault Blank timer not enabled for open-LED condition. 6Undercurrent when VSENSEL < VIDL– VUCL , where VSENSEL is the voltage across the LED current sense resistor RSL.

Automotive High Current LED ControllerA6265 7Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. Functional Description The A6265 is a DC-DC converter controller that is designed to drive series-connected high power LEDs in automotive applica- tions. It provides programmable constant current output at load voltages and currents limited only by the external components. For automotive applications optimum performance is achieved when driving between two and ten LEDs at currents up to 1 A. The A6265 can be configured as a standard boost converter or as a supply referenced boost converter. In the supply referenced configuration the load voltage is the difference between the boost voltage and the supply voltage. This difference can be greater than, equal to, or less than the supply voltage, effectively provid- ing a buck-boost capability. This configuration provides seam- less, uninterrupted operation over the wide supply voltage range possible in automotive applications and, because the output is ref- erenced to the positive supply, there is no load current to ground. This ensures that there is no leakage path through the LEDs when in shutdown and no inrush current at power-up. The A6265 integrates all necessary control elements to pro- vide a cost-effective solution using a single external logic-level MOSFET and minimum additional external passive components. The LED current is set by selecting an appropriate value for the sense resistor value and using the EN input to provide simple on-off control or for PWM brightness control using a suitable externally generated PWM signal. The LED current can be reduced in a single step by reducing the voltage between the IREF pin and GND to less than 1 V . The pin functions and circuit operation are described in detail in the following sections. Pin Functions VIN Supply to the control circuit. A bypass capacitor must be connected between this pin and GND. GND Ground reference connection. This pin should be connected directly to the negative supply. EN Logic input to enable operation. Can be used as direct PWM input. Chip enters low power sleep mode when low for longer than the disable time, t DIS. FF1 Fault Flag output and isolation control. Open drain current sink output, when high impedance indicates detection of a critical circuit fault. An external pull-up resistor should be connected to a suitable logic supply for simple logic fault flag operation or to the source of the PMOS FET used to isolate the load from the supply. Table 1 defines when FF1 is active. If FF1 is pulled low when an output short fault is indicated then the output disable will be overridden. FF2 Fault Flag output. Open drain output, when high impedance indicates detection of a circuit fault. An external pull-up resistor should be connected to a suitable logic supply. If VREG is not used, then the logic supply should not be pulled 300 mV above VREG. Table 1 defines when FF2 is active. If FF2 is pulled low when an open LED fault is indicated then the output disable will be overridden. OSC Resistor to ground to set the internal oscillator or clock input from external oscillator. When connected to VREG the oscillator runs at typically 350 kHz. Higher accuracy in the frequency is possible by connecting a resistor from this pin to ground or by driving this pin with an external precision oscillator. CKOUT Logic output at the oscillator frequency with phase shift. Used to drive succeeding controllers to interleave switching instants. IREF LED current reference modifier. A voltage input that can be used to reduce the LED current sense voltage. When connected to VREG, the current sense voltage, V IDL, and the value of the sense resistor, RSL , define the maximum LED current. SG Gate drive for external logic-level MOSFET low-side switch that connects the inductor to ground. SP, SN Sense amplifier connections for switch current limit sense resistor, RSS . LP Positive sense amplifier connection for LED current limit sense resistor, RSL . This pin is also the bias supply for the LED current sense amplifier. LN Negative sense amplifier connection for LED current limit sense resistor, RSL . The voltage at LN also determines whether the boost or buck-boost mode is configured. VREG Compensation capacitor for internal 5 V regulator. LA Anode reference connection to LEDs. Using an external resis- tor divider with the same ratio as the number of LEDs provides a measurement of the voltage across all LEDs in the load. This is compared to the voltage on the LF pin to provide shorted LED detection. In addition, it is compared against voltage references to

Worcester, Massachusetts 01615-0036 U.S.A. provide open circuit or shorted LED string detection. ing a setpoint reference for the inner loop. controlling the peak switch current on a cycle-by-cycle basis. level for the peak switch current is modified by a slope generator. by a small amount as the duty cycle increases (refer to figure 1). fixed frequency current control scheme. current and thus increasing the current delivered to the LEDs. low frequency and is not harmful. peak sense voltage (typical values) changes over a period of 3 μs. through the switch would be 360 mV/RSS . Figure 1. Slope compensation for peak switch current control.

Worcester, Massachusetts 01615-0036 U.S.A. to VREG, or one-step dimming by use of a single logic control. LEDs, this will have some effect on the color of the LEDs. audible noise due to the piezoelectric effect of the capacitor. any large variations in the input current. path remaining will be the path through the MOSFET. lator is designed to run between 100 and 700 kHz. Figure 2 shows the resulting fOSC for various values of ROSC. frequency will be set internally to approximately 350 kHz. the CKOUT pin of another A6265 as the external clock source. current pulses will be spread across the oscillator period. Figure 2. Internal oscillator frequency when set by ROSC

10Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. greater than the disable time, or a power cycle occurs. down, for example in the external switching MOSFET. required current level, FF2 will go low.

• the voltage across a single reference LED: V LED = VLF – VLP (5)
• the ratio of the voltage across all LEDs in a single string: VSTR = VLA – VLP (6) The voltage, VSTR , is derived from the voltage across all LEDs in the string, by an external resistor divider with a ratio equal to the quantity of LEDs in the string. To minimize the effects of the bias currents introducing an offset voltage, it is recommended that the resistor between LP and LA should be approximately 560 Ω. So for example, if eight LEDs were used, the ratio required would be an eighth, therefore the resistor connected between LA and the anode end of the LED string would be 3.9 kΩ;

Table 1. Fault Table

11Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. described in turn in the following sections. trical Characteristics table. appearing in the power circuitry.

• for the first (reference) LED: V STR > VLED + VSCOR (8)
• for other than the first (reference) LED: VLED > VSTR + VSCO (9) where VSTR and VLED are as defined above, VSCO is the nonrefer- ence LED short offset voltage, and VSCOR is the reference LED short offset voltage. VSCO and VSCOR are defined in the Electrical Characteristics table. When a short is present, the fault flag FF2 is high impedance, but the regulator continues to operate and drives the remaining LEDs with the correct regulated current. FF2 will remain high imped- ance while the short circuit condition is present. A short circuit on one or more LEDs will not cause a hazard because the output is current-controlled. If one LED fails and becomes a short circuit, then the remaining LEDs will continue to be lit with the same current through, and voltage across, each LED. Note—Accuracy: The output status monitor relies on all the LEDs in the load having a similar forward voltage drop. Where possible all the LEDs forming the load for a single controller should be taken from the same voltage bin. With only two or three LEDs a wider variation in forward voltage is acceptable, but the selection of LEDs from the same bin is more critical when higher numbers of LEDs are used in a single string. Shorted LED String or Output Short – A short circuit across the LED string, is detected when: V STR < VSCL (10) An output short can consist of the LP, LN, or LF terminals of the LED string being shorted, either to the battery terminal or to ground. Either a shorted LED string or output short will be latched and will only be cleared by pulling EN low for a period greater than the disable time, or by cycling the power. If either a shorted LED string, or an output short is detected, the A6265 will stop the switching action by pulling SG low. The fault flag FF1 will go high impedance and should be pulled up to the supply with suitable external pull-up resistors to indicate the fault. Either a shorted LED string or output short will be latched and will only be cleared by pulling EN low for a period greater than the disable time, or by cycling the power. The FF1 output can also be used with pull-up resistors and a P-channel MOSFET in the supply, to isolate the switching ele- ments and the load from the supply. This MOSFET should be connected, as shown in figure 3, with the source connected to the supply and the drain connected to the inductor of the converter. To FF1 VBAT To VIN

Figure 3. Example of a supply isolation MOSFET

Automotive High Current LED ControllerA6265 12Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. Two pull-up resistors are used to limit the voltage across the gate- source junction during high input voltages or load dump condi- tions. If the battery voltage is restricted, one resistor across the gate-source junction can be used. The FF1 provides a sink current of up to 1.3 mA. This circuit can be used to avoid most hazardous conditions and protect the circuit components from over-stress. Note that under extreme cases, the circuit cannot protect against certain fault con- ditions. For example, when any of the following occurs:

• In boost mode: ▫ If the cathode end of the LED string is shorted to VBAT, the LED sense resistor effectively appears between VBAT and ground. Depending on the current limit of the source supply or the input fuse rating, the fault current may damage the resistor. ▫ If the LF node is shorted to VBAT, the reference LED and the LED sense resistor effectively appear between VBAT and ground. Depending on the current limit of the source supply or the input fuse rating, the fault current may damage the resistor and /or the reference LED. ▫ If the cathode end of the LED string is shorted to ground. A fault current determined by the impedance of the shorting link (now effectively the LED sense resistor) flows through the power circuit. The fault current will either: be limited by the maximum switch current sense, V IDS , or if the source supply cannot maintain this current, be limited by the source supply. (The source supply will either: fold back, or if the current exceeds the input fuse rating, the fuse will blow, creating an open circuit).
• In buck-boost mode: ▫ If the cathode end of the LED string is shorted to ground, the LED sense resistor effectively appears between VBAT and ground. Depending on the current limit of the source supply or the input fuse rating, the fault current may damage the resistor. ▫ If the LF node is shorted to ground, the reference LED and the LED sense resistor effectively appear between VBAT and ground. The reference LED will be reversed biased and will probably be damaged. ▫ If the cathode end of the LED string (LP) is shorted to VBAT, the impedance of the short appears in parallel with the series combination of the series protection MOSFET and the LED sense resistor. This will tend to reduce the effective impedance of the LED sense resistor and correspondingly increase the LED current. To ensure the A6265 inputs (LP and LN) are not damaged dur- ing any of the above faults, it is necessary to add a differential resistor in series with the LN connection (between the sense resistor and the LN pin). This resistor value should be approxi- mately 150 Ω. If an output short is detected but it is necessary to keep the output active, the FF1 output can be pulled low. This will override the output disable but will not clear the fault.

Automotive High Current LED ControllerA6265 13Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A.

Application Information

External component selection is critical to the successful appli- cation of the LED driver. Although the inductor, the switching MOSFET, and the output capacitor are the most critical elements, the specification of the rectifying diode and sense resistors should also be carefully considered. The starting point for component selection is to define the maxi- mum LED current, the voltage across the LEDs, and the input operating voltage range. This then allows the average inductor current under worst case conditions to be calculated. The induc- tor value is then selected based on the acceptable inductor ripple current. The amount of ripple current will then determine the maximum inductor current under worst case conditions. From this current the switch current sense resistor can be calculated. LED Current Sense Resistor (R LS) If the voltage at the IREF pin, VIREF , is greater than 1V , or if IREF is tied to VREG, then the value of the LED current sense resistor, RLS , can be calculated from: R LS = VIDL / ILED(max) (11) where VIDL is the differential voltage across the LED current sense amplifier and ILED(max) is the maximum LED current. If VIREF is less than 1 V , then the value of the LED current sense resistor can be calculated from: RLS = VIREF / (10 × ILED(max) ) (12) The typical value for VIDL is 100 mV . Examples of various sense resistor values are given in table 2. In boost mode, the power loss in the current sense resistor is worse at the lowest input voltage: PLOSS = (VLED / VIN(min) ) × RLS × I 2 LED (13) In buck-boost mode, the power loss in the current sense resistor is worse at the lowest input voltage: P LOSS = ( [ VIN + VLED ] / VIN ) × RLS × I 2 LED (14) The resistors should be of a low inductance construction. Surface mount chip resistors are usually the most suitable, however, axial or radial leaded resistors can be used provided that the lead length is kept to a minimum. Inductor Selection Selecting the correct inductance is a balance between choosing a value that is small enough to help reduce size and cost, but high enough to ensure that the inductor current ripple is kept to an acceptable level. A reasonable target for the ripple current is 20% of the maximum average current. The inductor current equations differ slightly depending on whether the A6265 is configured as a boost or as a buck-boost converter.

• In a boost converter configuration: ▫ The maximum average inductor current is approximately: I L(av)(max) = ILED(max) × VLED / VIN(min) (15) ▫ The inductor current ripple is approximately: ILRIP = VIN × (VLED – VIN ) / (fOSC × L × VLED ) (16) ▫ The inductor value is therefore: L = VIN × (VLED –VIN ) / (fOSC × ILRIP × VLED ) (17)
• In a buck-boost configuration: ▫ The maximum average inductor current is approximately: IL(av)(max) = ILED(max) × (VIN(min) + VLED) / VIN(min) (18) ▫ The inductor current ripple is approximately: ILRIP = VIN × VLED / (fOSC × L × [VIN + VLED ] ) (19) ▫ The inductor value is therefore: L = VIN × VLED / (fOSC × ILRIP × [VIN + VLED ] ) (20) where: VLED is the voltage across the LED string, VIN is the supply voltage, VIN(min) is the minimum supply voltage, L is the inductor value, and f OSC is the oscillator frequency. With an internal oscillator frequency of 350 kHz, the value of the inductor for most cases will be between 20 and 50 μH. The maximum inductor current can then be calculated as: IL(PK) = IL(av)(max) + (IRIP / 2) (21)

Table 2. Sense Resistor Values

Automotive High Current LED ControllerA6265 14Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. This defines the minimum peak switch current as set by the switch current sense resistor. The current rating for the inductor should be greater, by some margin, than the peak value above, IL(PK) . When selecting an inductor from manufacturers datasheets, there are two current levels usually defined, the smallest value being the figure to work with:

• Saturation level, where the inductance value typically drops by 10%, or
• Temperature rise, where the part experiences a certain rise in temperature at full rated current. This parameter can be defined between a 20°C and 50°C rise in temperature. It is important to understand how manufacturers define the maximum operating temperature, because this can often incorporate the self-heating temperature rise. In most cases the limiting current is usually the saturation value. To improve efficiency, the inductor should also have low winding resistance, typically < 50 mΩ, and the core material will usually be ferrite, with low losses at the oscillator frequency. Recommended inductor manufacturers/series are:
• Coilcraft/ MSS1278T
• TDK/ SLF12575 type H Diode The diode should have a low forward voltage, to reduce conduction losses, and a low capacitance, to reduce switching losses. Schottky diodes can provide both these features if care- fully selected. The forward voltage drop is a natural advantage for Schottky diodes and reduces as the current rating increases. However, as the current rating increases, the diode capacitance also increases so the optimum selection is usually the lowest cur- rent rating above the required maximum, in this case I L(PK). Switch Current Sense Resistor (RSS) Neither the absolute value of the switch current nor the accuracy of the measurement is important, because the regulator will continuously adjust the switch current, within a closed loop, to provide sufficient energy for the output. For maximum accuracy the switch sense resistor value should be chosen to maximize the differential signal seen by the sense amplifier. The input limit of the sense amplifier, V IDS , and the maximum switch current, IS(max), therefore define the maximum value of the sense resistor as: RSS = VIDS / IS(max) (22) Where IS(max) is the maximum switch current and should be set above the maximum inductor current, IL(PK) . This represents the maximum measurable value of the switch (and inductor) current; however, the peak switch current will always be less than this, set by the control circuit, depending on the input voltage and the required load conditions. Because the switch current control is within a closed loop, it is possible to reduce the value of the sense resistor to reduce its power dissipa- tion. However this will reduce the accuracy of the regulated LED current. In Boost mode, the power loss in the switch sense resistor is worse at the lowest input voltage: P LOSS = (VLED [VLED – VIN(min)] / VIN(min)2) × RSS × I 2 LED (23) In Buck Boost mode, the power loss in the switch sense resistor is worse at the lowest input voltage: PLOSS = (VLED / VIN(min))(VLED + VIN(min)) × RSS × I 2 LED (24) External Switch MOSFET A logic-level N-channel MOSFET is used as the switch for the DC-to-DC converter. In the boost configuration the voltage at the drain of the MOSFET is equal to the maximum voltage across the string of LEDs. In the buck-boost configuration the output voltage is referenced to the positive supply. This means that the voltage at the drain of the MOSFET will reach a voltage equal to the sum of the LED voltage and the supply voltage. Under load dump conditions, up to 90 V may be present on this node. In this case the external MOSFET should therefore be rated at greater than 100 V . The peak switch current is defined by the maximum inductor cur- rent, I L(PK) . However in most cases the MOSFET will be chosen by selecting low on-resistance, which usually results in a current rating of several times the required peak current. In addition to minimizing cost, the choice of MOSFET should consider both the on-resistance and the total gate charge. The total gate charge will determine the average current required from the internal regulator and thus the power dissipation.

15Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. selecting the output capacitor. be low, allowing the use of ceramic capacitors. connection of the original isolation MOSFET. to the FET, or a second resistor added as shown in figure 3. Figure 4. Example of a supply isolation MOSFET

Automotive High Current LED ControllerA6265 16Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. Package LP 16-Pin TSSOP with Exposed Thermal Pad A

1.20 MAX

0.15 0.00 0.30 0.19 0.20 0.09 0.60 ±0.15

1.00 REF

C SEATING PLANEC0.10 16X

0.65 BSC

0.25 BSC

5.00±0.10 GAUGE PLANE SEATING PLANE A Terminal #1 mark area B For Reference Only; not for tooling use (reference MO-153 ABT) Dimensions in millimeters Dimensions exclusive of mold flash, gate burrs, and dambar protrusions Exact case and lead configuration at supplier discretion within limits shown B C Exposed thermal pad (bottom surface); dimensions may vary with device 6.10 0.65 0.45 1.70 3.00 3.00 Reference land pattern layout (reference IPC7351 SOP65P640X110-17M); All pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary to meet application process requirements and PCB layout tolerances; when mounting on a multilayer PCB, thermal vias at the exposed thermal pad land can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5) PCB Layout Reference View C Branded Face

3 NOM

Automotive High Current LED ControllerA6265 17Allegro MicroSystems, Inc. Worcester, Massachusetts 01615-0036 U.S.A. For the latest version of this document, visit our website: www.allegromicro.com Copyright ©2012, Allegro MicroSystems, Inc. Allegro MicroSystems, Inc. reserves the right to make, from time to time, such de par tures from the detail spec i fi ca tions as may be required to per- mit improvements in the per for mance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that the information being relied upon is current. Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the failure of that life support device or system, or to affect the safety or effectiveness of that device or system. The in for ma tion in clud ed herein is believed to be ac cu rate and reliable. How ev er, Allegro MicroSystems, Inc. assumes no re spon si bil i ty for its use; nor for any in fringe ment of patents or other rights of third parties which may result from its use.

Revision History

Revision Revision Date Description of Revision Rev. 5 July 26, 2012 Update application information